Resonant Flying Capacitor Multi-Level Converter Using Rerouted Capacitor

Abstract

The present invention relates to a DC-DC converter that reduces conduction losses and improves power density by introducing a repositioned capacitor, a GaN switch, and a negative voltage gate driver to have the same equivalent capacitance and resonant frequency in all phases, thereby solving the problem of limiting maximum output power caused by non-uniform resonant frequency and peak current resulting from different equivalent capacitances for each phase in a conventional resonant FCML (Flying Capacitor Multi-Level) converter.

Description

Technology Field

[0001] The present invention relates to a resonant FCML-based converter that reduces conduction losses and improves efficiency by introducing a repositioned capacitor stage to ensure a uniform resonant phase duration and peak phase current. Background Technology

[0003] Recently, switched capacitor (SC) converters have been attracting significant attention due to their high power density and efficiency. Since capacitors possess significantly higher energy density compared to inductors, research on SC converters that utilize only capacitors in the energy conversion process has been actively conducted.

[0004] However, to apply SC converters to high-power applications, critical issues such as electromagnetic interference (EMI) and charge sharing losses must be resolved. These problems stem from large current spikes occurring during switching transitions, which are inversely proportional to the switching frequency and capacitance.

[0005] Accordingly, Resonant Switched-Capacitor (ReSC) converters, which use inductors to prevent sudden current spikes and enable soft-charging operation, are widely utilized. Resonant switching operation not only reduces charge sharing losses and EMI issues but also improves power density and efficiency. Since the switching resistance is independent of the resonant operating frequency, resonant switching operation can reduce conduction losses even at low switching frequencies.

[0006] Due to these advantages, various types of ReSC topologies, such as Cascaded, Dickson, and Multi-resonant, have been proposed. Among these, the Flying Capacitor Multi-Level (FCML) converter is characterized by the lowest voltage stress applied to the switch.

[0007] However, conventional FCML topologies have limitations in improving power density due to non-uniform resonant frequencies during resonant operation. Since the equivalent capacitance of each phase differs, the resonant frequencies of the inductor and capacitor vary by phase. This multi-resonant frequency characteristic of FCML results in non-uniform phase currents and resonant periods in each phase. Consequently, the peak inductor current, which is limited by the inductor's saturation current, increases, thereby limiting the maximum output power.

[0008] FIGS. 1 to 3 are for explaining an FCML converter according to the prior art, FIG. 1 is a circuit diagram, FIG. 2 is a gate signal and inductor current, and FIG. 3 is an equivalent circuit diagram according to each phase.

[0009] Based on this, the operation of a resonant flying capacitor multilevel (FCML) converter according to conventional technology is explained as follows.

[0010] The FCML converter uses an input voltage (V IN ) is divided into N-levels using flying capacitors and switches. Pairs of switches operating complementarily (S NA / S NB The operation timing and inductor current of )(100) are as shown in FIG. 2. In N-to-1 step-down resonant switching operation, from phase φ1 to φ N The equivalent circuit that changes to is as shown in Fig. 3.

[0011] The resonant frequency (fres,φi) at phase φi is calculated as shown in Equation 1 below.

[0012]

[0013] This means that all flying capacitors have the same capacitance (C F1 = ... = C F(N-1) = C F With ), the flying capacitance is much smaller than the output capacitance (C F << C OUT ) is under the assumption.

[0014] Therefore, the resonance frequency relationship at each phase is derived as shown in Equation 2.

[0015]

[0016] For zero-current switching (ZCS) operation to eliminate VI overlap losses, the duration of each phase must be equal to half of the resonance period of each phase, as shown in Equation 3.

[0017]

[0018] Peak inductor current (I) of each phase pk,φi ) is derived as shown in Equation 4 by considering the charge balance of the flying capacitor.

[0019]

[0020] From mathematical equations 3 and 4, phase φ i The ratio of peak inductor current to output current k at φi It is derived as in mathematical formula 5.

[0021]

[0022] Conversion ratio (N) and peak inductor current to output current ratio k φi The relationship is shown in Fig. 4. In ideal resonant operation operating at a constant resonant frequency and phase duration, k φi The ratio is ð / 2, but N In a 3-in-1 FCML structure, this ratio is much larger. Since the inductor peak current must be smaller than the inductor saturation current, a large k φi The ratio limits the maximum output current.

[0023] In other words, the resonance frequency (f) of each phase res,φi ) means that all flying capacitors have the same capacitance (C F output capacitance (C) with ) OUTUnder the assumption that it is much smaller than ), it has different values ​​depending on the phase, which causes non-uniformity of the resonance period and peak current.

[0024] New solutions are required to overcome the limitations of such conventional technology. The problem to be solved

[0026] The present invention was conceived against this technical background and aims to solve problems such as maximum output power limitation, high conduction loss, and power density constraints caused by non-uniform resonant frequency and peak current resulting from the difference in equivalent capacitance for each phase in conventional resonant FCML converters. means of solving the problem

[0028] A resonant FCML-based converter using a relocated capacitor according to the present invention comprises: a main switching unit in which a first switch, a second switch, a third switch, and a fourth switch are sequentially connected in series from an input voltage toward the input terminal of an inductor; a first flying capacitor connected between the first and second switches of the main switching unit, a second flying capacitor connected between the second and third switches, and a third flying capacitor connected between the third and fourth switches; a fifth switch, one end of which is connected to ground connected to the input voltage, and the other end of which is connected to the first flying capacitor; and a relocated capacitor unit connected in series with the first to third flying capacitors, wherein the relocated capacitor unit comprises a relocated capacitor, sixth to eighth switches, and ninth and tenth switches connected in parallel with the relocated capacitor.It includes, wherein one end of the sixth switch is connected to a first connection point where the other end of the fifth switch and the lower end of the first flying capacitor are interconnected, and the other end is connected to a second connection point where the lower end of the second flying capacitor and the upper end of the relocation capacitor are interconnected, and one end of the seventh switch is connected to the second connection point, and the other end is connected to a third connection point where the lower end of the third flying capacitor and one end of the eighth switch are interconnected, one end of the eighth switch is connected to the third connection point, and the other end is connected to a fourth connection point between the fourth switch and the inductor, one end of the ninth switch is connected to the ground, and the other end is connected to the relocation capacitor, and one end of the tenth switch is connected to a fifth connection point where the other end of the ninth switch and the lower end of the relocation capacitor are interconnected, and the other end is connected to the fourth connection point, and in the first phase interval, the input voltage - the first switch - the first flying capacitor - the sixth A series current path is formed through the switch, the relocation capacitor, and the 10th switch, and in the 4th phase section, a series current path is formed through the ground, the 9th switch, the relocation capacitor, the 7th switch, the 3rd flying capacitor, and the 4th switch.

[0029] In the second phase section, a series current path is formed through the ground, the fifth switch, the first flying capacitor, the second switch, the second flying capacitor, the seventh switch, and the eighth switch, and in the third phase section, a series current path is formed through the ground, the fifth switch, the sixth switch, the second flying capacitor, the third switch, the third flying capacitor, and the eighth switch.

[0030] The first to fourth switches are sequentially turned on, and the ON period of each switch is 1 / 4 of the total switching cycle.

[0031] The DC voltage of the first flying capacitor above is three times V OUT , the DC voltage of the second flying capacitor is twice V OUT , the DC voltage of the third flying capacitor is 1 times V OUT It is charged to a voltage of , where, V OUT is the voltage value output from the above inductor.

[0032] The first and third flying capacitors have a first capacitance value, the relocation capacitor and the second flying capacitor have a second capacitance value, and the first capacitance value and the second capacitance value are the same.

[0033] The peak value of the current flowing through the inductor is the same in all phase sections, including the first phase section, the second phase section, the third phase section, and the fourth phase section.

[0034] The above-mentioned repositioning capacitor is charged in the first phase interval and discharged in the fourth phase interval.

[0035] A resonant FCML-based converter using a relocated capacitor according to the present invention comprises at least one of the switches being a GaN switch and driven by a negative voltage gate driver, wherein the negative voltage gate driver generates a negative voltage through a self-rectifying circuit using the voltage of the relocated capacitor.

[0036] In a resonant FCML-based converter using a relocated capacitor according to the present invention, the negative voltage can be generated at a voltage as high as the diode voltage from the lowest voltage of the relocated capacitor.

[0037] According to the resonant FCML-based converter using a relocated capacitor according to the present invention, the drain-source threshold voltage of a GaN switch driven by the negative voltage gate driver is adjusted, thereby increasing the ripple voltage range of the relocated capacitor. Effects of the invention

[0039] The present invention connects a relocated capacitor in series with existing flying capacitors to have the same equivalent capacitance in all phases, thereby achieving a uniform resonant frequency and peak current, which significantly reduces the ratio of peak inductor current to output current compared to existing ones, and also reduces conduction losses by reducing the series connection switch resistance. Brief explanation of the drawing

[0041] FIGS. 1 to 3 are for explaining an FCML converter according to the prior art, FIG. 1 is a circuit diagram, FIG. 2 is a gate signal and inductor current, and FIG. 3 is an equivalent circuit diagram according to each phase. Figure 4 shows the conversion ratio (N) and the peak inductor current to output current ratio k. φi It illustrates the relationship of. FIG. 5 is a circuit diagram of a resonant FCML-based converter using a relocated capacitor according to the present invention. Figures 6 and 7 show the voltage and inductor current of the fly capacitor and the equivalent circuit diagram in each phase. FIGS. 8 to 11 illustrate the current paths in each phase. Figure 12 shows the normal current path for each phase and the interference path between the auxiliary capacitor and the switches. Fig. 13 shows a GaN device (650V GaN Systems) According to , This is a graph showing the characteristics. Figure 14 shows the gate-source voltage of each switch and the flying capacitor, auxiliary capacitor voltage, and inductor current at that time. Figure 15 shows the operation waveform according to the output current of a two-way system with an adaptive self-charging negative voltage rectifier circuit applied. Specific details for implementing the invention

[0042] Embodiments of the present invention will be described in detail below with reference to the drawings. However, detailed descriptions of known functions or configurations that may obscure the essence of the present invention in the following description and the attached drawings are omitted. Additionally, throughout the specification, the term 'comprising' a component means that, unless specifically stated otherwise, it does not exclude other components but may include additional components.

[0043] Additionally, terms such as first, second, etc. may be used to describe various components, but said components should not be limited by said terms. said terms may be used for the purpose of distinguishing one component from another component. For example, without departing from the scope of the present invention, the first component may be named the second component, and similarly, the second component may be named the first component.

[0044] The terms used in this invention are used merely to describe specific embodiments and are not intended to limit the invention. The singular expression includes the plural expression unless the context clearly indicates otherwise. In this application, terms such as "comprising" or "comprising" are intended to specify the existence of the described features, numbers, steps, actions, components, parts, or combinations thereof, and should be understood as not precluding the existence or addition of one or more other features, numbers, steps, actions, components, parts, or combinations thereof.

[0045] Unless specifically defined otherwise, all terms used herein, including technical or scientific terms, have the same meaning as generally understood by those skilled in the art to which the present invention pertains. Terms such as those defined in commonly used dictionaries should be interpreted as having a meaning consistent with their meaning in the context of the relevant technology, and should not be interpreted in an ideal or overly formal sense unless explicitly defined in this application.

[0047] The multiple resonant frequency characteristics of conventional resonant FCML converters occur because the equivalent series-connected capacitors differ in each phase, as explained in the prior art section. Therefore, the converter of the present invention (Fig. 5) has a relocated capacitor (C FO ) is introduced to match the equivalent capacitance in order to achieve a constant resonant frequency.

[0048] The equivalent circuit for each phase is shown in FIG. 7, and the detailed current flow is illustrated in FIG. 8 to 11. In the present invention, the inductor current always passes through two capacitors connected in series from phase φ1 to φ4. The converter resonant frequency according to the present invention from phase φ1 to phase φ4 is derived as shown in Equations 6 to 9.

[0049]

[0050]

[0051]

[0052]

[0053] From mathematical equations 6 to 9, the constant resonant frequency condition is derived as in mathematical equation 10.

[0054]

[0055] Here is. As a result, four capacitors are combined into two pairs (C) with the same value (10).A and C B By matching with ), uniform resonant operation and uniform resonant inductor current can be achieved. Switching operation with a constant resonant frequency and uniform phase duration results in a peak inductor current to output current ratio k φi It reduces by 20%, which is ð / 2 from 1.897 of the existing 4-to-1 resonant FCML topology ( It decreases to 1.571), which is shown in Fig. 4.

[0056] The converter of the present invention also improves efficiency by reducing the resistance of the series-connected switches. FIGS. 8 through 11 show the current paths of each phase. In the second phase section φ2 and the third phase section φ3, the inductor current flows through four series-connected switches, similar to a conventional 4-to-1 FCML converter. On the other hand, in the first phase section φ1 and the fourth phase section φ4, the inductor current flows through three series-connected switches along with a relocated capacitor. Detailed switch conduction loss P COND It can be derived as in mathematical formula 11:

[0057]

[0059] Here, T SW is the switching period, and R SW,φi is phase φ i In this case, it is the total equivalent series resistance of the switches. Assuming that all switches have the same resistance, the total equivalent series resistance R of the present invention is SW,φi It can be expressed as in mathematical formula 12.

[0060]

[0061] Since the phase duration and peak inductor current are the same, the phase duration ratio t φi / T SW is equal to 1 / 4, and k of the present invention φiThe ratio is equal to π / 2. From Equations 11 and 12, the switch conduction loss P of the proposed topology COND,Proposed It can be derived as shown in mathematical formula 13.

[0062]

[0063] On the other hand, since the inductor current always flows through the four switches, R of the conventional FCML converter SW,φi It is derived as shown in mathematical formula 14.

[0064]

[0065] From mathematical equations 2, 3, 5, 11, and 14, the switch conduction loss P of a conventional FCML converter COND,Conventional It is derived from mathematical formula 15.

[0066]

[0067] The present invention reduces conduction loss by approximately 15% compared to the prior art (existing FCML converter). The main reason for the improvement is that the series-connected switch resistance is reduced, as expressed in Equations 12 and 14. In addition, the reduced RMS inductor current of the proposed topology also improves conduction loss.

[0069] Hereinafter, a resonant FCML-based converter using a relocated capacitor according to the present invention, based on the above description, will be described. First, the circuit configuration of the converter is explained with reference to FIG. 5 as follows.

[0070] The converter is input voltage (V IN A main switching unit in which a first switch (s1), a second switch (s2), a third switch (s3), and a fourth switch (s4) are sequentially connected in series from ) toward the input terminal of an inductor (L), and the first and second switches (s) of the main switching unit 1, s2) The first flying capacitor (C) connected between F1 ), second and third switches(s 2, s3) The second flying capacitor (C) connected betweenF2 ), and the 3rd and 4th switches(s 3, s4) The third flying capacitor (C) connected between F3 ) and, one end is the above input voltage (V IN It is connected to ground connected to ), and the other end is connected to the first flying capacitor (C F1 A fifth switch (s5) connected to ) and the first to third flying capacitors (C F1, C F2, C F3 A relocation capacitor section connected in series with ), wherein the relocation capacitor section is a relocation capacitor (C F0 ), 6th to 8th switches (s6, s7, s8), and the relocation capacitor (C F0 The 9th and 10th switches (s9, s) connected in parallel with ) 10 It is configured to include a relocation capacitor section including ). Here, one end of the sixth switch (s6) is connected to the other end of the fifth switch (s5) and the first flying capacitor (C F1 The lower end of ) is connected to a first connection point that is interconnected, and the other end is connected to a second flying capacitor (C F2 The bottom of ) and the repositioning capacitor (C F0 The upper end of ) is connected to a second connection point that is interconnected. One end of the seventh switch (s7) is connected to the second connection point, and the other end is connected to the third flying capacitor (C F3 The bottom of ) and one end of the 8th switch (s8) are connected to a third connection point where they are interconnected. One end of the 8th switch (s8) is connected to the third connection point, and the other end is connected to a fourth connection point between the 4th switch (s4) and the inductor (L). One end of the 9th switch (s9) is connected to ground, and the other end is connected to the relocation capacitor (C F0 It is connected to ). The 10th switch(s 10 ) is one end of the other end of the ninth switch (s9) and the relocation capacitor (C F0 The lower end of ) is connected to the fifth connection point that is interconnected, and the other end is connected to the fourth connection point.

[0071] Meanwhile, FIGS. 6 and 7 show the voltage and inductor current of the fly capacitor and the equivalent circuit diagram at each phase, and FIGS. 8 to 11 show the current path at each phase. Based on this, the operation of a resonant FCML-based converter using a repositioned capacitor according to the present invention is described as follows.

[0072] In the first phase (Phase φ1), as shown in FIG. 8, the input voltage (V IN From ) the first switch (s1), the first flying capacitor (C F1 ), 6th switch (s6), relocation capacitor (C F0 ), 10th switch(s 10 A series current path is formed through the ) and the inductor (L), and current flows. At this time, the current forms a rerouted path that passes through only three switches.

[0073] In the second phase (Phase φ2), as shown in FIG. 9, the fifth switch (s5) and the first flying capacitor (C) from ground F1 ), second switch (s2), second flying capacitor (C F2 A series current path is formed through the 7th switch (s7), the 8th switch (s8), and the inductor (L), and current flows. At this time, the current forms a general path passing through four switches, as in a conventional FCML converter.

[0074] In the third phase (Phase φ3), as shown in FIG. 10, the fifth switch (s5), the sixth switch (s6), and the second flying capacitor (C) from ground F2 ), third switch (s3), third flying capacitor (C F3 A series current path is formed through the 8th switch (s8) and the inductor (L), and current flows. In this phase, the current also forms a general path through the four switches.

[0075] In the fourth phase (Phase φ4), as shown in FIG. 11, the ninth switch (s9) and the relocation capacitor (C) from ground F0 ), 7th switch (s7), 3rd flying capacitor (C F3 A series current path is formed through the ), the fourth switch (s4) and the inductor (L), and current flows. As with the first phase, in this phase, the current also forms a relocated path that passes through only three switches.

[0076] By the above operation, the relocated capacitor (C F0 The capacitor is charged in the first phase and discharged in the fourth phase, and operates in series with other flying capacitors in the second and third phases. Due to the operation of this repositioned capacitor, the same equivalent capacitance is formed in all phases, allowing for uniform resonant frequency characteristics to be obtained.

[0077] In particular, in the first and fourth phases, a current path is formed that passes through only three switches, thereby reducing conduction losses, and a uniform peak current is formed due to the same resonant frequency in all phases, which can be achieved to improve power density.

[0078] Meanwhile, in configuring the resonant FCML-based converter of the present invention as described above, an interference problem may occur between switches connected in parallel with the charging voltage of the auxiliary capacitor.

[0079] Figure 12 shows the normal current path (blue) and the interference path (red) between the auxiliary capacitor and the switches for each phase. Since general switches have a body diode path between the drain and source, current can flow even when the switch+ is turned off if a voltage of approximately -0.7V is applied. This body diode path creates greater conduction loss compared to the switch path, so this must be prevented. The auxiliary capacitor voltage and switch threshold voltage conditions for preventing current from flowing through the body diode path for each phase are given by Equation 16.

[0080]

[0081] Auxiliary capacitor (V CF0 The ripple voltage of ) increases proportionally to the output current and has a relationship with the drain-source threshold voltage of each switch as shown in Equation 17.

[0082]

[0083] In the present invention, the reverse current threshold voltage ( GaN switches and negative voltage gate drivers are utilized to increase ). A typical silicon switch is -0.7V by the body diode. It has. However, GaN switches do not have a body diode, and reverse current flows through a phenomenon called self-commutated reverse conduction. When reverse current flows, there is no current path before the switch conducts, so the drain voltage decreases as the parasitic capacitor at the switch's drain is discharged. The drain voltage is the gate-drain threshold voltage ( If it decreases further than ), current can flow between the source and drain, causing the reverse path to conduct. At this time, the gate-source off voltage , threshold voltage between gate and drain , reverse current threshold voltage( ) has the same relationship as mathematical formula 18.

[0084]

[0085] And, Fig. 13 shows a GaN device (650V GaN Systems) According to , As a characteristic, the relationship of Equation 16 is shown for each gate voltage. In the present invention, which uses a negative voltage gate driver and a GaN switch, the characteristic of being able to adjust the drain-source threshold voltage of the GaN switch is utilized.

[0086] Figure 14 shows the gate-source voltage of each switch and the corresponding flying capacitor, auxiliary capacitor voltages, and inductor current. The gate voltage of the S5 GaN switch with a negative voltage gate driver applied is in the off-state. It acquires a voltage. Therefore, by adjusting the negative voltage of the gate driver, the ripple voltage range of the auxiliary capacitor can be widened, and the maximum output current can be increased.

[0087] Equation 19 below shows the relationship between the drain-source threshold voltage and the auxiliary capacitor voltage of a GaN switch with a negative voltage gate driver applied.

[0088]

[0089] Meanwhile, a negative voltage power supply for a gate driver to increase the drain-source threshold voltage of a GaN switch ( ) can be generated not only by external application but also through a self-rectification circuit using auxiliary capacitor voltage( This is explained with reference to Fig. 15 as follows.

[0090] Fourth phase ( When the 9th switch (S9) is in the ON state in ), the auxiliary capacitor voltage (V CF0 ) is discharged as a negative voltage and transferred as output current. Auxiliary capacitor voltage (V CF0The voltage drops to negative, and the negative voltage power supply through the diode via the adaptive self-charging negative voltage rectifier circuit proposed in this invention Discharges.

[0091] The voltage is the lowest voltage of the auxiliary capacitor as shown in Equation 20 through the negative voltage rectifier circuit ( diode voltage at ) It is discharged at a voltage as high as )

[0092]

[0093] Therefore, through mathematical equations 16 to 20, the maximum chargeable voltage of the auxiliary capacitor satisfies an inequality that always holds true as in mathematical equation 20, so a gate driver that generates a negative voltage adaptively to the output current can be made.

[0094]

[0096] Meanwhile, Figure 16 shows the operation waveform according to the output current of a two-way system with an adaptive self-charging negative voltage rectifier circuit applied. It can be confirmed that it is currently operating at a negative voltage having an absolute value lower than the diode voltage (0.3V) and the ripple voltage of the auxiliary capacitor. At this time, the fifth switch (S5) is The switch can be maintained in the off-state with an additional margin equal to the voltage. That is, the adaptive self-charging negative voltage of the present invention can generate an optimal off-state gate voltage that increases in proportion to the voltage of the auxiliary capacitor changing according to the output current, thereby optimizing gate charge losses.

[0098] The present invention has been described above with reference to various embodiments. Those skilled in the art will understand that the present invention may be implemented in modified forms without departing from the essential characteristics of the invention. Therefore, the disclosed embodiments should be considered in an illustrative rather than a restrictive sense. The scope of the invention is defined by the claims, not by the foregoing description, and all variations within the scope of the claims should be interpreted as being included in the invention.

Claims

Claim 1 A main switching unit in which a first switch, a second switch, a third switch, and a fourth switch are sequentially connected in series from the input voltage toward the input terminal of an inductor; a first flying capacitor connected between the first and second switches of the main switching unit, a second flying capacitor connected between the second and third switches, and a third flying capacitor connected between the third and fourth switches; a fifth switch, one end of which is connected to ground connected to the input voltage, and the other end of which is connected to the first flying capacitor; and a relocation capacitor unit connected in series with the first to third flying capacitors, wherein the relocation capacitor unit comprises a relocation capacitor, sixth to eighth switches, and ninth and tenth switches connected in parallel with the relocation capacitor.It includes, wherein one end of the sixth switch is connected to a first connection point where the other end of the fifth switch and the lower end of the first flying capacitor are interconnected, and the other end is connected to a second connection point where the lower end of the second flying capacitor and the upper end of the relocation capacitor are interconnected, the seventh switch has one end connected to the second connection point and the other end connected to a third connection point where the lower end of the third flying capacitor and one end of the eighth switch are interconnected, the eighth switch has one end connected to the third connection point and the other end connected to a fourth connection point between the fourth switch and the inductor, the ninth switch has one end connected to the ground and the other end connected to the relocation capacitor, the tenth switch has one end connected to a fifth connection point where the other end of the ninth switch and the lower end of the relocation capacitor are interconnected, and the other end connected to the fourth connection point, and in the first phase section, the input voltage - the first switch - the first flying capacitor - the sixth A resonant FCML-based converter using a relocated capacitor, wherein a series current path is formed through the switch, the relocated capacitor, and the 10th switch, and in the 4th phase section, a series current path is formed through the ground, the 9th switch, the relocated capacitor, the 7th switch, the 3rd flying capacitor, and the 4th switch. Claim 2 A resonant FCML-based converter using a relocated capacitor, wherein in the first phase section, a series current path is formed through the ground - the fifth switch - the first flying capacitor - the second switch - the second flying capacitor - the seventh switch - the eighth switch, and in the third phase section, a series current path is formed through the ground - the fifth switch - the sixth switch - the second flying capacitor - the third switch - the third flying capacitor - the eighth switch. Claim 3 A resonant FCML-based converter using a repositioned capacitor according to claim 1, wherein the first to fourth switches are sequentially turned on, and the ON period of each switch is 1 / 4 of the total switching cycle. Claim 4 In claim 1, the DC voltage of the first flying capacitor is three times V OUT , the DC voltage of the second flying capacitor is twice V OUT , the DC voltage of the third flying capacitor is 1 times V OUT It is charged to a voltage of , where, V OUT A resonant FCML-based converter using a relocated capacitor, wherein is the voltage value output from the inductor. Claim 5 A resonant FCML-based converter using a relocated capacitor, wherein, in claim 1, the first and third flying capacitors have a first capacitance value, the relocated capacitor and the second flying capacitor have a second capacitance value, and the first capacitance value and the second capacitance value are identical to each other. Claim 6 A resonant FCML-based converter using a repositioned capacitor, wherein, in all phase intervals including the first phase interval, the second phase interval, the third phase interval, and the fourth phase interval, the peak value of the current flowing through the inductor is the same. Claim 7 A resonant FCML-based converter using a relocated capacitor, wherein the relocated capacitor is charged in a first phase interval and discharged in a fourth phase interval in claim 1. Claim 8 A resonant FCML-based converter using a relocated capacitor according to claim 1, wherein at least one of the switches is a GaN switch and is driven by a negative voltage gate driver, and the negative voltage gate driver generates a negative voltage through a self-rectification circuit using the voltage of the relocated capacitor. Claim 9 In claim 8, a resonant FCML-based converter using a relocated capacitor, wherein the negative voltage is generated from the lowest voltage of the relocated capacitor to a voltage as high as the diode voltage. Claim 10 A resonant FCML-based converter using a relocated capacitor, wherein, in claim 8, the drain-source threshold voltage of a GaN switch driven by the negative voltage gate driver is adjusted so as to increase the ripple voltage range of the relocated capacitor.

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