Walk Error Compensation Circuit of ToF LiDAR

Abstract

The LiDAR work error compensation circuit of the present invention includes a transimpedance amplifier that receives an optical signal reflected back from an avalanche photodiode (APD), amplifies the output current signal, converts it into a voltage signal, and outputs it; an automatic gain control amplifier that automatically gain-controls the voltage signal output by the transimpedance amplifier to amplify it into a signal of constant amplitude and outputs it; and a zero-crossing discriminator that amplifies the amplitude-corrected signal output by the automatic gain control amplifier and differentiates it to extract the ToF timing.

Description

Technology Field

[0001] The present invention relates to a technology for compensating for work errors of a LiDAR sensor. Background Technology

[0002] Time of Flight (ToF) LiDAR (Light Detection and Ranging) calculates the distance between the LiDAR and an object by emitting a high-energy laser pulse for a very short duration and measuring the time it takes for the pulse to be reflected by an object and return to the LiDAR. Since laser pulses travel at the speed of light, the distance to the target object can be determined by dividing the measured time in half and then dividing by the speed of light.

[0003] The timing used by the LiDAR's Time-to-Digital Converter (TDC) occurs when the level of the signal output by the Avalanche Photodiode (APD), which detects the reflected laser pulse, exceeds a certain threshold. However, a consistent threshold cannot be used because the amplitude of the received signal varies depending on the distance to the target object or optical characteristics such as the surface reflectivity and angle of incidence. This variation in amplitude creates an error in the timing at which the signal is recognized as an active signal. The error in distance measurement results caused by this minute difference in ToF timing is called a walk error. Light travels 1 cm in a vacuum in 33.4 ps. Therefore, even if the signal crosses the threshold late by just a very short time of 1 ns, an error of 15 cm occurs in the distance measurement result. Therefore, when extracting ToF timing without separate compensation, the lidar may perceive an object that is bright, has a flat surface, and is positioned perpendicular to the direction viewed by the lidar as being closer, while conversely, it may perceive a dark, curved, and obliquely positioned object as being further away, even though the two objects are separated by the same distance. For the same reason, the lidar may perceive objects that are close as being closer than their actual distance, and objects that are far away as being further away than their actual distance.

[0004] The most fundamental method for determining reception timing using signals received from an APD is the leading edge discriminator method. This structure involves a transimpedance amplifier (TIA) receiving the current input from the APD, converting it into voltage, and amplifying it (or outputting it after further amplification via an appropriate post-amplifier if necessary), and using a single comparator with a single threshold line to output the ToF timing as the TDC. This is a simple and easy-to-use method for extracting ToF timing. However, the leading edge discriminator method does not compensate for work errors. Consequently, work errors significantly occur depending on the amplitude of the input signal. As higher threshold levels are used for greater noise immunity, work errors become more aggravated and the dynamic range decreases.

[0005] Two threshold lines can be used to compensate for work errors. By using two comparators with different threshold lines and outputting two leading edge timings as the TDC, the signal start point can be estimated. This method has the advantage of being relatively simple in structure compared to other methods and capable of compensating for work errors even when the signal is saturated. However, increasing the gap between the two threshold lines to achieve a higher work error compensation rate involves a trade-off in which the dynamic range is limited. Additionally, regarding the signal input to the comparator, V, which is the comparator's threshold line threshold Countermeasures are needed for the occurrence of another delay time depending on how quickly it is passed, that is, for the comparator overdrive characteristics.

[0006] As another method to compensate for work errors, a Constant Fraction Discriminator (CFD) method utilizing the intersection of attenuated and delayed signals can be used. This method enables operation independent of signal amplitude, allowing for the elimination of work errors as long as the signal rise time remains constant. However, in practice, this method also generates additional work errors due to the comparator's overdrive characteristics. To reduce this dependency, a CFD method incorporating an Auto Gain Control (AGC) amplifier circuit at the input is being proposed. The AGC amplifier processes the input signal into a signal with nearly uniform amplitude and slope, outputting it to the ToF timing discriminator circuit, thereby reducing additional work errors caused by the comparator's characteristics.

[0007] There is also another method to compensate for work errors using the Zero Crossing Discriminator (ZCD) method, which identifies the point at which the polarity of the signal's derivative changes by utilizing differentiation tools such as CR differentiation networks. The ZCD method is one of the good methods for capturing the location of the maximum amplitude of a received signal without significant delay. However, it has the problem of failing to properly detect peak points when the signal is excessively strong (saturating) or excessively weak. Furthermore, unlike the previously mentioned methods, it does not compensate for additional delay time that may occur in the comparator depending on the signal amplitude and slope. The problem to be solved

[0008] The present invention aims to provide a work error compensation circuit that uses an automatic gain control amplifier and a zero-crossing discriminator to increase the range of detectable input signals while maintaining the same amplitude and slope of the signals.

[0009] In addition, another objective of the present invention is to provide a work error compensation circuit capable of reducing additional time delays caused by the overdrive characteristics of a comparator within a zero-crossing discriminator. means of solving the problem

[0010] A lidar work error compensation circuit according to one aspect of the present invention includes a transimpedance amplifier, an automatic gain control amplifier, and a zero-crossing discriminator.

[0011] A transimpedance amplifier receives the optical signal reflected back from an avalanche photodiode (APD), amplifies the output current signal, converts it into a voltage signal, and outputs it.

[0012] An automatic gain control amplifier automatically gain-controls the voltage signal output by a transimpedance amplifier to amplify it into a signal of constant amplitude and output it.

[0013] The zero-crossing discriminator amplifies the amplitude-corrected signal output by the auto-gain control amplifier and then differentiates it to extract the ToF timing.

[0014] According to an additional aspect of the present invention, the lidar work error compensation circuit may further include an AC coupling capacitor disposed before the transimpedance amplifier to remove the DC signal component from the output signal of the avalanche photodiode.

[0015] Specifically, the transimpedance amplifier may include a first OP amplifier that amplifies an input current signal and converts it into a voltage signal for output, a first feedback resistor connected to the output terminal and the inverting input terminal of the first OP amplifier to determine the current-voltage exchange ratio, and a first feedback capacitor connected in parallel with the first feedback resistor to stabilize the output waveform of the transimpedance amplifier.

[0016] Additionally, the automatic gain control amplifier may include a peak detector that maintains and outputs peak information of the output signal of the transimpedance amplifier, a gain calculator that calculates a gain using the peak information output from the peak detector, a variable gain amplifier that variably amplifies the output signal of the transimpedance amplifier by applying the gain calculated by the gain calculator, and a delay line that delays the output signal of the transimpedance amplifier by the amount of time generated by the gain calculator while calculating the gain and inputs it to the variable gain amplifier.

[0017] Additionally, the peak detector includes a second OP amplifier, a diode, a second feedback capacitor, and a second feedback resistor, wherein the output terminal of the transimpedance amplifier is connected to the non-inverting input terminal of the second OP amplifier, the input terminal of the diode is connected to the output terminal of the second OP amplifier, the output terminal of the diode is connected to the output terminal of the peak detector, the second feedback capacitor is connected to the output terminal of the diode and the inverting input terminal of the second OP amplifier to store peak information, and the second feedback resistor may be connected in parallel with the second feedback capacitor. Effects of the invention

[0018] According to the present invention, a work error compensation circuit can increase the range of detectable input signals by using an automatic gain control amplifier and a zero-crossing discriminator, while maintaining the same amplitude and slope of the signal.

[0019] Additionally, according to the present invention, the work error compensation circuit can also reduce additional time delay caused by the overdrive characteristics of the comparator in the zero-crossing discriminator. Brief explanation of the drawing

[0020] FIG. 1 is a block diagram of a ToF rider work error compensation circuit according to one aspect of the present invention. FIG. 2 briefly illustrates an example of the circuit configuration of the transimpedance amplifier of the present invention. FIG. 3 briefly illustrates an example of the peak detector circuit configuration of the present invention. Figure 4 briefly illustrates an example of the gain calculator circuit configuration of the present invention. FIG. 5 briefly illustrates an example of the configuration of a variable gain amplifier circuit of the present invention. Figure 6 briefly illustrates an example of the zero-crossing discriminator circuit configuration of the present invention. Specific details for implementing the invention

[0021] The foregoing and additional aspects are embodied in the embodiments described with reference to the attached drawings. It is understood that the components of each embodiment may be combined in various ways within the embodiment unless otherwise stated or contradictory. Each block in the block diagram may represent a physical part in some cases, but in others, it may be a logical representation of a part of the function of a single physical part or a function spanning multiple physical parts. Sometimes, the entity of a block or part thereof may be a set of program instructions. These blocks may be implemented in whole or in part by hardware, software, or a combination thereof.

[0022] FIG. 1 illustrates a block diagram of a ToF LiDAR work error compensation circuit according to one aspect of the present invention. A LiDAR work error compensation circuit (1) according to one aspect of the present invention includes a transimpedance amplifier (11), an automatic gain control amplifier (13), and a zero-crossing discriminator (15).

[0023] According to an additional aspect of the present invention, the lidar work error compensation circuit (1) may further include an AC coupling capacitor (17) disposed before the transimpedance amplifier (11) to remove the DC signal component from the output signal of the avalanche photodiode.

[0024] The current signal output from the APD biased at approximately 200V is input to the transimpedance amplifier (11) through a 0.1μF AC coupling capacitor. This structure allows the AC coupling capacitor (17) to cancel out DC components such as the dark current, ambient light, and leakage of the APD, provided that there is sufficient time for the sudden fluctuation in the amount of charge generated while passing the APD signal to be recovered, thereby enabling the maximum utilization of the APD dynamic range.

[0025] The transimpedance amplifier (11) receives the optical signal reflected back from the avalanche photodiode (APD), amplifies the output current signal, converts it into a voltage signal, and outputs it.

[0026] FIG. 2 briefly illustrates an example of the circuit configuration of the transimpedance amplifier of the present invention. As shown in FIG. 2, the transimpedance amplifier (11) can be composed of an inverting amplifier and a voltage follower configured using an OP amplifier.

[0027] Specifically, as illustrated in FIG. 2, the transimpedance amplifier (11) comprises a first OP amplifier (111) that amplifies an input current signal and then converts it into a voltage signal for output, a first feedback resistor (113) connected to the output terminal and the inverting input terminal of the first OP amplifier (111) to determine the current-voltage exchange ratio, a first feedback capacitor (115) connected in parallel with the first feedback resistor (113) to stabilize the output waveform of the transimpedance amplifier (11), and a voltage follower using negative feedback.

[0028] The value of the first feedback resistor (113) of the first OP amplifier (111) of the transimpedance amplifier (11) determines the gain of the transimpedance amplifier (11), that is, the current-voltage exchange ratio, as transimpedance. For example, if a resistor of 25k ohm is used as the first feedback resistor (113), the transimpedance amplifier (11) outputs a voltage of 25 mV when a current of 1 μA is input from the APD, and outputs a voltage of 1 V when a current of 40 μA is input. Since the maximum gain of this transimpedance amplifier (11) is limited by the output swing, slew rate, GBWP (Gain Bandwidth Product), etc. of the first OP amplifier (111) of the transimpedance amplifier (11), a feedback resistor that is too large cannot be used. Conversely, considering the noise immunity after conversion to a voltage signal, which is relatively more susceptible to noise interference than a current signal, it is likewise difficult to apply feedback resistors that are too small.

[0029] The transimpedance amplifier (11) of the present invention connects a feedback capacitor in parallel with a feedback resistor to stabilize the output waveform. If there is no capacitor or the capacitance is too small, the output becomes unstable, but if the capacitance is too large, the signal is delayed and dulled. Therefore, it is desirable for the first feedback capacitor (115) to have a capacitance value of tens to hundreds of fF.

[0030] The voltage follower of the transimpedance amplifier (11) acts as a buffer that takes the output voltage of the inverting amplifier as the input voltage and transmits it to the output terminal of the transimpedance amplifier (11).

[0031] The automatic gain control amplifier (13) automatically gain-controls the voltage signal output by the transimpedance amplifier (11) to amplify it into a signal of constant amplitude and output it.

[0032] The automatic gain control amplifier (13) includes a delay line (131), a variable gain amplifier (133), a peak detector (135), and a gain calculator (137) as shown in FIG. 1.

[0033] The peak detector (135) maintains and outputs peak information of the output signal of the transimpedance amplifier (11).

[0034] FIG. 3 briefly illustrates an example of the peak detector circuit configuration of the present invention. As shown in FIG. 3, the peak detector (135) includes a second OP amplifier (1351), a diode (1357), a second feedback capacitor (1353), and a second feedback resistor (1355).

[0035] The output terminal of the transimpedance amplifier (11) is connected to the non-inverting input terminal of the second OP amplifier (1351), the input terminal of the diode (1357) is connected to the output terminal of the second OP amplifier (1351), the output terminal of the diode (1357) is connected to the output terminal of the peak detector (135), the second feedback capacitor (1353) is connected to the output terminal of the diode (1357) and the inverting input terminal of the second OP amplifier (1351) to store peak information, and the second feedback resistor (1355) can be connected in parallel with the second feedback capacitor (1353).

[0036] The peak detector (135) maintains the peak information of the signal output by the transimpedance amplifier (11) even after the output signal of the APD has ended, so that the gain calculator (137) can calculate and input a suitable gain to the variable gain amplifier (133).

[0037] The peak detector (135) has a structure in which a diode (1357) is added to a voltage follower as shown in FIG. 3. This structure exhibits superior peak voltage storage capability compared to other peak detectors composed only of a combination of diode (D), resistor (R), and capacitor (C) without an OP amplifier, because the second OP amplifier (1351) receives direct feedback of the peak voltage stored in the second feedback capacitor (1353) by bypassing the diode (1357).

[0038] The values ​​of the second feedback capacitor (1353) that stores peak information and the second feedback resistor (1355) connected in parallel therewith can be determined such that the original purpose of storing peak voltage for a certain period of time of the peak detector is achieved, while being sufficiently discharged before the next APD signal sequence begins so as not to affect the next signal.

[0039] The gain calculator (137) can calculate the gain using peak information output from the peak detector (135). FIG. 4 briefly illustrates an example of the circuit configuration of the gain calculator of the present invention. As shown in FIG. 4, the gain calculator (137) may be configured to include a log amplifier, an anti-log amplifier, an inverting summing amplifier, and a non-inverting amplifier with added subtraction function. The gain calculator (137) of the present invention is characterized by utilizing the fact that multiplication of an analog signal is equivalent to applying a logarithmic function to the analog signal, adding it, and then returning it through an exponential function.

[0040] The gain calculator (137) circuit configured as shown in FIG. 4 has a response time of approximately 100 to 200 ns, which is slower than the APD output signal having a length of approximately 20 ns to 40 ns. As a result, it is necessary to make the variable gain amplifier (133) receive the APD output signal with a delay until the gain calculator (137) completes the calculation of the gain to be input to the variable gain amplifier (133). Additionally, it is necessary to make the peak detector (135) maintain peak information for that duration of the delay. However, this makes it possible for the gain calculator (137) to calculate the gain finely and to variablely amplify the output signal of the variable gain amplifier (133) so that the amplitude remains constant despite fluctuations in the APD output amplitude.

[0041] If the values ​​of the resistors used in the gain calculator (137) circuit of Fig. 4 are the same and the diodes are also designed in the same way, the output gain of the gain calculator (137) is calculated as in <Equation 1>.

[0042]

[0043] At this time V C It is calculated as in <Equation 2> by a non-inverting amplifier.

[0044]

[0045] This V C It was added to the gain calculator (137) according to the determination of the gain control parameter to be described later. By substituting <Equation 2> into <Equation 1>, the output gain of the gain calculator (137) can be calculated as <Equation 3>.

[0046]

[0047] Since Vg of the VCA824 of the variable gain controller shown in Fig. 5 has an input range of -1 V to +1 V, it is necessary to regulate the amplitude of the gain calculator output to be input as Vg. Therefore, the power supply to the OP amp after the log amplifier output, where the signal amplitude decreases, is limited to +3.3 V, -3.3 V, or lower, thereby inducing amplitude clamping in the anti-log amplifier where the amplitude increases again, and protecting against abnormal operation such as calculation errors and noise caused by exceeding the range when the output gain is input to the VCA824.

[0048] FIG. 5 briefly illustrates an example of a variable gain amplifier circuit configuration of the present invention. The variable gain amplifier (133) variably amplifies the output signal of the transimpedance amplifier (11) by applying the gain calculated by the gain calculator (137).

[0049] The variable gain controller illustrated in FIG. 5 variably amplifies the signal received from the transimpedance amplifier by applying the appropriate gain calculated by the gain calculator (137) described above. That is, the variable gain amplifier (133) plays a substantial role in reducing additional time delay caused by the overdrive characteristics of the comparator in the zero-crossing discriminator (15) by keeping the amplitude and slope of the signal as equal as possible.

[0050] The voltage-controlled variable amplifier (hereinafter VCA824) used in the circuit illustrated in Fig. 5 is capable of controlling gain much more precisely and linearly than the method of controlling gain by replacing the feedback resistor of a general OP amp amplifier with a variable resistor such as a TR or FET.

[0051] The VCA824 sets the maximum gain through Rg and Rf, and selects the actual gain to be applied through the Vg input, which is the gain calculated within that range. The minimum gain is 0, and the maximum gain is defined as 2Rf / Rg, up to 240 V / V. This range is mapped back to Vg from -1 V to +1 V. Since the range and bandwidth of the input signal amplitude are limited as the maximum gain of the VCA824 is set higher, it is important to appropriately select the variable amplification range by compromising with the fixed gains of the amplifiers before and after the variable gain amplifier (133) or other parameters.

[0052] The variable gain controller receives a signal delayed by about 100 ns, which is due to the time taken to scan the signal of the APD until one cycle is completely finished and to calculate the appropriate gain in order to calculate the gain to be input to the VCA824 in the gain calculator (137).

[0053] The delay line (131) delays the output signal of the transimpedance amplifier (11) by the time the gain calculator (137) generates the gain and inputs it to the variable gain amplifier (133).

[0054] FIG. 6 briefly illustrates an example of the zero-crossing discriminator circuit configuration of the present invention. The zero-crossing discriminator (15) amplifies the amplitude-corrected signal output by the automatic gain control amplifier (13) and then differentiates it to extract the ToF timing.

[0055] The zero-crossing discriminator (15) shown in FIG. 6 is configured to include a differential amplifier and a comparator, and extracts the actual ToF timing by amplifying the amplitude-corrected signal output from the automatic gain control amplifier (13) once more through a non-inverting amplifier and then differentiating it.

[0056] The differentiation network has a trade-off relationship where, as the values ​​of the differentiation resistor and differentiation capacitor become smaller, the differentiation response time and work error are reduced, but the amplitude after differentiation is significantly reduced. If the amplitude of the APD signal is too small, the comparator may not detect the signal after differentiation, or even if it does detect it, work error may occur again due to the overdrive characteristics of the comparator. In particular, the automatic gain control amplifier (13), which outputs a constant amplitude of the input signal, has no choice but to lower the amplitude of the target output to keep the output amplitude constant over a wider range of input amplitudes due to the maximum gain limit. This is the reason why the signal output from the automatic gain control amplifier (13) must be amplified through an additional OP amp before reaching the differentiation network of the zero comparison detector. In other words, the pre-amplifier of the zero comparison discriminator completes the true meaning of 'amplifying weak signals and regulating strong signals' for the automatic gain control amplifier (13).

[0057] Also, for the same reason, the fixed amplification rate of the preamplifier and the value of each component of the differentiation network need to be tuned together with the amplification rates of the other amplifiers and the minimum amplitude of the target detectable APD output signal.

[0058] Ideally, the crossover point of the result of differentiating the signals output from the automatic gain control amplifier (13) for APD signals of various amplitudes should be at the 0V point, but in reality, it is shifted by tens to hundreds of μV due to the input offset voltage of the comparator. Therefore, the work error can be further reduced by inputting a bias voltage to the comparator so that the zero crossover point can be manually specified. This can reduce the work error to about 80% compared to using the comparator's threshold line as 0V.

[0059] The parameters to be determined for gain control are the maximum gain (Rg, Rf) of the variable gain amplifier (133), the target output peak voltage of the variable gain amplifier (133), and V required for calculating the appropriate gain of the gain calculator (137). A , V B , V subtractor , V D am.

[0060] Determining the gain control parameters begins with calculating the maximum gain of the automatic gain control amplifier (13), that is, the ratio of Rg to Rf of the variable gain amplifier. The maximum gain of the variable gain amplifier (133) is related to bandwidth, but is also significantly related to the detectable lowest and highest APD output peak signal amplitudes. This is because the signal amplitude range that can be input to the VCA824 is limited to the value obtained by dividing the output amplitude range (+-3.9 V) by the maximum gain.

[0061] The detection-possible lowest and highest APD output peak signal amplitudes are targeted to be 1 μA and 40 μA, respectively, and the predetermined transimpedance = 25k ohm, considering the noise immunity of the signal converted into voltage in the transimpedance amplifier (11), converts them to 25 mV and 1 V, respectively, and outputs them. When considering amplitude attenuation factors such as the delay line (131) in the path from the signal output from the transimpedance amplifier (11) to the input to the VCA, the maximum gain of the automatic gain control amplifier (13) can be expressed as <Equation 4>.

[0062]

[0063] However, if the APD output signal of the target detectable minimum amplitude is lowered when passing through the differentiation network and the amplitude is reduced, it falls short of the minimum required amplitude that enables stable detection in the comparator, it can be adjusted again along with other parameters such as the differentiation network, the transimpedance of the transimpedance amplifier (11), and the fixed amplification rate of other amplifiers.

[0064] Next, the amplitude at which the maximum gain of the VCA824 will be applied and the amplitude at which the gain will begin to be reduced must be set. In other words, the target output peak voltage of the variable gain amplifier (133) is set. Because the VCA824 has a limit of maximum gain, the gain cannot be increased indefinitely to obtain the same output amplitude for a very small APD signal amplitude range. To mitigate additional work errors caused by the overdrive characteristics of the comparator in the zero-crossing detector, the amplitude of the output signal of the automatic gain control amplifier (13) must be kept as constant as possible. To keep the output signal amplitude constant for a wider input signal amplitude range, the target output signal amplitude must inevitably be lowered. Therefore, coordination between the fixed amplification rate of the pre-amplifier of the zero-crossing detector and the maximum gain of the VCA824 is required.

[0065] Finally, the parameter V of the gain calculator (137). A , V B , V subtractor , V D The gain control of the VCA824 is set by inputting the voltage corresponding to the desired gain to the Vg pin, with the range from 0 V / V to the maximum gain set through the ratio of Rg and Rf corresponding to the range of -1 V to +1 V, respectively. Therefore, it can be expressed as in <Equation 5> and <Equation 6>.

[0066]

[0067]

[0068] At this time A vmax is the pre-set maximum gain of the VCA824, and A v is the appropriate gain value to be calculated. And for computational convenience, V g The gain of the automatic gain control amplifier (13) when = 0 V is A v0 Defined as.

[0069]

[0070] A v It can be calculated using <Mathematical Formula 9>.

[0071]

[0072]

[0073] At this time V op is the target, that is, the output signal peak voltage of the automatic gain control amplifier (13) that is expected to be regulated to this value, V op0 is V g = 0 V is the peak voltage of the output signal of the automatic gain control amplifier (13).

[0074] V op0 It is equal to <Mathematical Formula 10>.

[0075]

[0076] V ip V is the peak voltage of the signal input to the automatic gain control amplifier (13), and is a value received in advance from the peak detector (135). p0 is an offset voltage included in the peak voltage received from the peak detector (135), and is a constant added for circuit reasons.

[0077] Rearranging <Equation 7> through <Equation 10> yields V g It can be expressed as in <Mathematical Formula 11>.

[0078]

[0079] Meanwhile, the output gain of the gain calculator (137) is mentioned in <Equation 3>.

[0080] According to the circuit connection structure described in Fig. 1, V calculated from <Equation 11> g is equal to the gain in <Equation 3>, and also PEAKOUT is V ip It is the same as.

[0081]

[0082]

[0083] Therefore, the gain calculator parameters are determined as follows.

[0084]

[0085]

[0086]

[0087]

[0088] However, V subtractor It can be adjusted by several tens of mV to prevent the denominator from becoming zero.

[0089] Although the present invention has been described above with reference to embodiments with reference to the accompanying drawings, it is not limited thereto and should be interpreted to encompass various modifications that can be obviously derived from them by those skilled in the art. The claims are intended to encompass such modifications. Explanation of the symbols

[0090] 1: ToF LiDAR Walk Error Compensation Circuit 11 : Transimpedance amplifier 111: 1st OP amp 113: 1st feedback resistor 115: First feedback capacitor 13: Automatic gain control amplifier 131 : Delay line 133 : Variable gain amplifier 135 : Peak detector 1351: 2nd OP amp 1353: Second feedback capacitor 1355: Second feedback resistor 1357 : Diode 137 : Profit Calculator 15 : Zero-crossing discriminator 17 : AC coupling capacitor

Claims

Claim 1 A ToF LiDAR work error compensation circuit comprising: a transimpedance amplifier that receives a light signal reflected back from a photodiode (APD), amplifies the output current signal, converts it into a voltage signal, and outputs it; an automatic gain control amplifier that automatically gain-controls the voltage signal output by the transimpedance amplifier to amplify it into a signal of constant amplitude and outputs it; and a zero-crossing discriminator that amplifies the amplitude-corrected signal output by the automatic gain control amplifier, differentiates it, and extracts the ToF timing at the zero-crossing point of the differentiated result. Claim 2 A ToF lidar work error compensation circuit according to claim 1, further comprising an AC coupling capacitor disposed in front of a transimpedance amplifier to remove a DC signal component from the output signal of an avalanche photodiode. Claim 3 A ToF LiDAR work error compensation circuit according to claim 1, wherein the transimpedance amplifier comprises a first OP amplifier that amplifies an input current signal and then converts it into a voltage signal for output, a first feedback resistor connected to the output terminal and the inverting input terminal of the first OP amplifier to determine a current-voltage exchange ratio, and a first feedback capacitor connected in parallel with the first feedback resistor to stabilize the output waveform of the transimpedance amplifier. Claim 4 In claim 1, the automatic gain control amplifier comprises a peak detector that maintains and outputs peak information of the output signal of a transimpedance amplifier, a gain calculator that calculates a gain using the peak information output from the peak detector, a variable gain amplifier that variably amplifies the output signal of the transimpedance amplifier by applying the gain calculated by the gain calculator, and a delay line that delays the output signal of the transimpedance amplifier by the time generated by the gain calculator while calculating the gain and inputs it to the variable gain amplifier, thereby forming a ToF lidar work error compensation circuit. Claim 5 In claim 4, the peak detector comprises a second OP amplifier, a diode, a second feedback capacitor, and a second feedback resistor, wherein the output terminal of the transimpedance amplifier of the amplifier is connected to the non-inverting input terminal of the second OP amplifier, the input terminal of the diode is connected to the output terminal of the second OP amplifier, the output terminal of the diode is connected to the output terminal of the peak detector, the second feedback capacitor is connected to the output terminal of the diode and the inverting input terminal of the second OP amplifier to store peak information, and the second feedback resistor is connected in parallel with the second feedback capacitor, a ToF lidar work error compensation circuit. Claim 6 In claim 1, the zero-crossing discriminator further comprises: a differential amplifier; and a comparator that compares the input of the differential amplifier with a threshold line which is a bias voltage that can be manually specified; a ToF lidar work error compensation circuit.

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