Ultra wideband (UWB) link configuration systems

Abstract

Ultra-wideband (UWB) technology is a wireless technology that transmits large amounts of digital data over short distances with very low power across a wide spectrum. However, to support their deployment in wide-range applications, it is beneficial to provide solutions that utilize multiple directional antennas pointing in different directions to ensure spatial filtering of unwanted signals and increase signal strength; utilize dynamic configuration of multi-pulse beams for transmitting bits / symbols within packets to improve link service quality; utilize dynamic configuration of one or more frequency bands in which the transmitter operates; and utilize antenna subsystems that provide omnidirectional radiation patterns, enabling filtering and baffled functionality in a small size and low cost.

Description

Ultra-wideband link configuration method and system This invention pertains to ultra-wideband wireless communication systems, and more specifically to ultra-wideband transmitters and receivers configured to enhance the performance of ultra-wideband wireless links, as well as antennas for such communication systems. Ultra-wideband (UWB) technology is a wireless technology that transmits large amounts of digital data as modulated-coded pulses over short distances over a very wide spectrum at very low power. This type of pulse-based transmission is an alternative to transmission using sine waves, which are then turned on or off to present a digital state, as is used in current wireless communication standards and systems such as IEEE 802.11 (Wi-Fi), IEEE 802.15 Wireless Personal Area Network (PAN), IEEE 802.16 (WiMAX), Universal Mobile Telecommunications System (UMTS), Global System for Mobile Communications (GSM), General Packet Radio Service (GPRS), and those networks accessing the Industrial, Scientific and Medical (ISM) bands and International Mobile Telecommunications 2000 (IMT-2000). UWB systems are well-suited for short-range applications in a variety of environments, such as those depicted in Figure 1, including peripheral and device interconnection exemplified by the first residential environment 110, sensor networks exemplified by the second residential environment 120, control and communication exemplified by the industrial environment 130, medical systems exemplified by medical imaging 150, and PANs exemplified by Personal Area Networks (PANs) 140. Due to the low emission levels permitted by regulatory authorities, such UWB systems tend to be suited for short-range indoor applications; however, it is clear that a variety of other applications could be considered in the absence of such regulatory restrictions, addressing military and civilian needs such as communication between individuals, electronic devices, control centers, and electronic systems. Therefore, it is beneficial for UWB transmitters, UWB receivers, and UWB transceivers to utilize multiple directional antennas oriented in different directions to ensure spatial filtering of unwanted signals and increase signal strength. Further beneficial for multiple directional antennas is the integration of baluns (also known as baluns). A further benefit for UWB transmitters, UWB receivers, and UWB transceivers is the ability to utilize dynamic configuration of multi-pulse beams for transmitting bits / symbols within packets. Such dynamic configuration is used to enhance link service quality. A further advantage for UWB transmitters, UWB receivers, and UWB transceivers is the ability to dynamically configure one or more frequency bands operated by the transmitter. Further beneficial for UWB transmitters, UWB receivers, and UWB transceivers is the use of an antenna subsystem that provides an omnidirectional radiation pattern, wherein the antenna subsystem has a printed circuit board implementation that provides filtering and provides small size and low cost balun functionality. Other aspects and features of the invention will become apparent to those skilled in the art when the following description of specific embodiments of the invention is read in conjunction with the accompanying drawings. The purpose of this invention is to alleviate the limitations of the prior art in relation to ultra-wideband wireless communication systems, and more specifically, in relation to the configuration of ultra-wideband transmitters and receivers to enhance the performance of ultra-wideband wireless links and the antennas used in such communication systems. According to an embodiment of the present invention, an antenna is provided, comprising: a substrate; a first antenna structure formed on the substrate, operating within a predetermined frequency range, including a first pattern on a first surface of the substrate and a first ground plane formed on a second surface of the substrate opposite to the first surface of the substrate; a second antenna structure formed on the substrate, operating within the predetermined frequency range, including a second pattern on the first surface of the substrate and a second ground plane formed on the second surface of the substrate; a first feed structure electrically coupled to the first antenna structure and a first input feed portion; and a second feed structure electrically coupled to the second antenna structure and the second input feed portion, wherein the first antenna structure and the second antenna structure are arranged at 90° to each other with respect to the axis of the substrate; and the first input feed portion and the second input feed portion are aligned and arranged on opposite sides parallel to the axis of the substrate. According to an embodiment of the present invention, a transmitter for pulse radio is provided, comprising: a radio frequency (RF) signal generator for transmission, the RF signal generator receiving a data signal to be transmitted and a clock signal characterized by a clock frequency and coupled to an RF antenna and a control circuit; the control circuit controlling the generation of transmitted data such that each transmitted bit includes multiple data generated by the RF signal generator. 1 pulse, of which Each pulse in a pulse is in multiple A predetermined frequency, having a predetermined amplitude, and having a predetermined pulse length; wherein, and ; and It is an integer; the pulse repetition rate of the RF signal generator is determined based on the clock frequency; integer. It depends on the duration of the data signal bits and the pulse repetition rate of the RF signal generator; multiple Each pulse is emitted during the duration of a bit in the data signal, and includes pulses in multiple... Pulses at various frequencies; and multiple Each pulse is within the predetermined frequency band. According to an embodiment of the present invention, an antenna subsystem is provided, comprising: an ultra-wideband radio circuit (UWB) providing a balanced output signal over a predetermined frequency range; a monopole antenna having a single-ended input for providing an omnidirectional radiation pattern over the predetermined frequency range; and intermediate circuitry including a balun and a filter for converting the balanced output signal of the UWB radio circuit into the single-ended input of the monopole antenna. According to an embodiment of the present invention, an antenna subsystem is provided, comprising: an ultra-wideband radio circuit (UWB) providing a balanced output signal over a predetermined frequency range; a monopole antenna having a differential input for providing an omnidirectional radiation pattern over the predetermined frequency range; a differential low-pass filter having a differential input and a differential output, wherein the differential output of the differential low-pass filter is coupled to the differential input of the monopole antenna; and an intermediate circuit including a differential feed line for coupling the UWB radio circuit to the differential low-pass filter, the intermediate circuit including a notch filter for attenuating the second harmonic of the signal applied to the intermediate circuit over the predetermined frequency range. According to an embodiment of the present invention, an antenna subsystem is provided, comprising: an ultra-wideband radio circuit (UWB) providing a balanced output signal over a predetermined frequency range; a monopole antenna providing an omnidirectional radiation pattern over the predetermined frequency range; a differential low-pass filter having differential inputs and differential outputs, wherein a first output of the differential output of the differential low-pass filter is coupled to a first side of the monopole antenna, and a second output of the differential output of the differential low-pass filter is coupled to a second far side of the monopole antenna; and an intermediate circuit including differential feed lines for coupling the UWB radio circuit to the differential low-pass filter, the intermediate circuit including a notch filter for attenuating the second harmonic of the signal applied to the intermediate circuit over the predetermined frequency range. Other aspects and features of the invention will become apparent to those skilled in the art from the following description of specific embodiments of the invention taken in conjunction with the accompanying drawings. This invention pertains to ultra-wideband wireless communication systems, and more specifically to ultra-wideband transmitters and receivers configured to enhance the performance of ultra-wideband wireless links, as well as antennas for such communication systems. The following description provides exemplary embodiments only and is not intended to limit the scope, applicability, or configuration of this disclosure. Rather, the following description of exemplary embodiments will provide those skilled in the art with enabling descriptions for implementing the exemplary embodiments. It should be understood that various changes can be made to the function and arrangement of the elements without departing from the spirit and scope set forth in the appended claims. 0. Pulse Radio Ultra Wideband System As mentioned above, UWB offers many potential advantages, such as high data rates, low-cost implementation, low transmit power, ranging, multipath immunity, and low interference. The U.S. Federal Communications Commission (FCC) regulations on UWB reserve unlicensed frequency bands between 3.1 GHz and 10.6 GHz for indoor UWB wireless communication systems, where low-modulated transmit power allows such UWB systems to coexist with other licensed and unlicensed narrowband systems. Therefore, limited spectrum resources can be utilized more efficiently. On the other hand, UWB systems with their ultra-wide bandwidth offer significantly greater capacity for short-range applications than current narrowband systems. Two possible technologies for implementing UWB communication are pulse radio (IR) UWB and multi-carrier or multi-band (MB) UWB. IR-UWB utilizes the transmission of ultra-short (nanosecond-level) pulses, but in some cases, more than one pulse represents one symbol to increase processing gain. Conversely, MB-UWB systems use orthogonal frequency division multiplexing (OFDM) technology to transmit information on each sub-band within the sub-band. While OFDM possesses several desirable characteristics, including high spectral efficiency and robustness against RF and multipath interference, it also suffers from several drawbacks. These include the need for up-conversion and down-conversion, the requirement for mixers and their associated high power consumption, and extreme sensitivity to inaccuracies in frequency, clock, and phase. Similarly, nonlinear amplification disrupts the orthogonality of OFDM. Therefore, MB-UWB is not suitable for low-power and low-cost applications. In contrast, IR-UWB offers several advantages, including unlicensed use of several gigahertz of spectrum, providing great flexibility in spectrum usage, and self-tuning transceiver designs that can optimize system performance based on data rate, operating range, available power, desired quality of service, and user preferences. Furthermore, multi-Gb / s data rate transmission over very short distances is possible, and due to the ultrashort pulses within IR-UWB, it is highly robust to multipath interference, and in some implementations, more multipath components can be resolved at the receiver, resulting in higher performance. Additionally, ultrashort pulses support sub-centimeter ranging, while the lack of up-conversion and down-conversion allows for reduced implementation costs and lower-power transceiver implementations. Advantageously, the ultrashort pulses and low-power transmission make IR-UWB communications difficult to eavesdrop on. As described below with reference to Figures 2 and 3 regarding embodiments of the invention, the IR-UWB transmitter utilizes a pulse generator followed by an on-demand oscillator to upconvert pulses from the generated pulses, thus avoiding the need for a separate mixer. Both the pulse generator and the on-demand oscillator can be digitally tuned, provided they can be implemented in standard CMOS logic, to offer control over the pulse bandwidth and center frequency. Furthermore, by utilizing a digitally controlled ring oscillator for the on-demand oscillator, the IR-UWB transmitter is designed to allow very fast frequency adjustments on the order of pulse repetition rate (PRR). Advantageously, this technique offers the same advantages as MB-OFDM in terms of spectral configurability, achieved by sequentially changing the transmit spectrum using a frequency hopping scheme, while retaining the advantages of IR-UWB. Moreover, by providing a high duty cycle with fast power-up time combined with on-off shift keying (OOK) modulation, the IR-UWB according to embodiments of the invention allows for significant power reduction by utilizing the low duty cycle of UWB symbols and the fact that only half of the symbols require transmit energy. In addition to defining the operating frequency range of UWB systems, different regulatory agencies specify and implement specific power spectral density (PSD) masks for UWB communications. The PSD mask that may be used for embodiments of this invention is the FCC mask, for which Table 1 below summarizes masking data in the 3100MHz-10600MHz (3.1GHz-10.6GHz) range. Table 1: Indoor and outdoor FCC shielding in different frequency bands. Therefore, it is clear that the upper limit of -41.3 dB / MHz in the 3.1 GHz–10.6 GHz frequency range is the same as the limit imposed on unintentional radiation at a given frequency to avoid interfering with other radios. Essentially, for a given frequency, UWB radios operate at permissible noise levels, which... Transmit energy per pulse, maximum spectral power ,bandwidth bit rate and the number of pulses per unit The relationship between them is shown in formula (1): (1). IEEE has published several standards for the physical layer (PHY) of UWB radio in Personal Area Networks (IEEE 802.15.4a-2007), Body Area Networks (IEEE 802.15.4a-2007), and Radio Frequency Identification (IEEE 802.15.4f-2012). These standards primarily use relatively large pulses, resulting in a relatively narrow bandwidth, which is up-converted to a specific center frequency to fill a predetermined channel. Data is encoded using Pulse Position Modulation (PPM), and Dual Phase Shift Keying (BPSK) is used to encode redundant data. Each bit consists of one or more pulses phase-scrambled according to the target data rate. These standards allow considerable flexibility in terms of channel availability and data rate. The standard also defines the preamble, header, and ranging protocol for data packets. These IEEE standards, designed with multiple users in mind and using different channels to transmit data, impose strict constraints on pulse bandwidth and limit the transmitted energy. Existing technology for non-standard transmitters attempts to better utilize the available spectrum by using narrow pulses, thus achieving a wider bandwidth and increasing the maximum transmit energy according to formula (1). Therefore, these transmitters are non-standard and designed for different data rates, frequencies, pulse widths, etc. Furthermore, they use various coding schemes, most notably PPM, OOK, or BPSK. In the work described below, the inventors have developed improvements to UWB systems, UWB transmitters, and energy-based UWB receivers that generate and adapt to a variety of IR-UWB pulse and bit coding schemes, thereby supporting communication from both IEEE-compliant and non-standard IR-UWB transmitters. These improvements are made for UWB transmitters, UWB receivers, UWB transceivers, and UWB systems, such as those described and depicted by the inventors in the following documents: WO / 2019 / 000,075 UWB "High-Efficiency Ultra-Wideband Pulse Radio System and Method" (PCT / CA2018 / 000,135 filed June 29, 2018); WO 2016 / 191,851 "System and Method of Spectrum-Efficient and High-Efficiency Ultra-Wideband Pulse Radio with Scalable Data Rate" (PCT / CA2016 / 000,161 filed May 31, 2016); and WO / 2015 / 103,692 "System and Method Related to Ultra-Wideband Broadcasting Including Dynamic Frequency and Bandwidth Jumps" (PCT / CA2015 / 000,007 filed January 7, 2015). 1. IR-UWB Transmitter circuit Referring to Figure 2, an exemplary architecture of an IR-UWB transmitter 200 according to an embodiment of the present invention is schematically depicted, consisting of five main blocks plus an antenna. First, when a data signal from gate 210 is high based on a control signal provided to gate 210, pulse generator 230 generates programmable pulses at clock intervals. Then, the pulses from pulse generator 230 are up-converted by a programmable multi-ring digitally controlled ring oscillator (DCRO) 240. The output from DCRO 240 is then coupled to a variable gain amplifier (VGA) 250 to compensate for any frequency dependence of the pulse amplitude. Finally, driver 260 feeds antenna 270, thereby overcoming typical package parasitic effects, such as those resulting from encapsulating the transceiver in a quad flat no-lead (QFN) package. To further reduce the power consumption of the IR-UWB transmitter (IR-UWB-Tx) 200 according to an embodiment of the present invention, power loop controller 220 dynamically turns these functional blocks on or off when the data signal is low. Referring now to FIG. 3A, a block diagram 300 of an exemplary IR-UWB transmitter supporting biphasic phase scrambling according to an embodiment of the present invention is schematically depicted. Compared to the non-biphasic phase-shifted IR-UWB transmitter 200 for IR-UWB according to an embodiment of the present invention in FIG. 2, the biphasic phase-shifted IR-UWB (BPS-IR-UWB) transmitter comprises six main blocks instead of five main blocks plus an antenna. First, when the data signal from gate 310 is high based on a control signal provided to gate 310, pulse generator 330 generates programmable pulses at clock intervals. Then, a programmable multi-ring digitally controlled ring oscillator (DCRO) 340 up-converts the pulses from pulse generator 330. The output from DCRO 340 is then coupled to a dual output amplifier (VGA) 350 to compensate for any frequency dependence of the pulse amplitude, and also generates a dual-phase-shifted output signal coupled to a switch 360, which, under the action of a switch control signal "S" applied to switch 360, selects to couple one of the two signals to an output power amplifier (driver) 380. Note that a similar phase selection scheme can be achieved by influencing the startup conditions of DCRO 340 to provide two phases. This would eliminate the need for switch 360 at the cost of adding a startup condition control signal to DCRO 340. Output power amplifier 380 feeds antenna 370, thereby overcoming typical packaging parasitic effects, such as those resulting from encapsulating the transceiver within a quad flat no-lead (QFN) package. To reduce power consumption of the BPS-IR-UWB transmitter represented by block diagram 300 according to an embodiment of the invention, power loop controller 320 dynamically turns these functional blocks on or off when the data signal "PC" is low. Therefore, the BPS-IR-UWB transmitter according to an embodiment of the invention transmits pulses with or without phase shift based on a control signal "S" applied to switch 360. If this control signal is now fed from a scrambled data generator or a pseudo-scrambled data generator, the resulting pulses coupled to the antenna of the BPS-IR-UWB transmitter will be pseudo-randomly or randomly phase-shifted. Referring now to FIG. 3B, a block diagram of an exemplary IR-UWB transmitter 3000 according to an embodiment of the present invention is schematically depicted. As depicted, a pulse mode block 3010 maintains a configuration for the pulses representing the current symbol. Multiple phases are generated by a delay-locked loop (DLL) 3030 according to the symbol rate clock (i.e., 20 MHz). The rising edge of each clock phase indicates the start of a pulse in the symbol pulse beam. A multiplexer 3020 is triggered by the edge of the clock phase and selects the configuration of the current pulse from the pulse mode block 3010. A pulse generator (Pulser) 3050 generates a pulse having a pulse width set by the multiplexer 3020 and enables a digitally controlled oscillator (DCO) 3040 and a power amplifier (PA) 3060. When enabled, the DCO 3040 generates a Gaussian-shaped pulse with a frequency set by the multiplexer 3020, which is then amplified by the PA 3060 and radiated by the antenna 3070. Therefore, the pulse mode block 3010 establishes pulses for symbols or symbol sequences. In this way, the pulse mode block 3010 is updated to adjust the pulse sequence for each symbol, and thus, the pulse mode block 3010 can be dynamically updated based on one or more factors, including but not limited to network environment information, predetermined sequence, date, time, geographical location, signal-to-noise ratio (SNR) of the received signal, and regulatory masking. Referring to FIG3C, a multi-pulse symbol UWB protocol according to an embodiment of the present invention is schematically depicted. Referring to the first image 3100A, bits 3160 including a series of sub-pulses 3160A to 3160C are depicted, each sub-pulse being at a frequency Therefore, the multipulse spectrum 3180 of the symbol (bit 3160) is depicted in the second image 3100B, which is conceptually obtained by adding the individual pulse spectra of sub-pulses 3160A to 3160C (phase scrambling is omitted for clarity). This increases bandwidth while increasing the total symbol duration compared to existing single-pulse methods, while maintaining maximum power below the UWB mask 3120. This allows for maximizing symbol energy while relaxing the timing requirements and synchronization levels required by the receiver. Any number of pulses with different parameter sets can be included in the bundle to tailor the pulse spectrum to meet given requirements. Referring now to FIG3D, the measurement pulse shapes of a UWB transmitter according to an embodiment of the present invention (such as those described and depicted in FIG2 or FIG3A, without biphase scrambling according to WO / 2015 / 103,692) are depicted at three different frequency settings. The first to third traces 3200A to 3200C represent single pulses at 2.4 GHz, 3 GHz, and 4 GHz, respectively. Referring to Figure 3E, a pulse beam representing the transmitted bits is depicted in the first trace 3300A, as described in WO / 2015 / 103,692, while the second trace 3300B depicts the resulting power spectral density (PSD). Clearly, this allows the PSD of the final signal to be managed through the parameters of each pulse within the frequency-hopping sequence. The depicted pulse sequence includes four pulses at 2.4 GHz, two pulses at 3.5 GHz, and three pulses at 4.0 GHz. Across the entire frequency band, the resulting PSD fills the spectrum with approximately -58 dBm. Referring now to FIG3F, the power spectrum and pulse train of a pulse beam, as illustrated in FIG3A and according to WO 2016 / 191,851, are depicted. This pulse beam supports operation in a frequency range of approximately 3 GHz to approximately 7 GHz. First images 3400A and second images 3400B represent the power spectrum and pulse sequence, respectively, where no random frequency or phase scrambling occurs during generation and transmission. Third images 3400C and fourth images 3400D depict the results of random frequency and random phase scrambling of the pulses, where data established by a scrambled data generator sets the phase for each pulse. Therefore, it is evident that, when comparing the first image 3400A and the third image 3400C, the introduction of random frequency and random phase shift significantly reduces the spectral lines within the transmission spectrum of the UWB transmitter according to an embodiment of the present invention. 2. IR-UWB receiver Referring to Figure 4, the architecture of an IR-UWB receiver 400 according to an embodiment of the present invention is schematically depicted. Thus, a signal from an IR-UWB transmitter is received via antenna 410 and coupled to a low-noise amplifier (LNA) 420, and then coupled to a first amplifier 430, where the resulting signal is squared by a squaring circuit 440 to assess the amount of energy in the signal. The output of the squaring circuit 440 is then amplified by a second amplifier 450, integrated by an integrator circuit 460, and evaluated by a flash memory ADC 470 to generate an output signal. A power loop controller 480 is also depicted, which dynamically powers on and off the LNA 420, the first amplifier 430 and the second amplifier 450, the squaring circuit 440, and the flash memory ADC 470 in a manner similar to the power loop controller 220 of the IR-UWB transmitter 200 in Figure 2, respectively, to further reduce power consumption according to circuit requirements. Referring to Figure 5, a schematic diagram of a receiver 500 according to an embodiment of the present invention is depicted. The RF signal from antenna 510 is initially amplified by a low-noise amplifier (LNA) 520 and then passed to a two-stage RF amplifier (AMP1) 530. A first square mixer (MIX1) 540 multiplies the signal by itself to convert it to an intermediate frequency (IF). A three-stage variable gain amplifier (VGA) 550 further amplifies the signal and performs a bandpass filter function. The output of VGA 550 is then coupled to a second square mixer (MIX2) 560, which downconverts the signal to a baseband frequency. Processing circuitry 580 includes a pair of parallel integrators (INT1 and INT2) that sum the signal energy, which is then digitized by analog-to-digital converters (ADC1 and ADC2) and sent to a digital document processor (not depicted for clarity). 3. IR-UWB receiver As described in WO / 2019 / 000,075 and WO 2016 / 191,851, the inventors established design parameters for a startup time in the millisecond range from sleep mode and a startup time in the microsecond range from idle mode by creating a custom integrated DC / DC converter and duty cycle transceiver circuit. This custom integrated DC / DC converter and duty cycle transceiver circuit can achieve low ( ) and medium data rate ( To achieve fast circuit startup / shutdown for optimal power consumption. To maintain good energy efficiency, the components of the total UWB transceiver (such as the transceiver 600 depicted in FIG. 6 according to an embodiment of the invention) have been designed for low quiescent sleep current and fast start / sleep time. Referring to FIG. 6, the battery ( (Not depicted for clarity) The low-frequency crystal oscillator 615, sleep counter 620, and bandgap reference 610 are powered. Although the bandgap reference 610 may have a duty cycle in other embodiments of the invention without changing the scope of the claimed invention, the low-frequency crystal oscillator, sleep counter, and bandgap reference are all generally always operational. Their power consumption limits the minimum power consumption of the system to the sub-microwatt level. When the system is not in sleep mode, the integrated buck DC-DC converter 605 is battery powered, and this provides the power supply voltage to the rest of the system with high conversion efficiency. The startup time of the DC-DC converter 605 is on the order of several symbol cycles to minimize wasted energy. Between sleep cycles, the PLL 655 starts up to provide the base clock for the system. The receiver 625 and DLL 660 have dedicated power-down controllers and are only activated during frame transmission / reception. In addition, the transmitter also performs power cycling through its all-digital architecture, which is not described as having a separate controller. Due to the low base clock (e.g., 20 MHz), the power consumption of the digital synthesis module is low. In principle, the power cyclic transceiver (PLL) achieves constant energy efficiency by linearly scaling power consumption with the data rate. With a fixed frame size, multiple data rates are obtained by adjusting the length of the sleep cycle, where the maximum achievable data rate is determined by the symbol rate within the frame itself. To conserve energy efficiency, power consumption during sleep must be lower than the average power consumption. For high data rates, the PLL does not need to be powered down when its power consumption does not significantly reduce overall efficiency. For low data rates, the entire system, except for the bandgap reference, crystal oscillator, and sleep counter, can be shut down during sleep mode. In this case, the millisecond-range startup time of the PLL may be negligible compared to the sleep cycle, and overall efficiency will not decrease significantly. As depicted, the UWB transceiver 600 also includes a receive / transmit switch 690 coupled to an antenna to selectively couple the transmitter 6000 or the receiver 625 to the antenna during transmission and reception, respectively. The UWB transceiver 600 also includes spectrum configuration circuitry 665 (equivalent to pulse mode 3010 in transmitter 3000 in FIG. 3B), PHY processing circuitry 650, link controller 645, buffer and interface circuitry 640, analog-to-digital converter (ADC) 630, multiplexer (MUX) 670, pulse generator 675, mixer 680, power amplifier (PA) 685, and PHY formatting circuitry 635. The UWB transceiver 600 communicates with the user terminal 695 via the link controller 645. Thus, the link controller 645 can communicate with the user terminal 695, for example, using a wired protocol (e.g., Serial Peripheral Interface (SPI)). 4. For UWB Compact Dipole Antenna Diversity of the System 4A. Overview Indoor wireless communications are frequently affected by various propagation phenomena, which, if not properly addressed, can degrade signal quality. In indoor environments, multipath interference is a significant source of fading that severely degrades link quality. When radio waves propagate through the air, they are received at the receiver via different paths. Based on their propagation paths, obstacles, and reflecting media, these received signals will have varying amplitudes and time delays (phase differences). When combined at the receiver, these signals from different paths may partially or, in some cases, completely cancel each other out, resulting in a significant deterioration in signal quality. This fading effect is exacerbated in dynamic environments with moving objects in the room. Therefore, addressing this issue at both the system level and the radio and antenna levels is crucial. Antenna diversity techniques are frequently used to mitigate this problem. In this method, more than one antenna is used to transmit and receive radio signals. Typically, to achieve the maximum possible diversity, the antennas are orthogonally oriented to provide polarization diversity. Furthermore, placing the antennas at approximately a quarter-wavelength distance between adjacent antennas ensures spatial diversity, thereby improving system efficiency. In this configuration, signals are received from two antennas and compared with each other, and the stronger signal is selected. This technique has been shown to effectively mitigate multipath fading in both dynamic and static environments. Numerous architectures for ultra-wideband (UWB) antenna diversity have been proposed in the literature. However, most of these architectures are based on monopole antennas with relatively omnidirectional radiation patterns. Using omnidirectional antennas for diversity reduces the efficiency of this technique because the radiation patterns of the antennas will overlap, and signals will still arrive at both antennas via various paths. Therefore, it is preferable to use multiple directional antennas oriented in different directions to ensure spatial filtering of unwanted signals. Therefore, the inventors have developed a novel compact antenna diversity architecture for use with UWB transmitters, UWB receivers, and UWB transceivers utilizing techniques according to embodiments of the invention (those described and depicted in WO / 2019 / 000,075, WO2016 / 191,851, and WO / 2015 / 103,692) and other aspects of these UWB devices not described herein or elsewhere. The antenna system of the present invention consists of two UWB planar microstrip dipole antennas that provide spatial diversity and polarization diversity by having directional radiation patterns. The key characteristics of this architecture are its compact structure, wideband impedance matching, stable gain over the bandwidth, and perfect envelope correlation between the antennas, making it a strong candidate for high data rate applications. 4B. Antenna structure The antenna structure is depicted in Figure 7. It consists of two broadband dipole antennas with integrated balusters. The two dipoles are placed at a 90° angle to each other to ensure maximum polarization diversity. As can be observed in Figure 7, the overall structure is compact, low-profile, and low-cost, making it suitable for a variety of high data rate indoor applications. 4C. Simulation results The antennas were simulated using the full-wave finite element method (FEM) in ANSYS HFSS, a 3D electromagnetic field simulator for RF and wireless designs. The |S11| (dB) results are depicted in Figure 8A, showing perfect matching between the two antennas within the 3.4 GHz to 5.8 GHz band (covering the desired target frequency band of UWB radio waves developed by Spark Microsystems in Montreal, Canada). Figure 8B clearly shows that the antennas are perfectly isolated, with S21 values ​​below -25 dB between 3.5 GHz and 5.5 GHz. This is primarily due to the directional nature of the antennas radiating in two different directions. Good isolation between the antennas ensures low envelope correlation and thus higher diversity gain. Figures 9A to 9C show the antenna radiation patterns for different frequencies and three different planes: φ=0°, φ=90°, and θ=90°. As can be observed from these figures, the radiation patterns are completely directional and illuminate only one side of the UWB radio at a time. This allows for attenuation of multipath signals received from the other direction. When the antenna is switched, the other side of the radio is activated. Therefore, complete spatial diversity is achieved while the combination of the two antennas covers the entire space surrounding the radio wave module. In addition to the features mentioned above, the antenna has a uniform radiation pattern across various frequencies, which ensures stable link quality throughout the desired frequency band. The peak realized gain of the antenna is depicted in Figure 10A. Notably, the peak realized gain remains around 4 dB across the 3.5 to 5.5 GHz range. Therefore, by using this antenna, radio waves will be able to utilize a single output power distribution across the bandwidth. Furthermore, the antenna will attenuate signals outside the desired bandwidth. Due to the antenna's high realized gain, the coverage of radio waves is enhanced in both indoor and outdoor environments. The antenna's radiation efficiency is also depicted in Figure 10B, exhibiting a radiation efficiency exceeding 90% over the desired frequency range. Antenna diversity performance can be analyzed using a metric called envelope correlation coefficient (ECC). The measurement shows how the antennas correlate. Since the antennas have completely different polarizations and are well isolated, a very low ECC is expected for this configuration. The ECC can be calculated using the S-parameters of the two antennas using formula (1): (1). The resulting ECC is depicted in Figure 11. As expected, a very low ECC value was achieved over the operating bandwidth, resulting in very good diversity efficiency for the antenna. 5. With parasitic elements UWB Loop antenna ( Loop Antenna ) 5A. Overview The antenna of this invention uses a differentially driven loop element. As depicted in FIG12, the loop element 1210 is coupled to a differential feed (or differential feed section) 1250. A first parasitic (non-driven) element 1220 and a second parasitic element 1230 are arranged on both sides of the loop element, and a third parasitic element 1240 is located in the center of the loop element. The first parasitic element 1220 and the second parasitic element 1230 are coupled to the underlying ground plane and are used to tune the low-frequency cutoff. The central third parasitic element 1240 is used to tune the higher cutoff frequency. Furthermore, the first parasitic element 1220 to the third parasitic element 1240 respectively allow for a reduction in the overall size of the antenna. 5B. performance Figure 13A depicts the |S11dd| (dB) results of a UWB loop antenna with parasitic elements, measured and simulated in the 2 GHz to 6 GHz frequency range. Referring to Figure 13B, the effect of the central resonator on the nominal |S11dd| is depicted, while Figure 13C depicts the effect of the side resonators on the nominal |S11dd|. Therefore, Figure 13D depicts the performance of antennas with and without all resonators. The antenna is omnidirectional and has a low gain that is stable across the frequency range. The gain mode can be tuned by changing the position and length of the resonator. Figure 14A depicts the antenna gain versus frequency. Referring to Figure 14B, it is clear that the antenna matching is minimally affected by the width of the ground plane, which depicts the |S11dd| (dB) results for different ground plane sizes from 35 mm to 200 mm in the 2 GHz to 6 GHz frequency range. Figures 13A through 13D and Figures 14A through 14B depict |S11dd|, which is a mixed-mode S-parameter related to the power reflected in the differential mode. Similar results were obtained when a flesh-like object was brought near the antenna, indicating that the antenna can be used near body parts without detuning. These results are depicted in Figure 14C. Referring to Figures 15A and 15B, two antenna designs employing the same design methodology are depicted. In both cases, the substrate width is 40 mm (1.6 inches). The antenna design in Figure 15A is intended to operate in the frequency range of 6 GHz to 9 GHz and has dimensions of 10 mm x 16 mm (~0.4 inches x 0.65 inches), while the antenna design in Figure 15B is intended to operate in the frequency range of 3.5 GHz to 5.5 GHz and has dimensions of 19 mm x 34 mm (~0.75 inches x 1.4 inches). Schematic diagrams of the two antennas are depicted in Figures 15C and 15D, respectively. 5C. Filtering The inventors also modified the loop antenna to add band-stop filtering and allow low-frequency cutoff tuning and high-frequency cutoff tuning via parasitic elements (such as the first parasitic element 1220 to the third parasitic element 1240 in Figure 12, respectively). Therefore, the inventors employed a stub-line filter as illustrated in Figure 16A and a slot-line filter as illustrated in Figure 16B. As clearly seen in Figure 16A, the stub-line filter is implemented as a notch within the central third parasitic element, while the slot-line filter is implemented within the edge portion of the ground plane below the differential feed of the loop antenna. Thus, the band-stop filter increases the antenna height by 1 mm (~0.05 inches), while the common-mode rejection slot line remains unchanged in size. As clearly seen in Figure 16C, the band-stop filter reduces efficiency near the 2.5 GHz and 5.75 GHz ISM bands. The slotted line filter reflects common-mode signals, resulting in the performance depicted in Figure 16D. 6. Broadband with common-mode absorption and harmonic suppression RAT-RACE Belle 6A. Overview As described above in Section 5, a loop antenna forming part of a diversity antenna according to an embodiment of the invention has a single feed (or a single-feed section). Therefore, radio waves using such an antenna may require balunning (balanced to unbalanced) to couple the differential circuitry of the radio wave electronics to the single-ended feed of the loop antenna. For example, the differential output of an RF signal generator may be coupled to a single-ended transmitting antenna, or a single-ended receiving antenna may be coupled to the differential input of the front end of an RF receiver. A balun is a reversible device, meaning it can be used in both directions, and therefore in embodiments of the invention, radio waves can be simultaneously transmitted and received using a single diversity antenna, or in other embodiments of the invention, different antennas can be used to implement the transmitting and receiving functions, and in still other embodiments of the invention, a single antenna can be used in conjunction with either a receive-only radio wave or a transmit-only radio wave. If the balanced port of a balun is excited by two similar signals with the same phase, they are either suppressed or absorbed. This is called the common-mode rejection function of the balun. In some cases, it is desirable to absorb common-mode signals and guide them to the ground plane. Several methods exist for designing baluns at microwave frequencies. In most of these methods, the balun is typically designed with three ports and usually reflects common-mode signals. However, in cases where it is desirable to absorb common-mode signals, a four-port structure is required, from which both differential and common-mode signals can be extracted. In this section, the inventors describe an innovative, compact, folded planar baffle design based on a microstrip 180° rat-race hybrid junction, exhibiting low loss and wideband performance. The 180° hybrid junction is a four-port network that can be designed in various configurations. The rat-race hybrid junction can be designed in a planar form using microstrip or coplanar transmission lines, offering the benefits of a low profile and compact design. 6B. Folded balusters and simulation results The key characteristic of the rat-race hybrid connector is that if it is energized at ports 2 and 3, the sum of the signals will be located at port 1, while port 4 will output their difference. These ports are depicted in Figure 17A, which shows a plan view of the rat-race hybrid baffle. Therefore, this feature can be used to design baffles from the rat-race hybrid connector. If ports 2 and 3 are energized with differential signals, an unbalanced signal will appear at port 4, and the load connected at port 1 can absorb the common-mode signal, as depicted in Figure 17A, with a resistor R_absorb coupled between port 1 and ground. If the hybrid connector is perfectly matched, there will be no reflection of the common-mode signal, and signal integrity issues can be avoided. Figure 17B depicts a 3D drawing of the rat-race ballast structure, showing that it employs a rectangular folded structure, thus reducing the overall size of the structure. The dimensions shown in Figure 17B are 22mm x 12.5mm (0.85 inches x 0.5 inches), representing a folded rat-race ballast designed for operation from 3.6 GHz to 5.1 GHz. Figure 18A depicts the simulated differential and common-mode reflection coefficients, while Figure 18B depicts the insertion loss of the rat-race ballast. As can be clearly seen from Figure 18A, using this structure, a good differential S11 with a range below -10 dB from 3.1 GHz to 5.6 GHz is achieved, and common-mode matching with a range below -10 dB from 3.7 GHz and above is also achieved. This means that common-mode signal reflection is avoided, and as mentioned earlier, common-mode reflection can be absorbed from port 1. Furthermore, as can be seen from Figure 18B, this structure has extremely low insertion loss, with a maximum of -1.5 dB at 5.5 GHz. This ensures system efficiency when the balun is integrated with other circuit elements such as filters and antennas. Two other important characteristics of the bainoid to be analyzed are amplitude balance and phase balance, with analog values ​​depicted in Figures 19A and 19B, respectively. Amplitude balance represents the difference in amplitude of the signal received at port 4 when excited from ports 2 and 3. Ideally, this value should be 0 dB across the entire band of interest, but typically a maximum difference of about 1.5 dB is acceptable in most applications. This value is emphasized in the proposed designs from 3.6 GHz to 5.3 GHz, and is considered to be for broadband performance. At the center frequency, the phase difference at the output port should be exactly 180°, and the phase difference should be close to this value for the remainder of the desired bandwidth. This balun provides a phase difference of 180° at 4.3 GHz, while at the lower and higher frequencies of interest (3.5 GHz and 5.5 GHz, respectively), the phase differences are 196° and 150°. Therefore, the phase variation over the desired bandwidth is acceptable. Advantageously, the progressive folded rat-race bangle according to embodiments of the invention exhibits good performance over a wide frequency range of interest (from 3.5 GHz to 5.5 GHz) while providing significant advantages over currently commercially available bangles, namely, low cost, printability on PCBs, low loss, and immunity to common-mode reflections. 7. Multipath of multi-frequency pulse band / Cross-fading mitigation As described above with respect to sections 0 through 3, the inventors have established advanced radio waves utilizing multi-frequency pulse beams for each transmit bit, thereby allowing the PSD of each bit to be constructed through the cumulative spectral characteristics of multiple pulses within each multi-pulse beam. These concepts are also described in WO / 2019 / 000,075, WO2016 / 191,851 and WO / 2015 / 103,692. Referring to Figure 20, the first image 2000A depicts a frequency band from 3 GHz to 11 GHz, as well as the upper and lower frequency limits specified by the FCC for an unreserved band, within which the radio wave transmission spectrum (WRES) is depicted. The first WRES 2010 represents radio wave operation such that the pulse beam substantially fills the FCC-unreserved spectrum from 3.1 GHz to 10.6 GHz. In contrast, the second WRES 2020 and the third WRES 2030 represent radio wave operation in smaller portions of this FCC-unreserved spectrum (3.6 GHz to 5.1 GHz and 8.3 GHz to 9.8 GHz, respectively). The third WRES 2030 also depicts a scenario where the radio wave establishes a notch at 9 GHz in its PSD. It will be apparent to those skilled in the art that the WRES for radio will be established by local transmission regulations for wireless devices. It will be apparent to those skilled in the art that radio waves utilizing embodiments of the present invention can operate on different total frequency ranges, which are defined by the FCC according to their deployment scenarios (such as those defined by factors such as interoperability standards, applications, environment, and jurisdiction). Furthermore, it will be apparent to those skilled in the art that radio waves utilizing embodiments of the present invention can operate on different frequency ranges, which are part of those total frequency ranges that are defined or not defined based on factors such as interoperability standards, applications, environment, and jurisdiction. Therefore, radio waves according to embodiments of the present invention can operate, for example, on bandwidths such as 3.1 GHz to 10.6 GHz, 3.6 GHz to 5.6 GHz, and 5.25 GHz to 5.75 GHz within environments covered by FCC regulations. Referring to the second image 2000B, the frequency range from 3.1 GHz to 10.6 GHz is depicted as being divided into 15 frequency bands 2050, I, II, ..., XIV, XV. Each frequency band 2050 is 500 MHz wide. It will be apparent to those skilled in the art that the number of frequency bands can be varied, such that there exists a limited number of frequency bands whose width is defined by dividing the total frequency range by the number of frequency bands, or frequency bands can be specified within a limited frequency range (e.g., 500 MHz, 400 MHz, 1 GHz, 250 MHz, etc.), and thus the number of frequency bands is defined by dividing the total frequency range by this limited frequency range. Although the description with respect to Figures 20 to 25 is relative to frequency bands all having equal bandwidth, it will be apparent that in other embodiments of the invention, frequency bands may be defined in unequal ways in terms of bandwidth on regular or irregular grids. For example, even if bandwidth is defined in an equal or unequal manner, each bandwidth can be continuous, or the bandwidth can be discontinuous (equal or unequal on a regular or irregular grid). Therefore, referring to FIG. 21, the transmitter Tx_1 2140 according to an embodiment of the invention can transmit one or more pulse beams for each data block, which provides transmission covering the same frequency band (band IV in this example) for each sequential data block 2110. In embodiments of the invention, data block 2110 can be a bit, a group of bits, a packet, or a predetermined portion of data being transmitted. A receiver within another radio wave can similarly tune to band IV to receive the transmitted pulse beams from the transmitter Tx_1 2140. However, another transmitter, Tx_3 2160, can transmit one or more pulse beams for each data block, providing transmission covering the same frequency band (band VIII in this example) for each sequential data block. Again, a data block can be a bit, a group of bits, a packet, or a predetermined portion of data being transmitted. Therefore, a receiver within another radio wave can similarly tune to band VIII to receive the transmitted pulse beams from the transmitter Tx_3 2160. However, in another embodiment of the invention, transmitter Tx_2 2150 can transmit one or more pulse beams for each data block, providing a transmission range that varies within the occupied frequency band for each sequential data block across the entire frequency range. As depicted, Tx_2 2150 transmits data blocks according to a cyclical sequence of four blocks in frequency bands II, VII, XIII, and X. Again, a data block can be a bit, a group of bits, a packet, or a predetermined portion of the data being transmitted. However, compared to Tx_1 2140 and Tx_3 2160, transmitter Tx_2 2150 now uses frequency band hopping for each data block. Therefore, a receiver in another radio wave can similarly tune to frequency bands II, VII, XII, and X in the same sequence as transmitter Tx_2 2150 IV to receive the transmitted pulse beams from transmitter Tx_2 2150. It will be apparent to those skilled in the art that the number of available frequency bands within a cyclic sequence can vary depending on factors such as how the frequency bands are subdivided, the total frequency range, and the bandwidth of each frequency band. Furthermore, it will be apparent to those skilled in the art that the number of available frequency bands within the cyclic sequence can vary and can be 2, 3, 4, 5, 6, 7, 8, etc., and in some embodiments of the invention, each frequency band within the sequence is unique, while in other embodiments, a portion of the frequency bands within the sequence can occur multiple times, although the resulting sequence variations make the total length of the cyclic sequence very large. In embodiments of the invention, the frequency band sequence (such as the number of steps within the cyclic sequence, the number of frequency bands used, the actual sequence of frequency bands within repetitions, etc.) can be determined by the transmitter based on configuration settings provided to the transmitter or, as described below with respect to FIG25, by a spectrum sensing process as part of the transmitter configuration process. As is evident from the first inserted image 2120 and the second inserted image 2130, the transmitter then designs pulse beams for data blocks within each of frequency bands VII and XII to fill that frequency band. Thus, for example, by including frequencies The pulse beam establishes frequency band VII, while through frequency band VII... The pulse beam establishes frequency band XII. The number of frequencies used for each frequency band can be constant or variable. The number of pulses for each frequency in a frequency group used for a frequency band can be constant or variable. Furthermore, the parameters of each pulse at a given frequency can be constant, or, as described below with respect to other embodiments of the invention, the parameters of each pulse can be deterministically or pseudo-randomly varied, including but not limited to pulse width, pulse power, jitter frequency offset relative to the nominal frequency, jitter time offset relative to the nominal pulse position, and phase. Furthermore, the frequency sequence within the frequency band can be deterministically or pseudo-randomly varied such that, for example, in a subset of the pulse beam used for frequency band VII, the pulse sequence adopts a frequency... The pulse below, while another subset of the pulse beam can use a frequency The pulse below, and another subset of the pulse beam can be used It will be apparent to those skilled in the art that the more frequencies used within a pulse beam, the more combinations of frequency sequences can be used within different pulse beams, even if they are nominally in the same frequency band. It will also be apparent that, in some embodiments of the invention, a pulse sequence in a given frequency band can employ a different number of pulses at each frequency. For example, for a frequency sequence... and pulse counting sequence 3,3,3 (i.e., Three pulses were applied. Three pulses were applied. (3 pulses at the beginning), other pulse beams can be 2, 4, 3; 4, 3, 3; 3, 4, 3; 3, 3, 4; etc., provided that the accumulated power spectral density (PSD) meets regulatory requirements. Optionally, the frequency range that can be used can be limited by the bandwidth of the receiver filter. Referring now to FIG22, a transmission sequence from a transmitter according to an embodiment of the invention is schematically depicted, wherein during a first sequence 2250, the transmitter transmits pulse beams for each sequential data block 2210 according to frequency band sequences II, VII, XII, and X. However, the transmitter subsequently changes to a second sequence 2260 of frequency bands IV, VII, XII, and X. In a similar manner as described above, a receiver in another radio wave can similarly tune to the first sequence 2250 of frequency bands II, VII, XII, and X to receive the transmitted pulse beams from the transmitter, and then, for each sequential data block 2220 within the second sequence 2260, is notified to change to the second sequence 2260 of frequency bands IV, VII, XII, and X, the spectra of frequency bands IV and XIII being depicted as a first response 2230 and a second response 2240, respectively. A similar scenario is depicted in Figure 23, where during the first sequence 2350, the transmitter transmits pulse beams for each sequential data block according to a sequence of four frequency bands, including bands II, VII, VIII, and X. However, the actual frequency band sequence within each four-band sequence now changes. Thus, within the first sequence 2350 for bundles N-7 to N-4, the frequency bands are sequences II, VII, VIII, and X, while for bundles N-3 to N, the frequency bands are X, VIII, VII, and II. The sequence of each frequency band within the four-band sequence can be deterministic or pseudo-random. As mentioned above, the number of frequency bands within a sequence and the actual frequency bands can vary in other scenarios. However, the transmitter subsequently changes to a second sequence 2360 based on a configuration adjustment to bands II, IV, VIII, and X, which is transmitted to the radio wave by another radio wave, a network controller, etc., or established by the radio wave through, for example, spectrum sensing or monitoring of Quality of Service (QoS). Then, at another point in time, the transmitter changes to a third sequence 2370, which uses frequency bands III, IV, VIII, and X for beams N+M, N+M+1, N+M+2, and N+M+3. Thus, in a similar manner as described above, a receiver within another radio wave can similarly tune to a first sequence 2350 using frequency bands II, VII, XII, and X to receive the transmitted pulse beam from the transmitter, then be notified to change to a second sequence 2360 using frequency bands II, IV, VIII, and X, and then be notified to change to a third sequence 2370 using frequency bands III, IV, VIII, and X. Referring now to FIG24, an illustrative band transmission sequence according to an embodiment of the present invention is depicted, wherein the transmitter initiates a spectrum sensing phase Sense #1 2410 (Sense #1 2410) before using a pulse beam to transmit data blocks, wherein the transmitter transmits a pulse beam covering all or a predetermined portion of each band I to VII using spectrum sensing pulse beams Spec_Tx(1)2450 to Spec_Tx(N)2460. Thus, the transmitter is configured to operate in band I, for example, according to link data provided to the transmitter from another transmitter associated with a receiver receiving spectrum sensing pulse beams Spec_Tx(1)2450 to Spec_Tx(N)2460 within another radio wave, or from a network controller receiving one or more receivers (radio waves) receiving spectrum sensing pulse beams Spec_Tx(1)2450 to Spec_Tx(N)2460 via a communication interface. Once configured to transmit in band I, the transmitter transmits a signal within a first sequence Transmit #1 2420 (Transmit #1 2420) of a cyclic sequence comprising TX_A 2470 and TX_B 2480, each cyclic sequence comprising a series of four pulse beams transmitted on subbands of band I, these subbands being subbands AH, where Tx_A 2470 comprises subbands A, D, F, and H, and Tx_B 2480 comprises subbands B, C, E, and G. These are repeated sequentially, for example as described and depicted in Figures 22 and 23, until a second spectrum sensing phase Sense #2 2430 at a subsequent time point, where the transmitter transmits pulse beams covering all or predetermined portions of each band I through VII using spectrum sensing pulse beams Spec_Tx(1) 2450 to Spec_Tx(N) 2460. Therefore, assuming a spectral conflict is identified, the transmitter is reconfigured to transmit in band IV, where the transmitter transmits a signal within the sequence Transmit #2 2440 (Transmit #2 2440), which includes a sequence Tx_C 2490 consisting of four pulse beams transmitted on a sub-band of band IV, which is AH. As depicted, this causes the transmitter to switch to the second sequence Transmit #2 2440, where the pulse beam is now within band IV, and the pulse beams are sequentially located in sub-bands A, D, F, and H of this new band. Alternatively, the second spectrum sensing phase Sense #2 2430 may have already made a decision that there is no change in the band, or it may have determined that switching to one of the other bands II, III, and V through VII is more appropriate. Subsequent spectrum sensing phases may be automatically triggered, for example, after a predetermined number of pulse beams have been transmitted, a predetermined number of packets have been transmitted, or a predetermined period of time has elapsed since the last spectrum sensing process was performed. Alternatively, the determination to trigger another spectrum sensing process may be established based on data provided to the transmitter from a receiver forming part of the link with the transmitter, data received from a network controller, or another receiver. Optionally, the decision-making process within UWB radio waves according to embodiments of the present invention can be controlled, for example, by monitoring QoS through techniques such as packet error rate (PER) monitoring or signal-to-noise ratio (SNR) monitoring, either alone or in combination with spectrum sensing. Therefore, referring to Figure 25, three different exemplary scenarios are depicted. In the first scenario, the transmitter using mode Transmit #1 2510 determines after the spectrum sensing process that no reconfiguration is needed, and therefore continues to transmit using the same Transmit #2 2520 as Transmit #1 2510. In the second scenario, the transmitter using mode Transmit #3 2530 determines after the spectrum sensing process that no reconfiguration is needed, and therefore switches to transmitting using Transmit #4 2540, which is now in band II. In the third scenario, the transmitter using mode Transmit #5 2550 determines after the spectrum sensing process that reconfiguration is needed, and therefore transmits using Transmit #6 2560, which is now in band VII. Figure 26 depicts another scenario according to an embodiment of the invention, wherein the transmitter according to the embodiment operates on frequency bands I through XV and is initially configured to transmit on frequency bands II, VII, X, and XIII using the sequence Transmit #1 2610 (Transmit #1 2610). Subsequently, a spectrum sensing process generates a decision to transition to Transmit #2 2620 (Transmit #2 2620), where this sequence includes frequency bands III, VI, XI, and XII. Then, again at a subsequent time point, another spectrum sensing process generates a decision to transition to Transmit #3 2630, where this sequence includes frequency bands IV, VI, VII, and X. Thus, the scenario depicted in Figure 26 can involve multiple frequency bands such as those described in Figures 20 through 25, or it can involve a series of sub-bands within the frequency bands described in Figures 20 through 25. However, in other embodiments of the invention, spectrum sensing and adjustment can be related to frequencies within a frequency band or sub-band. Therefore, a wireless link according to an embodiment of the present invention can be configured to operate on multiple frequency bands and / or multiple sub-frequency bands, wherein the transmitter and receiver are configured to operate on the same frequency band and / or sub-frequency band, such that the transmitter and receiver are aligned with the same frequency band and / or sub-frequency band through one or more processes, said processes including but not limited to: ● The transmitter and receiver being synchronized by a network controller; ● The transmitter transmitting a sequence identifier of a predetermined sequence stored in the receiver's memory within the packet header of the data transmitted by the transmitter to the receiver; ● The transmitter transmitting to the receiver an identifier of the current frequency band and / or sub-frequency band of the current packet; and ● The transmitter transmitting to the receiver an identifier of the current frequency band and / or sub-frequency band used within the current packet. In embodiments of the invention, the transmitter in the first radio wave and the receiver in the second radio wave can utilize a discovery protocol at the start of communication to ensure that the transmitter and receiver find each other not only in time but also in which frequency band. 8. Multipath of multipulse beam / Cross-fading mitigation Referring to FIG27, an exemplary pulse beam according to an embodiment of the present invention is depicted, the pulse beam comprising first to ninth pulses 2710 to 2790, which are positioned within the pulse beam and are respectively timed t. 1, t 2, t 3, t 4. t 5. t 6. t 7. t 8 and t Centered on 9. These pulses each have a width W 1. W 2. W 3. W 4. W 5. W 6. W 7. W 8 and W 9. The first pulse 2710 to the third pulse 2730 are at the first frequency f. 1. The fourth pulse 2740 to the sixth pulse 2760 are at the second frequency f. 2, while the seventh pulse 2770 to the ninth pulse 2790 are in f 3. However, in embodiments of the present invention, in a general configuration, the pulse beam may include: ● a plurality of pulses generated by an RF signal generator. pulses, of which Each pulse in a pulse is in multiple A predetermined frequency, having a predetermined amplitude, and having a predetermined pulse length; wherein, ● [[;]]and ; ● and It is an integer; ● The pulse repetition rate of the RF signal generator is determined based on the clock frequency applied to the RF signal generator; ● Integer Depending on the duration of the data signal bits and the pulse repetition rate of the RF signal generator; and ● multiple Each pulse is emitted within the duration of a bit in the data signal, and includes multiple pulses. Pulses at a frequency of [number]. Figure 2 illustrates an RF signal generator for generating such a pulse beam according to an embodiment of the present invention. Furthermore, each pulse within the pulse beam may have a phase ϕ associated with it. 1. ϕ 2, ϕ 3, ϕ 4. ϕ 5. ϕ 6. ϕ 7、ϕ 8 and ϕ 9. Using the RF signal generator depicted in Figure 3, each pulse is assigned to either a first phase or a second phase according to the settings of switch 360. As described in WO / 2015 / 103,692, switch 360 is fed a pseudo-random signal such that the phases of the pulses within the pulse beam have pseudo-randomly assigned phases. However, in a wireless link between a first radio wave including a transmitter according to an embodiment of the present invention and a second radio wave including a receiver according to an embodiment of the present invention, the wireless signal may experience signal degradation such as multipath interference and cross fading. Referring now to Figure 28, the first pulse beam 2810 to the fourth pulse beam 2840 are depicted, each of which includes a first frequency f. 1. Three pulses at the second frequency f 2. Three pulses and the third frequency f The three pulses are 3. Since the first pulse beam 2810 to the fourth pulse beam 2840 each include the same frequency and the same number of pulses at each frequency, the transmit power spectrum 2850 of each pulse beam is the same, regardless of the phase of the pulses, provided that each pulse beam in the first pulse beam 2810 to the fourth pulse beam 2840 has the same amplitude. However, it is obvious that the actual pulse sequences of each pulse beam in the first pulse beam 2810 to the fourth pulse beam 2840 are different, and they are: ● The first pulse beam 2810 includes a frequency sequence f 1, f 1, f 1, f 2, f 2, f 2, f 3, f 3, f 3; ● The second pulse beam 2820 includes a frequency sequence f 2, f 2, f 2, f 1, f 1, f 1, f 3, f 3, f 3; ● The third pulse beam 2830 includes the frequency sequence f 1, f 1, f 1, f 2, f 3, f 3, f 3, f 2, f 2; and ● The fourth pulse beam 2840 includes the frequency sequence f 2, f 1, f 1, f 3, f 2, f 1, f 3, f 2, f 3. Referring now to Figure 29, a grouping 2900 is depicted, comprising a header 2910 and multiple data blocks 2920(1), 2920(2), 2920(3) to 2920(N). As depicted: ● The first data block 2920(1) comprises a pulse sequence defined by the first pulse beam 2810 in Figure 28; ● The second data block 2920(2) comprises a pulse sequence defined by the second pulse beam 2820 in Figure 28; ● The third data block 2920(1) comprises a pulse sequence defined by the third pulse beam 2830 in Figure 28; and ● The fourth data block 2920(1) comprises a pulse sequence defined by the fourth pulse beam 2840 in Figure 28. Referring to FIG30A, a series of packets 3010B to 3040B transmitted by a transmitter according to an embodiment of the present invention are depicted. In the described embodiment of the invention, the frequencies within each of the packets 3010 to 3040 are the same, but their sequences are different. Thus, as depicted: ● A first packet 3010B includes a header and a plurality of data blocks using a frequency sequence defined by a first pulse beam 2810 in FIG28; ● A second packet 3020B includes a header and a plurality of data blocks using a frequency sequence defined by a second pulse beam 2820 in FIG28; ● A third packet 3030B includes a header and a plurality of data blocks using a frequency sequence defined by a third pulse beam 2830 in FIG28; and ● A fourth packet 3040B includes a header and a plurality of data blocks using a frequency sequence defined by a fourth pulse beam 2840 in FIG28. Referring now to FIG30B, a series of packets 3052 to 3058 transmitted by a transmitter according to an embodiment of the invention are depicted. In the embodiments of the invention described, as each packet is located in a different frequency band of the UWB link spectrum accessible according to constraints defined by, for example, the UWB transmitter and receiver along with regulatory requirements, not only does the sequence of pulses vary (i.e., it is not a constant sequence with respect to frequency), but the frequency itself also varies. For example, this is a repeating sequence of frequency bands, as described above with respect to FIGS. 21 to 26 respectively. Thus, as depicted: ● The first packet 3052 includes a header and uses a frequency f 1, f 2 and f 3. Multiple data blocks of the first frequency sequence; ● The second group 3054 includes a header and uses a frequency f 4, f 5 and f 6. Multiple data blocks of the second frequency sequence; ● The third group 3056 includes a header and uses a frequency f 7, f 8 and f The third frequency sequence of 9 consists of multiple data blocks; and the fourth group 3058 includes a header and uses a frequency f 10 f 11 and f 12 Multiple data blocks of the fourth frequency sequence. Furthermore, as described above with respect to Figure 27, it is evident that the pulses within each of the first group 3010B to the fourth group 3040B in Figure 30A, or within each of the first group 3052 to the fourth group 3058 in Figure 30B, may also have other aspects of the pulses, such as pulse spacing, phase, pulse width, etc., which may vary pseudo-randomly or programmatically within the sequence. These are depicted as the first to fourth pulse beams 3060B to 3090B, respectively. Optionally, in other embodiments of the invention, programmable or fixed-time guard bands may be employed between symbols. Optionally, in other embodiments of the invention, the symbol repetition rate may be varied to accommodate locations where link quality is degraded due to inter-symbol interference. 8A. Programming and Pseudorandom In the preceding description of Figures 20 to 30, multi-frequency pulse beams are employed within each bit, each symbol, or each group, or across multiple bits, multiple symbols, or multiple groups. However, in Figures 22 to 23 and Figures 27 to 30, the multi-frequency beam can be within a single frequency band until a decision to adjust to another frequency band or switch one or more frequency bands within a repeating frequency band sequence can be made by a method such as the spectrum sensing described in Figures 24 to 26. However, in another embodiment of the invention, the decision to switch from one frequency band to another, switch one or more frequency bands within a repeating frequency band sequence, or switch the frequency sequence within a multi-pulse beam can be made based on the degradation of link characteristics between the transmitter and one or more receivers, which is fed back to the transmitter from one or more receivers or via a network controller. This degradation of link characteristics is caused by the aforementioned effects such as multipath interference and cross-fading. Therefore, the transmitter can adjust the frequency sequence within the multi-pulse beam (as described and depicted in Figures 29 and 30) during the training sequence or during data transmission to determine whether the adjustment of the frequency sequence of the pulses within the multi-frequency beam results in a reduction of the signal degradation observed, such as the absolute power received at the receiver and / or the effective signal-to-noise ratio between the pulse beam for "1" and the pulse beam for "0". In a similar manner, a decision can be made to switch to another frequency band among multiple frequency bands to determine whether using a different frequency band will similarly produce an improvement in the quality of the received signal. The adjustment from one frequency sequence to another can be programmed by a configuration sequence stored in memory associated with the transmitter, or it can be pseudo-randomly generated from a set of configuration sequences stored in memory. Alternatively, the adjustment from one frequency sequence to another can be determined discretely or in combination with one or more link metrics based on spectrum sensing. 8B. Multi-pulse beam parameter jitter As shown in Figure 27, the pulses within a multi-pulse beam are defined by their position within the pulse beam (which can be changed as described in Section 8A above), their frequency, pulse width, phase, and amplitude, as well as the pulse repetition rate from the RF signal generator within the transmitter. However, these parameters can be jittered, either programmatically or pseudo-randomly, to adjust the actual content of the multi-frequency pulse beam. This can be achieved, for example, by: ● jittering the pulse width of each pulse within the multi-frequency pulse beam within predetermined limits; ● jittering the frequency of each pulse within the multi-frequency pulse beam within predetermined limits; ● jittering the amplitude of the pulses within the multi-frequency pulse beam within predetermined limits; ● jittering the pulse position of each pulse within the multi-frequency pulse beam within predetermined limits; and ● adjusting the pulse repetition rate of the RF signal generator to change the interval between pulses within the multi-frequency beam. Therefore, when controlled in a programmed manner, adjusting these parameters allows the receiver to determine how link quality of service (QoS) metrics improve or deteriorate, and to sequentially adjust them to achieve improved link QoS. Furthermore, adjusting the PRR allows the transmitter to adjust the pulse interval within a multi-pulse beam to similarly determine how link QoS metrics improve or deteriorate. Optionally, the transmitter according to embodiments of the invention can dynamically adjust the symbol interval to adjust and seek enhanced QoS, albeit at the cost of a sustainable maximum data rate. 8C. Chirped pulse In the description relating to Figures 2 to 30, each pulse within the multi-frequency beam has been described and depicted as being at a single discrete frequency. However, in other embodiments of the invention, the pulses may be chirped such that a predetermined chirp is applied to each pulse. This could be an increase in frequency during an up-chirp of the pulse; a decrease in frequency during a down-chirp of the pulse; an up-chirp followed by a down-chirp; or a down-chirp followed by an up-chirp. The applied chirp may be linear, where the chip changes linearly with time during the pulse, or the chirp may be geometric, where the chip changes geometrically with time during the pulse. For example, a geometric chirp may be sinusoidal or exponential. Therefore, the transmitter according to embodiments of the present invention can apply linear frequency modulation of different degrees and forms to the pulses within a multi-pulse beam over time, so that QoS metrics can be measured and transmitter parameters can be adjusted to obtain improved QoS, and different aspects of the pulses can be continuously adjusted to continuously tune to obtain improved QoS. 9. UWB Training sequences for link configuration In the preceding description of Figures 20 to 30, multi-frequency pulse beams are employed within each bit, each symbol, or each group, or across multiple bits, multiple symbols, or multiple groups. However, in Figures 22 to 23 and Figures 27 to 30, the multi-frequency beam can be within a single frequency band until a decision to adjust to another frequency band or switch one or more frequency bands within a repeating frequency band sequence can be made by a method such as the spectrum sensing described in Figures 24 to 26. However, in another embodiment of the invention, the decision to switch from one frequency band to another, switch one or more frequency bands within a repeating frequency band sequence, or switch the frequency sequence within a multi-pulse beam can be made based on the degradation of link characteristics between the transmitter and one or more receivers, which is fed back to the transmitter from one or more receivers or via a network controller. This degradation of link characteristics is caused by the aforementioned effects such as multipath interference and cross-fading. Alternatively, in other embodiments of the invention, a decision can be made at the beginning of establishing the link between the transmitter and receiver, wherein an initial training sequence can be employed, causing the transmitter to cycle through a predetermined sequence of changes in the characteristics of the multi-frequency pulse beam, such as stepping through different frequency bands, so that the receiver determines the QoS metric for each frequency band and establishes the configuration subsequently used in the link. Optionally, in embodiments of the invention, the initial configuration can be used until the QoS drops below a predetermined threshold, wherein the transmitter can trigger a new training sequence or spectrum sensing process, etc., to establish a new transmitter configuration setting. 10. Vivaldi (Slotted) Diversity Antenna As described above with respect to Figure 4, diversity can provide enhanced link performance in situations where signal degradation occurs, such as due to multipath interference and fading. Therefore, when implementing a UWB transmitter, UWB receiver, and UWB transceiver, a directional antenna can be used to enhance the diversity antenna to provide improved spatial filtering. Thus, an antenna known as a Vivaldi antenna (or slotted antenna) not only provides a directional radiation pattern but also ultra-wideband performance. Therefore, the inventors have developed a diversity antenna structure combining two oppositely oriented Vivaldi antennas to provide maximum coverage around UWB radio waves. As depicted in Figure 31, the diversity antenna according to an embodiment of the invention includes a first Vivaldi antenna 3110 and a second Vivaldi antenna 3130, arranged side-by-side relative to each other, each fed by a coplanar waveguide transmission line. The coplanar waveguide is connected within the UWB device according to an embodiment of the invention to an RF switch, which connects either antenna to a transmitter and / or receiver circuit having a UWB transmitter, UWB receiver, and UWB transceiver. An exemplary antenna diversity system (the results of which are shown in Figures 32 to 35) was designed to fit within the finite dimensions of a PCB, thus making it a compact system. The structure was simulated in Ansys HFSS, a FEM (finite element method) based simulation software, with a total size of 67 mm × 34 mm. The simulated S11 (dB) is depicted in Figure 32, illustrating the ultra-wideband impedance matching performance from 3.5 GHz to 8 GHz, ensuring low-loss performance of the antennas. The two antennas are well isolated because their radiation patterns point in two different directions. This ensures a very low ECC, thereby increasing the diversity gain of the system. The simulated ECC is depicted in Figure 33. Figures 34A to 34C depict the antenna radiation patterns of a diversity antenna using the Vivaldi antenna according to an embodiment of the present invention in the frequency band of interest. These frequencies are 3.5 GHz, 4.5 GHz, and 5.5 GHz, respectively. It can be seen from these that uniformity of the radiation pattern is maintained over the desired bandwidth, while a peak realized gain range of 3.2 dBi to 3.5 dBi is obtained in the 3.5 GHz to 5.5 GHz range, which is a sub-band of the total operating bandwidth of the diversity antenna used in UWB wireless devices manufactured by Spark Microsystems, Inc., Montreal, Canada. Figures 35A and 35B depict the peak realized gain and radiation efficiency of the designed antenna. 11. Dynamic differential drive diversity antenna selection As described above in sections 4 and 10 regarding diversity antennas with dual antennas according to embodiments of the invention, a UWB transmitter can be connected to either of the two antennas in an exemplary diversity antenna configuration. However, in the case where the UWB transmitter includes a differential configuration, the UWB transmitter has a differential output. Thus, referring to FIG36, for a transmitter according to an embodiment of the invention, UWB circuit 3610 provides a differential RF output coupled to RF switch 3620, which provides functionality equivalent to a double-pole double-throw switch, wherein in one state, RF switch 3620 connects the differential RF output of UWB circuit 3610 to a first antenna (antenna #1 3630), and then in another state, RF switch 3620 connects the differential RF output of UWB circuit 3610 to a second antenna (i.e., antenna #2 3640). Antenna #1 3630 and antenna #2 3640 are two antennas within diversity antenna 3650. Based on the switching speed of the RF switch 3620, the UWB device with the configuration depicted in FIG36 can be reconfigured at each per-packet level. Optionally, if the UWB receiver supports an RF front-end with differential capability, the circuit depicted in FIG36 can be used to connect the dual outputs of the antenna according to an embodiment of the present invention to the receiver UWB circuitry. Now considering a UWB receiver, and referring to Figure 37, a diversity antenna 3700 is depicted, comprising antenna #1 3710 and antenna #2 3720. Antenna #1 3710 is connected to a first RF front end (RF receiver signal chain #1 3730), while antenna #2 3720 is connected to a second RF front end (RF receiver signal chain #2 3740). The outputs from each of RF receiver signal chains #1 3730 and #2 3740 are coupled to a control circuit 3750, which determines which of receiver signal chains #1 3730 and #2 3740 provides improved signal reception and thus de-energizes the other circuit. Due to the fast power cycling of the UWB receiver established by the inventors, the UWB receiver according to an embodiment of the invention can, for example, determine from the preamble within each packet which antenna to use, select the appropriate antenna, and de-energize the other antenna, such that the UWB receiver according to an embodiment of the invention can establish a configuration for each packet. Optionally, antenna #1 3710 and antenna #2 3720 may each provide a hemispherical radiation pattern, such that the pair forms a spherical coverage. 12. Chip-type balon antenna subsystem An antenna subsystem is required to establish a wireless link between the UWB transceiver (radio) and the outside world, such as that described and depicted in Figure 7. This subsystem may include several components essential for the overall system performance. Furthermore, the subsystem should shape the output spectrum of the UWB radio to meet stringent spectrum masking requirements. This spectrum masking may, for example, be associated with a geographical area. Figures 2 and 3A depict exemplary architectures of IR-UWB transmitters with and without dual-phase scrambling according to embodiments of the invention. In such IR-UWB transmitters, differential output signals may be employed, requiring baffles to convert balanced signals into unbalanced (single-ended) signals. As depicted in Figure 38, first image 3800A and second image 3800B respectively depict exemplary architectures of UWB radio modules utilizing chip (circuit) baffles in first image 3800A or printed circuit board (PCB)-based baffles in second image 3800B. The first image 3800A depicts a UWB radio 3810 that provides a differential output signal to a second harmonic suppression filter 3820, which then generates a differential output signal to a chip bangle 3830. The chip bangle 3830 generates a single-ended signal that is transmitted to an antenna 3850 via a low-loss passband filter 3840. The second image 3800B depicts a UWB radio 3860 that provides a differential output signal to a PCB bangle / filter 3870. The output of the PCB bangle / filter 3870 is a single-ended signal coupled to an antenna 3880. 12A. Chip-type balon antenna subsystem Referring to FIG39, first images 3900A to fourth images 3900D of a chip bailon antenna subsystem according to an embodiment of the present invention are depicted, wherein each of the first images 3900A to fourth images 3900D represents a different layer of a PCB incorporating UWB radio with a chip bailon antenna subsystem. First elements 3910 to ninth elements 3990 are depicted in these first images 390A to fourth images 3900D, respectively. These elements include: The first element 3910, located within the top layer of the PCB depicted in the first image 3900A, includes a monopole antenna with a single-frequency notch filter. In this embodiment, a UWB monopole antenna is used, which can provide an omnidirectional radiation pattern around the antenna, while a notch slot filter is designed on the antenna patch to suppress signals near 5.8 GHz. The second element 3920, located within the top layer of the PCB depicted in the first image 3900A, represents the top layer of the defective ground structure (DGS) bandpass filter used to reduce spurious out-of-band radiation while providing a low-loss and compact bandpass filter. The design established by the inventors using a cavity-backed defective ground structure has an extended suppression region up to 25 GHz, while providing low loss (typical insertion loss of 0.8 dB) and low radiation loss in a compact size of 9.0 x 2.0 mm (~0. inch x 0. inch). The top layer of the filter consists of interdigitated series capacitors and two stepped impedance resonators to suppress low-frequency signals. The third element 3930, located within the top layer of the PCB depicted in the first image 3900A, represents a sheet-type baffle. For example, this could be a variation of the design described in Figures 17A and 17B, respectively. The fourth element, 3940, located within the top layer of the PCB depicted in the first image 3900A, represents a second harmonic open-stub filter, where the feed line consists of two open-stubs tuned to the center frequency of the second harmonic (e.g., 14 GHz). This thus increases attenuation in that frequency region. Furthermore, this structure is tuned to provide better impedance matching between the UWB radio 3900E and the required subsequent 50 systems. The fifth element 3950, located within the top layer of the PCB depicted in the first image 3900A and within the third PCB layer depicted in the second image 3900B, includes a series of dumbbell-shaped slots for a DGS filter, wherein the series of dumbbell-shaped slots in the ground plane provides a wide stopband feature for suppressing spurious signals associated with the second and third harmonics. The seventh element, 3970, located within the third PCB layer depicted in the second image 3900B, includes a pair of ground plane slots. The electromagnetic interference (EMI) shielding element used in the UWB radio 3900E can function as a cavity resonator at certain frequencies, causing undesirable radiation. To mitigate this effect, a pair of 2 / 3 slots are formed within the ground plane, such that at the cavity resonant frequency, the excitation mode is coupled to the bottom layer of the PCB, and energy is dissipated rather than radiated. The eighth element 3980, located within the third PCB layer depicted in the second image 3900B, comprises a pair of tapered ground plane slots. Due to the large ground plane, radiation nulls may appear in the radiation pattern of the antenna module, which could have a detrimental effect on the performance of the entire system. To mitigate these nulls, the two tapered slots represented by the eighth element 3980 are designed into each side of the ground plane. By altering the path of the surface current, these slots improve the radiation performance of the module and provide a uniform radiation pattern. The ninth element 3990 includes a slot within the second PCB layer depicted in the third image 3900C, which is placed in the second PCB layer to couple the wave to the bottom below. Therefore, the combination of the second element 3920, the sixth element 3960, and the ninth element 3990, bounded by vias, forms a cavity-backed bandpass filter using DGS and interdigitated series capacitors. The fourth image, 3900D, represents the bottom PCB layer. Therefore, the subsystem shown in Figure 39 provides a uniform omnidirectional radiation pattern across the entire frequency range of the UWB radio 3900E, effectively filtering second and third harmonics as well as low-frequency unwanted radiation with low loss, and providing a shielding structure with suppression of cavity resonant modes to achieve low electromagnetic interference and a compact, low-loss ballast. Referring to Figures 40A to 40C, the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 39 according to an embodiment of the present invention is shown in the band of interest in the frequency range of 6.5 GHz to 8.5 GHz. Figures 41A and 41B depict S11 and peak gain of the UWB antenna subsystem depicted in Figure 39 according to an embodiment of the present invention. 12B. PCB Bellon antenna subsystem Referring to Figure 42, a PCB baffle / filter circuit 4200 for converting a balanced signal into a single-ended signal is depicted, such as the PCB baffle / filter 3870 in the second image 3800B of Figure 38. Within the PCB baffle / filter 4200, first elements 4210 to sixth elements 4260 are depicted respectively. The first element 4210, comprising a monopolar antenna having a single frequency notch filter, is used to provide an omnidirectional radiation pattern around the antenna. Notch slot filters are designed on antenna patches to suppress frequencies around 5.8 GHz. The second element 4220 , comprising a pair of curved ground plane slots. Due to the large ground plane, the radiation zero point appears in the radiation direction diagram of the antenna module, thereby having a destructive effect on the performance of the whole system. To mitigate these zero points, two curved ground plane slots are designed on each side of the ground plane, which improves the radiation performance of the module by changing the path of the surface current and provides a uniform radiation pattern. A third element 4230 , indicating an open-circuit stub filter on the output of PCB Bellenger / filter 4240 prior to the monopole antenna (first element 4210). The fourth element 4240, including a PCB Belle / filter, employs a “rat-race” hybrid joint where the incremental output is connected to the antenna and the summation port is grounded. Belleng is loaded with a pair of open-circuit stub filters (fifth element 4250) to suppress undesirable signals from secondary and tertiary harmonics. Four more open-circuit stub filters were added before and after Belleng to improve the level of suppression of the out-of-band signal (not marked for clarity, but clearly visible in the inner part of the “rat-race” hybrid joint). The fifth element 4250 , indicating an open-circuit stub filter, is located on the input of the PCB Belleng / filter (fourth element 4240), providing improved suppression of the common and differential mode signals before the common and differential mode signals enter the Belleng. The sixth unit 4260 is a UWB radio. Therefore, the present inventors have built a PCB Behring / filter circuit 4200, which provides excellent filtering capability for high-frequency stray harmonics and integrates the filtering and Behring functions into one PCB structure, resulting in a compact and low-cost solution. The PCB Belleng / Filter circuit 4200 provides an omnidirectional radiation pattern with uniform gain around the antenna module for the entire desired frequency band. Therefore, innovations within the PCB Belle / Filter circuit 4200 include the use of ground plane slots for smoothing of the radiation pattern combined with a compact PCB Belle / filter structure that excellently filters high-frequency spurious signals, providing uniform gain for all frequencies around the antenna. 43A to 43C depicts the antenna radiation pattern over the frequency band of interest of the UWB antenna subsystem depicted in FIG. 44A and 44B depict the S11 and peak gains of the UWB antenna subsystem depicted in Figure 39 according to an embodiment of the present invention. 13. Differential antenna with harmonic suppression As described above, UWB radios can be directly connected to a differential antenna or indirectly connected to a single-ended antenna using a bail. These single-ended antenna modules, as mentioned above, can have integrated filtering capabilities to suppress unwanted spurious signals. This is possible where the additional cost and insertion loss of the bail are acceptable. However, in some UWB systems, these costs are unacceptable. Therefore, the present invention has developed a novel UWB antenna module to address this issue, employing either the filtered loop antenna method described below in Section 13A or the differential antenna module described below in Section 13B. Each harmonic suppression is achieved through a differential low-pass filter, thereby eliminating the need for a bail, which in turn reduces the losses and manufacturing costs associated with this additional component. 13A. Filtered loop antenna module Referring to FIG45, an exemplary filtered loop antenna module 4500 according to an embodiment of the present invention is depicted, having an overall size of 20 mm (0.8 inches) × 18 mm (0.7 inches). Within the filtered loop antenna module 4500, first element 4510 to third element 4530 are depicted respectively. The first element 4510 is a differential feed line that connects the UWB radio to the front end of the antenna and includes an open stub attenuation notch filter to attenuate the second harmonic signal. The second element, 4520, is a differential low-pass filter that suppresses both common-mode and differential-mode spurious signals. The filter's structure is similar to the filter previously described for the monopole antenna subsystem. The input differential signal is separated between two 50Ω microstrip lines, then filtered by a defective ground structure (DGS) low-pass filter before being fed into the differential antenna (third element, 4530) and radiated. The third element 4530 is a differential loop antenna loaded with parasitic elements, wherein the structure of the antenna is similar to the loop antenna described above. Advantageously, the filtered loop antenna module 4500 provides a compact and low-loss structure for a novel differential low-pass filter, offering a wide suppression region for the second and third harmonics and a low-cost, high-efficiency structure that does not require ballast. Furthermore, the high-pass properties of the antenna, combined with the low-pass filter, provide bandpass radiation of the incident wave, as depicted in the accompanying figures, and can be observed in Figure 46B. Referring to Figures 46A and 46B, S11 and radiation efficiency of the UWB antenna subsystem depicted in Figure 45 according to an embodiment of the present invention are illustrated. It is thus evident that the filter ring module 4500 provides good impedance matching from 6.2 GHz to 8.35 GHz, covering the frequency bandwidth required for exemplary UWB radios (such as those manufactured by Spark Microsystems, Inc., Montreal, Quebec, Canada). Furthermore, the radiation efficiency curve in Figure 46B illustrates suppression of high-frequency and low-frequency signals under harmonics. Referring now to Figures 47A-47C, the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 45 according to an embodiment of the present invention is illustrated in the band of interest. 13B. Differential patch antenna module Similar structures to the feed line and filter described above with respect to FIG. 45 can also be used for different differential antenna types, such as the type depicted in FIG. 48. Therefore, FIG. 48 depicts an exemplary differential patch antenna module according to an embodiment of the present invention. Within the differential patch antenna module 4800, first elements 4810 to third elements 4830 are depicted respectively. The first element 4810 is a differential feed line that connects the UWB radio to the front end of the antenna and includes an open stub attenuation notch filter to attenuate the second harmonic signal. The second element, 4820, is a differential low-pass filter that suppresses both common-mode and differential-mode spurious signals. The filter's structure is similar to the filter previously described for the monopole antenna subsystem. The input differential signal is separated between two 50Ω microstrip lines, then filtered by a defective ground structure (DGS) low-pass filter before being fed into the differential antenna (third element, 4830) and radiated. The third element 4830 is a differential loop antenna loaded with parasitic elements, wherein the structure of the antenna is similar to the loop antenna described above. As is evident from Figure 48, this antenna provides a smaller, more compact module overall, and as can be observed from the radiation pattern curves in Figures 49A to 49C, the side antenna gain (Phi=0°, Theta=90°) is higher than that of a loop antenna. This, for example, improves coverage in indoor environments. Referring to Figures 49A to 49C, the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 48 according to an embodiment of the present invention is illustrated in the frequency band of interest. Now referring to Figures 50A and 50B, S11 and radiation efficiency of the UWB antenna subsystem depicted in Figure 48 according to an embodiment of the present invention are illustrated. Therefore, it is evident that the antenna exhibits good impedance matching in the frequency range of interest (6.0 GHz to 8.7 GHz), and the radiation efficiency reflects the bandpass properties of the overall structure, resulting in effective suppression of unwanted signals. Specific details are set forth in the foregoing description to provide a thorough understanding of the embodiments. However, it should be understood that the embodiments may be practiced without these specific details. For example, circuits may be shown in block diagrams so as not to obscure the embodiments with unnecessary details. In other instances, well-known circuits, processes, algorithms, structures, and techniques may be shown without unnecessary details to avoid obscuring the embodiments. The aforementioned technologies, blocks, steps, and devices can be implemented in various ways. For example, these technologies, blocks, steps, and devices can be implemented in hardware, software, or a combination thereof. For hardware implementation, the processing unit can be implemented in one or more application-specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), field-programmable gate arrays (FPGAs), processors, controllers, microcontrollers, microprocessors, other electronic units designed to perform the aforementioned functions, and / or combinations thereof. For purposes of illustration and description, the foregoing disclosure of exemplary embodiments of the invention has been provided. It is not intended to be exhaustive or to limit the invention to the precise forms disclosed. Based on the foregoing disclosure, many variations and modifications of the embodiments described herein will be apparent to those skilled in the art. The scope of the invention is defined only by the appended claims and their equivalents. Furthermore, in describing representative embodiments of the invention, the specification may have provided the methods and / or processes of the invention as a specific sequence of steps. However, the methods or processes should not be limited to the specific sequence of steps described herein, to the extent that they are independent of the specific order of steps set forth herein. Other sequences of steps are also possible, as will be understood by those skilled in the art. Therefore, the specific order of steps set forth in the specification should not be construed as a limitation on the scope of the claims. Moreover, the scope of the claims for the methods and / or processes of the invention should not be limited to the steps performed in the written order, and those skilled in the art will readily understand that the sequence can be varied and still remain within the spirit and scope of the invention. 110: First residential environment 120: Second residential environment 130: Industrial environment 140: Personal area network 150: Medical imaging 200: IR-UWB transmitter 210: Gate 220: Power loop controller 230: Pulse generator 240: Programmable multi-loop digitally controlled ring oscillator 250: Variable gain amplifier 260: Driver 270: Antenna 300: Block diagram 310: Gate 320: Power loop controller 330: Pulse generator 340: Programmable multi-loop digitally controlled ring oscillator 350: Dual output amplifier 360: Switch 370: Antenna 380: Output power amplifier 400: IR-UWB receiver 410: Antenna 420: Low noise amplifier 430: First amplifier 440: Squaring circuit 450: Second amplifier 460: Integrating circuit 470: Flash memory ADC 480: Power Cyclic Controller; 500: Receiver; 510: Antenna; 520: Low-Noise Amplifier; 530: Two-Stage RF Amplifier; 540: First-Square Mixer; 550: Variable Gain Amplifier; 560: Second-Square Mixer; 580: Processing Circuit; 600: UWB Transceiver; 605: DC-DC Converter; 610: Bandgap Reference; 615: Low-Frequency Crystal Oscillator; 620: Sleep Counter; 625: Receiver; 635: PHY Formatting Circuit; 640: Interface Circuit; 645: Link Controller; 650: PHY Processing Circuit; 655: PLL; 660: DLL 665: Spectrum Configuration Circuit; 670: Multiplexer; 675: Pulse Generator; 680: Mixer; 685: Power Amplifier; 690: Receive / Transmit Switch; 1210: Ring Element; 1220: First Parasitic Element; 1230: Second Parasitic Element; 1240: Third Parasitic Element; 1250: Differential Feed; 2000A: First Image; 2000B: Second Image; 2010: First WRES; 2020: Second WRES; 2030: Third WRES; 2050: Bandwidth; 2110: Data Block; 2120: First Inserted Image; 2130: Second Inserted Image; 2140: Transmitter Tx_1; 2150: Transmitter Tx_2; 2160: Transmitter Tx_3 2210: Sequential data block 2220: Sequential data block 2230: First response 2240: Second response 2250: First sequence 2260: Second sequence 2350: First sequence 2360: Second sequence 2370: Third sequence 2410: First spectrum sensing phase Sense #1 2420: First sequence Transmit #1 2430: Second spectrum sensing phase Sense #2 2440: Second sequence Transmit #2 2450: Spectrum sensing pulse beam Spec_Tx(1)2460: Spectrum Sensing Pulse Beam Spec_Tx(N) 2470: TX_A 2480: TX_B 2490: TX_C 2510: Transmit #1 2520: Transmit #2 2530: Transmit #3 2540: Transmit #4 2550: Transmit #5 2560: Transmit #6 2610: Transmit #1 2620: Transmit #2 2630: Transmit #3 2710: First pulse 2720: Second pulse 2730: Third pulse 2740: Fourth pulse 2750: Fifth pulse 2760: Sixth pulse 2770: Seventh pulse 2780: Eighth pulse 2790: Ninth pulse 2810: First pulse beam 2820: Second pulse beam 2830: Third pulse beam 2840: Fourth pulse beam 2900: Packet 2910: Header 2920 (1) ~ 2920 (N): Data block 3000: Transmitter 3010: Pulse mode block 3020: Multiplexer 3030: Delay-locked loop 3040: Digitally controlled oscillator 3050: Pulse generator 3052: First packet 3054: Second packet 3056: Third packet 3058: Fourth packet 3060: Power amplifier 3070: Antenna 3010B: First Group 3020B: Second Group 3030B: Third Group 3040B: Fourth Group 3060B: First Pulse Beam 3070B: Second Pulse Beam 3080B: Third Pulse Beam 3090B: Fourth Pulse Beam 3100A: First Image 3100B: Second Image 3110: First Vivaldi Antenna 3120: UWB Mask 3130: Second Vivaldi Antenna 3160: Bits 3160A~3160C: Subpulse 3180: Multipulse Spectrum 3200A: First Trace 3200B: First Trace 3200C: Third Trace 3300A: First Trace 3300B: Second Trace 3400A: First Image 3400B: Second Image 3400C: Third Image 3400D: Fourth Image 3610: UWB Circuit 3610 3620: RF switch; 3630: First antenna #1; 3640: Second antenna #2; 3650: Diversity antenna; 3700: Diversity antenna; 3710: Antenna #1; 3720: Antenna #2; 3730: RF receiver signal chain #1; 3740: RF receiver signal chain #2; 3750: Control circuit; 3800A: First image; 3800B: Second image; 3810: UWB radio; 3820: Second harmonic suppression filter; 3830: Chip ballast.3840: Low-loss passband filter; 3850: Antenna; 3860: UWB radio; 3870: PCB baluster / filter; 3880: Antenna; 3900: UWB radio; 3900A: First image; 3900B: Second image; 3900C: Third image; 3900D: Fourth image; 3900E: UWB radio; 3910: First element; 3920: Second element; 3930: Third element; 3940: Fourth element; 3950: Fifth element; 3960: Sixth element; 3970: Seventh element. Component 3980: Eighth Component; 3990: Ninth Component; 4200: PCB Bellagio / Filter Circuit; 4210: First Component; 4220: Second Component; 4230: Third Component; 4240: Fourth Component; 4250: Fifth Component; 4260: Sixth Component; 4500: Filtered Loop Antenna Module; 4510: First Component; 4520: Second Component; 4530: Third Component; 4800: Differential Patch Antenna Module; 4810: First Component; 4820: Second Component; 4830: Third Component; 6000: Transmitter Embodiments of the invention will now be described by way of example only with reference to the accompanying drawings, in which: Figure 1 depicts an application of a UWB transmitter, receiver, and system according to an embodiment of the invention; Figure 2 depicts a block diagram of a UWB transmitter according to an embodiment of the invention; Figure 3A depicts a block diagram of a UWB transmitter supporting biphasic phase scrambling according to an embodiment of the invention; Figure 3B depicts a block diagram of a UWB transmitter employing dynamically configurable and programmable pulse sequences according to an embodiment of the invention; Figure 3C schematically depicts a multi-pulse symbol UWB protocol according to an embodiment of the invention; Figure 3D depicts pulses from three different frequencies from a UWB transmitter according to an embodiment of the invention; Figure 3E depicts a pulse sequence of bits transmitted by a UWB transmitter according to an embodiment of the invention and its transmit power spectrum; Figure 3F depicts a pulse sequence of bits transmitted by UWB transmitters with and without biphasic phase scrambling according to an embodiment of the invention and its transmit power spectrum; Figure 4 depicts a block diagram of a UWB receiver according to an embodiment of the invention; Figure 5 depicts a receiver circuit diagram of a UWB receiver / transceiver according to an embodiment of the invention; Figure 6 depicts a circuit diagram of a UWB transceiver according to an embodiment of the invention. Figure 7 depicts a compact diversity dipole antenna according to an embodiment of the present invention; Figures 8A and 8B depict experimental results of a diversity dipole antenna designed according to Figure 7 for 3.5 GHz to 5.5 GHz; Figures 9A to 9C depict the antenna radiation patterns of the diversity dipole antenna designed as shown in Figure 7 for 3.5 GHz to 5.5 GHz for three different planes according to an embodiment of the present invention; Figures 10A and 10B depict the antenna gain and radiation efficiency of the diversity dipole antenna designed as shown in Figure 7 for 3.5 GHz to 5.5 GHz for three different planes according to an embodiment of the present invention; Figure 11 depicts the envelope correlation coefficient of the diversity dipole antenna according to an embodiment of the present invention; Figure 12 depicts a loop antenna with parasitic elements according to an embodiment of the present invention; Figures 13A to 13D depict the effects of central and lateral parasitic elements within the loop antenna depicted as shown in Figure 12 according to an exemplary embodiment of the present invention; Figure 14A depicts the gain mode changes caused by changes in the length and position of parasitic elements within the loop antenna depicted as shown in Figure 12 according to an embodiment of the present invention; Figure 14B illustrates the minimal impact of the loop antenna depicted in Figure 12, according to an embodiment of the present invention, on the ground plane; Figure 14C shows the effect of a flesh-like substance on the loop antenna depicted in Figure 12, according to an embodiment of the present invention; Figures 15A and 15B depict antennas manufactured according to embodiments of the present invention operating at 3.5-5.5 GHz and 6-9 GHz, respectively; Figures 15C and 15D depict schematic diagrams of antennas according to embodiments of the present invention operating at 3.5-5.5 GHz and 6-9 GHz, respectively, as depicted in Figures 15A and 15B;Figure 16A depicts a stub filter implemented within a ring filter according to an embodiment of the present invention; Figure 16B depicts a slotted wire filter implemented within a ring filter according to an embodiment of the present invention; Figure 16C depicts the result of a stub filter implemented within a ring filter according to an embodiment of the present invention; Figure 16D depicts the result of a slotted wire filter implemented within a ring filter according to an embodiment of the present invention; Figure 17A depicts a plan view of a rat-race hybrid balun according to an embodiment of the present invention; Figure 17B depicts a perspective view of a rat-race hybrid balun according to an embodiment of the present invention; Figure 18A depicts the return loss of a rat-race hybrid balun according to an embodiment of the present invention; Figure 18B depicts the insertion loss of a rat-race hybrid balun according to an embodiment of the present invention; Figures 19A and 19B depict the amplitude and phase balance of a rat-race hybrid balun according to an embodiment of the present invention; Figure 20 depicts the simulated output power spectrum of a UWB transmitter that exemplarily segments a frequency band into multiple sub-bands and operates on full-band, sub-band, and notch-filtered sub-bands; Figure 21 illustrates an exemplary allocation of the transmission frequency band of a UWB transmitter according to an embodiment of the present invention over time (with constant frequency band allocation or cyclic frequency band allocation); Figure 22 illustrates an exemplary allocation of the transmission frequency band of a UWB transmitter according to an embodiment of the present invention (with cyclic frequency band allocation and subsequent frequency band sequence changes); Figure 23 illustrates an exemplary allocation of the transmission frequency band of a UWB transmitter according to an embodiment of the present invention (with cyclic frequency band allocation and subsequent frequency band sequence changes); Figures 24 to 26 illustrate an exemplary spectrum sensing process and transmission frequency band allocation (with cyclic frequency band allocation and subsequent frequency band sequence changes and further spectrum sensing process) of a UWB transmitter according to an embodiment of the present invention; Figure 27 illustrates the characteristics of multipulse beams that can vary and / or jitter during transmission by a UWB transmitter according to an embodiment of the present invention; Figure 28 illustrates an exemplary variation of the pulse sequence of a single symbol multipulse beam having the same output spectrum distribution according to an embodiment of the present invention; Figure 29 illustrates a group transmitted by a UWB transmitter according to an embodiment of the present invention, with different pulse sequences used for different multipulse beams within the group. Figures 30A and 30B depict exemplary packets transmitted by a UWB transmitter according to an embodiment of the present invention, with different pulse sequences used for the multi-pulse beams within each packet; Figure 31 depicts the structure of a dual Vivaldi antenna system according to an embodiment of the present invention; Figure 32 depicts the analog S11 (dB) of the dual Vivaldi antenna system depicted in Figure 31 according to an embodiment of the present invention; Figure 33 depicts the ECC of the dual Vivaldi antenna system depicted in Figure 31 according to an embodiment of the present invention; Figures 34A to 34C depict the antenna radiation patterns of the dual Vivaldi antenna system depicted in Figure 31 according to an embodiment of the present invention in the band of interest;Figures 35A and 35B depict the peak realized gain and radiation efficiency of a dual Vivaldi antenna system according to an embodiment of the present invention; Figure 36 depicts a configuration of a UWB device utilizing a diversity antenna with dynamic antenna selection; Figure 37 depicts a configuration of a UWB receiver according to an embodiment of the present invention, utilizing parallel receiver channels with dynamic diversity antenna selection at each packet level; Figure 38 depicts an exemplary block diagram of two UWB antenna subsystems according to an embodiment of the present invention; Figure 39 depicts an exemplary layer stack of a printed circuit board (PCB) implementation of a UWB antenna subsystem employing a chip baffle according to an embodiment of the present invention; Figures 40A to 40C depict the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 39 according to an embodiment of the present invention in the band of interest; Figures 41A and 41B depict S11 and peak gain of the UWB antenna subsystem depicted in Figure 39 according to an embodiment of the present invention; Figure 42 depicts an exemplary PCB baffle according to an embodiment of the present invention; Figures 43A to 43C depict the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 42 according to an embodiment of the present invention in the band of interest; Figures 44A and 44B illustrate S11 and peak gain of the UWB antenna subsystem depicted in Figure 42 according to an embodiment of the present invention; Figure 45 depicts an exemplary filtered loop antenna module according to an embodiment of the present invention; Figures 46A and 46B illustrate S11 and radiation efficiency of the UWB antenna subsystem depicted in Figure 45 according to an embodiment of the present invention; Figures 47A to 47C depict the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 45 according to an embodiment of the present invention in the band of interest; Figure 48 depicts an exemplary differential patch antenna module according to an embodiment of the present invention; Figures 49A to 49C depict the antenna radiation pattern of the UWB antenna subsystem depicted in Figure 48 according to an embodiment of the present invention in the band of interest; and Figures 50A and 50B depict S11 and radiation efficiency of the UWB antenna subsystem depicted in Figure 48 according to an embodiment of the present invention. 4500: Filtered Loop Antenna Module 4510: First Component 4520: Second Component 4530: Third Component

Claims

1. An antenna comprising: substrate; A first antenna structure, formed on the substrate and operating within a predetermined frequency range, includes a first pattern on a first surface of the substrate and a first ground plane formed on a second surface of the substrate opposite to the first surface; a second antenna structure, formed on the substrate and operating within the predetermined frequency range, includes a second pattern on the first surface of the substrate and a second ground plane formed on the second surface of the substrate; and a first feed structure electrically coupled to the first antenna structure and a first input feed section. The first antenna structure and the second antenna structure are electrically coupled to the second antenna structure and the second input feed, wherein the first antenna structure and the second antenna structure are arranged at 90° to each other with respect to the axis of the substrate; and the first input feed and the second input feed are aligned and arranged parallel to either side of the axis of the substrate; and the first antenna structure and the second antenna structure each include: a ring structure; and a ring feed structure including different feed sections, each feed section including a pair of ports coupled to either end of the ring structure; and at least one of the following: a stub filter formed as a notch region within a first parasitic element; a first slotted line filter formed within an edge portion of the first ground plane below the differential feed section of the first antenna structure; and a second slotted line filter formed within an edge portion of the second ground plane below the differential feed section of the second antenna structure.

2. An antenna comprising: substrate; A first antenna structure, formed on the substrate and operating within a predetermined frequency range, includes a first pattern on a first surface of the substrate and a first ground plane formed on a second surface of the substrate opposite to the first surface; a second antenna structure, formed on the substrate and operating within the predetermined frequency range, includes a second pattern on the first surface of the substrate and a second ground plane formed on the second surface of the substrate; and a first feed structure electrically coupled to the first antenna structure and a first input feed section. The first antenna structure and the second antenna structure are electrically coupled to the second antenna structure and the second input feed section, wherein the first antenna structure and the second antenna structure are arranged at 90° to each other with respect to the axis of the substrate; the first input feed section and the second input feed section are aligned and arranged parallel to either side of the axis of the substrate; and each of the first antenna structure and the second antenna structure includes: a ring structure; a ring feed structure including a differential port coupled to either end of the ring structure; and a first parasitic element disposed in the ring structure. The first parasitic element is located within the center of the first antenna structure; a second parasitic element is disposed on one side of the annular structure; a third parasitic element is disposed on the other side of the annular structure; and at least one of the following: a stub filter formed as a notch region within the first parasitic element; a first slotted filter formed within the edge portion of the first ground plane below the differential feed section of the first antenna structure; and a second slotted filter formed within the edge portion of the second ground plane below the differential feed section of the second antenna structure.

3. An antenna comprising: substrate; A first antenna structure, formed on the substrate and operating within a predetermined frequency range, includes a first pattern on a first surface of the substrate and a first ground plane formed on a second surface of the substrate opposite to the first surface; a second antenna structure, formed on the substrate and operating within the predetermined frequency range, includes a second pattern on the first surface of the substrate and a second ground plane formed on the second surface of the substrate; and a first feed structure electrically coupled to the first antenna structure and a first input feed section. The first antenna structure and the second antenna structure are electrically coupled to the second antenna structure and the second input feed, wherein the first antenna structure and the second antenna structure are arranged at 90° to each other with respect to the axis of the substrate; and the first input feed and the second input feed are aligned and arranged parallel to either side of the axis of the substrate; wherein each of the first antenna structure and the second antenna structure includes: a ring structure; a ring feed structure including a differential port coupled to either end of the ring structure; a first parasitic element disposed in the center of the ring structure; a second parasitic element disposed outside the ring structure, located on one side of the ring structure; and a third parasitic element disposed outside the ring structure, located on the other side of the ring structure; the second parasitic element and the third parasitic element of the first antenna structure and the second antenna structure are respectively coupled to their respective first ground plane and second ground plane to tune the low-frequency cutoff of the antenna; and the first parasitic element is used to tune the high-frequency cutoff of the first antenna structure and the second antenna structure.

4. The antenna as claimed in claim 3 further includes a stub filter located in a notch region within the central first parasitic element.

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