Method for operating a three-phase electric machine without a transmitter

Abstract

The invention relates to a method for operating a three-phase electric machine having a stator and a rotor, in particular a synchronous motor or a reluctance machine having embedded magnets, without a transmitter by field-oriented regulation, wherein a d-component of a stator current is regulated during a first regulation and a q-component of the stator current is regulated during a second regulation, and wherein an n-phase stator voltage is determined from the d-component of the stator current and the q-component of the stator current, wherein a rotor position angle is determined from the n-phase stator voltage by means of a magnetic flux model (200), during which an integration (205, 207) of the n-phase stator voltage is performed, wherein the first regulation of the d-component of the stator current and / or the second regulation of the q-component of the stator current is performed in dependence on the determined rotor position angle, wherein at least one correction value for the integration (205, 207) is determined during at least one third regulation (259, 260) in dependence on a measured n-phase current of the three-phase electric machine, wherein the integration (205, 207) is performed in dependence on the at least one correction value.

Description

Technical Field

[0001] The present invention relates to a method for operating a three-phase motor, particularly a reluctance machine, without a transmitter by means of magnetic field-oriented adjustment, the three-phase motor having a stator and a rotor, as well as a computing unit and a computer program for performing the method. Background Technology

[0002] Three-phase motors, especially electric motors, can be used as drives for rotating shafts in various industrial applications, such as screw systems. In order to regulate speed and / or torque, it is typically necessary to know the current rotor orientation angle in such three-phase motors. While appropriate transmitters or sensors can be used for this purpose, there is an increasing pursuit of so-called transmitter-free regulation, as this eliminates the need for additional components, which often provide inaccurate values ​​based on vibrations during operation.

[0003] This transmitterless, field-oriented operation of a three-phase motor is illustrated, for example, in DE 195 31 771 B4 and EP 1037 377 A2. Summary of the Invention

[0004] According to the present invention, a method having the features of the independent claim, as well as a computing unit and a computer program for performing the method, is proposed for operating a three-phase motor, particularly a synchronous motor or reluctance machine having embedded magnets, without a transmitter via flux-oriented or field-oriented regulation (FOR), the three-phase motor having a stator and a rotor. Advantageous designs are the subject of the dependent claims and the following description.

[0005] Within the scope of this method, the d-component of the stator current is adjusted during the first adjustment period, and the q-component of the stator current is adjusted during the second adjustment period. For example, PI regulation can be set separately for this purpose. The stator voltage of the n-phase, and especially the rated value of the stator voltage of the n-phase, is determined from the d-component and the q-component of the stator current. Therefore, in particular, the rated value of the stator current is determined in the d / q coordinate system, and the rated value of the stator voltage of the n-phase is determined from this rated value of the stator current. In particular, the corresponding voltage is supplied to the stator according to the determined rated value of the stator voltage of the n-phase.

[0006] During flux-oriented or field-oriented regulation, the rotor attitude angle is determined using a flux model. Based on the determined rotor attitude angle, a first regulation of the d-component of the stator current and / or a second regulation of the q-component of the stator current are performed.

[0007] The d / q coordinate system provides the possibility of representing the parameters of an n-phase three-phase motor using mutually perpendicular axes d and q in a two-dimensional coordinate system with a fixed relative rotating magnetic field. The coordinate system utilizes the angular frequency Ω. 转子 It rotates together with the rotor. The n-dimensional coordinate system can be converted to a d / q coordinate system through a so-called d / q transformation or Park transformation. For the d / q transformation, the rotor attitude angle θ between the stator magnetic field and the rotor magnetic field is important so that the d / q coordinate system can rotate with the rotor at the correct angular velocity and phase.

[0008] In this method, the rotor attitude angle is determined using a flux model based on the stator voltage of the n-phase phases, specifically based on the rated values ​​of the n-phase stator voltages. The integration of the n-phase stator voltages, specifically the rated values ​​of the n-phase stator voltages, is performed during the flux modeling process. The rated values ​​of the n-phase stator voltages and the stator currents of the three-phase motor, specifically the measured n-phase currents, are provided as input parameters to the flux model. As output parameters, the flux model appropriately provides the rotor attitude angle, and furthermore, specifically, the rotational speed of the three-phase motor.

[0009] The rotor attitude angle can be determined using a flux model, particularly from the rotor flux, in which the rotor flux is determined by the current flowing in the stator (or its windings or phases), the voltage present in the stator (or its windings or phases), the ohmic resistance of the stator (or its windings or phases), and the inductance of the stator. The flux model can be implemented as a so-called observer in regulation techniques.

[0010] In the flux model, the rotor flux can be appropriately obtained by integrating with respect to the stator voltage, subtracting the product of the stator current and the stator's ohmic resistance from the stator voltage. From this integral value, in particular, the product of the stator inductance and the stator current is also subtracted. Using the arctangent function, the rotor attitude angle can be obtained from the rotor flux obtained in this way. Furthermore, the time derivative of the rotor attitude angle, divided by the number of pole pairs, yields the rotor's angular velocity, which corresponds to the rotational speed of the three-phase motor.

[0011] Within the scope of this invention, at least one correction value for integration is determined during at least one third adjustment period, and integration is performed based on this at least one correction value. The at least one correction value is determined during at least one third adjustment period based on the measured stator current of the n-phase of the three-phase motor. In particular, the determined at least one correction value is subtracted from the stator voltage before integration. Suitably, the product of the stator current and the stator's ohmic resistance can also be subtracted from the stator voltage before integration.

[0012] Therefore, within the scope of this invention, a transmitterless regulation of a three-phase motor is proposed, which is achieved through flux symmetry or symmetry of the integral flux component. Symmetry of the integral flux component occurs by comparing the integral flux component with a flux signal, which is alternatively calculated substantially from the current (or voltage). For example, a correction voltage for the flux integrator can be derived from this difference using a PI regulator.

[0013] The flux model can be improved by adjusting the values, and simple optimization of transmitter-free regulation can be achieved. Integral drift can be prevented by angle correction, and dynamic angle correction is particularly possible. Flux errors can be avoided even at very low speeds. Three-phase motors can be regulated with high efficiency without a transmitter, especially at very low speeds.

[0014] The correction value is suitably determined here using parameters that are already known or exist. It is suitable that additional parameters or parameters need to be measured using measurement techniques to determine the correction value, which could affect the regulation dynamics. Therefore, in particular, no additional expense is required for the measuring device. Especially, the correction value is determined in a suitable manner even without measurements of the stator voltage.

[0015] Furthermore, besides the d / q coordinate system, there is a possibility of transforming an n-dimensional coordinate system to an α / β coordinate system that is relatively stationary relative to the stator via the so-called Clarke transformation. In the Clarke transformation, a basic Cartesian coordinate system is chosen, similar to that of a stationary stator, and is depicted in the complex plane with real part α and imaginary part β. The axes of the n-dimensional coordinate system, usually U, coincide with the real axis α. The magnetic flux model is calculated, specifically in the α / β coordinate system.

[0016] Advantageously, a first integral of the α component and a second integral of the β component of the stator voltage are performed in the flux model. For this purpose, a Clarke conversion of the n-phase stator voltage, particularly a Clarke conversion of the stator voltage rating, is performed at the input of the flux model. Advantageously, a first correction value for the integral of the α component of the stator voltage and a second correction value for the integral of the β component of the stator voltage are determined during at least one third adjustment period. Suitably, PI regulation is performed separately to determine the correction values.

[0017] Suitablely, the first correction value is subtracted from the α component of the stator voltage before the first integration. Furthermore, in particular, the product of the α component of the stator current and the ohmic resistance of the stator can be subtracted from the α component of the stator voltage before the first integration.

[0018] Similarly, the second correction value is suitably subtracted from the β component of the stator voltage before the second integration. Furthermore, in particular, the product of the β component of the stator current and the ohmic resistance of the stator is subtracted from the β component of the stator voltage before the second integration.

[0019] Advantageously, the difference between the product of the integrated stator voltage and stator inductance of the n-phase phase and the measured stator current of the n-phase phase is determined. Preferably, at least one correction value for the integral is determined based on this difference during at least one third adjustment period. In particular, at least one correction value can be determined from this difference using a PI regulator.

[0020] In particular, the measured n-phase stator current at the input of the magnetic flux model is expressed in the α / β coordinate system using Clarke transformation. In this way, the α and β components of the stator current can be determined, multiplied by the stator's ohmic resistance, and the product can be subtracted from the α and β components of the stator voltage, respectively, before integration. Subsequently, the α and β components of the stator current are transformed to the d / q coordinate system.

[0021] Suitablely, the d-component of the stator current is multiplied by the difference between the d-component and the q-component of the inductance. The resulting product in the d / q coordinates is then converted to α / β coordinates. Therefore, the α-component of the product is determined, which is suitable for forming the difference with the integral α-component of the stator voltage. Correspondingly, the β-component of the product is used to form the difference with the integral β-component of the stator voltage.

[0022] Advantageously, at least one correction value for the integral is determined during at least one third adjustment based on the magnetic flux, particularly based on the rotor magnetic flux of the permanent magnet. In particular, this possibility applies to permanent magnet-excited synchronous motors having permanent magnets (IPMs) embedded below the surface of the rotor.

[0023] Preferably, the product of the stator inductance and at least one component of the measured stator current of the n-phase is determined; particularly preferably, the product of the d-component of the stator current and the difference between the d-component and the q-component of the inductance is determined. Preferably, magnetic flux is added to this product, and the difference between the integrated stator voltage of the n-phase and this sum is determined. Particularly preferably, the difference between the α-component of the integrated stator voltage and this sum is determined. During at least one third adjustment period, at least one correction value for the integral is advantageously determined based on this difference, particularly a first correction value for the first integral of the α-component of the stator voltage.

[0024] The correction value is determined based on the d / q coordinates of the inductor. This method is particularly suitable for synchronous reluctance machines (SynRM) and permanent magnet driven synchronous motors with magnets embedded below the surface of the rotor (IPM).

[0025] According to an advantageous embodiment, the arctangent function is used for the integration of the n-phase stator voltage, and at least one correction value for the integration is determined based on the components of the arctangent function. The result of the arctangent function is particularly concerned with a first component of absolute value and a second component of angle. The rotor attitude angle can be obtained, in particular, from the second component. Preferably, at least one correction value is determined based on the rotor attitude angle determined in this manner during at least one third adjustment.

[0026] Three-phase motors can be used, for example, to drive machines. Such machines can be configured as machine tools, such as welding systems, screw systems, wire saws, or milling machines, or as web processing machines, such as printers, newspaper printers, gravure printers, screen printers, inline flexographic printers, or packaging machines. The machine can also be configured as a (belt) device for manufacturing automobiles or for manufacturing automobile parts (such as internal combustion engines or control devices).

[0027] Furthermore, three-phase motors can also be used in vehicles, for example, where they can operate suitably as either generators or motors. In generator operation, the three-phase motor can absorb driving torque and convert mechanical energy into electrical energy. In motor operation, the three-phase motor can convert electrical energy into mechanical energy and generate driving torque.

[0028] The computing unit, such as the control device, according to the invention is configured, particularly in terms of programming technology, to execute the method according to the invention.

[0029] Implementing this method as a computer program is also advantageous because it results in particularly low costs, especially when the control device being implemented is also used for other tasks and therefore exists anyway. Suitable data carriers for providing the computer program are particularly magnetic, optical, and electrical storage devices, such as hard disks, flash memory, EEPROM, DVDs, etc. Downloading the program via computer networks (Internet, intranet, etc.) is also possible.

[0030] Further advantages and design solutions of the present invention are available from the specification and drawings.

[0031] It should be understood that the features mentioned above and those to be described below may be used not only in the combinations described separately, but also in other combinations or individually, without departing from the scope of protection of this invention. Attached Figure Description

[0032] The present invention is schematically illustrated in the accompanying drawings with reference to embodiments, and will subsequently be described in detail with reference to the drawings. Wherein:

[0033] Figure 1The illustration schematically shows a preferred embodiment of the method according to the invention for operating a three-phase motor without a transmitter when using a magnetic flux model as a block circuit diagram;

[0034] Figure 2 The magnetic flux model of a preferred embodiment of the method according to the invention is illustrated as a block circuit diagram. Detailed Implementation

[0035] Figure 1 The illustration schematically shows a preferred embodiment of the method according to the invention for operating a three-phase motor without a transmitter when using a magnetic flux model as a block circuit diagram.

[0036] The three-phase motor 150 can be configured, for example, as an asynchronous motor (ASM) or a permanent magnet synchronous motor (PMSM; SPM, IPM) or preferably as a synchronous reluctance motor (SynRM).

[0037] During this method, a preset value I for the d-component of the stator current is used. d *As input value 101, and preset the rated speed n for the three-phase motor 150 as input value 106.

[0038] In comparison position 102, the actual value I of the d-component of the stator current. d From the rated value i d *Subtracted, the actual value is determined using magnetic flux model 200 and measurement 115, as further explained below. Here, for example, in a regulator constructed as a PI regulator 103, the d-component of the stator current is adjusted during the first adjustment period, where the d-component U of the stator voltage is obtained as an output parameter. d It is fed to the input terminal 105 of the calculation unit 104, and the calculation unit performs the conversion from dq coordinates to UVW coordinates.

[0039] Furthermore, in comparison position 107, the actual value of the rotational speed is subtracted from the rated value n*, which is determined in the flux model 200 based on the rotor attitude angle θ, as further explained below. Here, speed regulation is performed, for example, in a regulator configured as a PI regulator 108, where the q component I for the stator current is obtained as an output parameter. q The nominal value. In comparison position 109, the actual value of the q component is subtracted from this nominal value, which is determined by means of magnetic flux model 200 and measurement 115.

[0040] In this example, in a regulator constructed as a PI regulator 110, the q-component of the stator current is adjusted during the second adjustment period, wherein the q-component U of the stator voltage is obtained as an adjustment parameter. qIt is transmitted to the input terminal 111 of the computing unit 104.

[0041] Furthermore, the actual value of the rotor attitude angle θ, determined in the flux model 200, is fed to the input terminal 112 of the calculation unit 104. At the output terminal 113, the calculation unit 104 provides the rated value V for the stator voltage of the three-phase motor 150. UVW The rated value is supplied to the power or regulating electronics 114 for the three-phase motor 150, so that the voltage is adjusted according to the rated value V. UVW exist.

[0042] The stator current I of phase n was measured during measurement 115. UVW The measured n-phase current is fed to input terminal 116 of the magnetic flux model 200. Furthermore, the stator voltage rating V is output by the calculation unit 104. UVW The flux is fed to the input terminal 117 of the flux model 200. At the output terminal 118, the flux model 200 provides the rotational speed of the three-phase motor 150, and at the output terminal 119, it provides the rotor attitude angle θ.

[0043] The rotor attitude angle θ is also fed to the input terminal 121 of the calculation unit 120, and the stator current I is measured at the input terminal 122. UVW The data is fed to the calculation unit. The calculation unit 120 performs d-q conversion and provides the actual value of the d component of the measured stator current at output 123 and the actual value of the q component of the measured stator current at output 124. These actual values ​​are fed to comparison positions 102 and 109.

[0044] Therefore, the first adjustment 103 of the d component of the stator current and the second adjustment 110 of the q component of the stator current are performed according to the determined rotor posture angle θ.

[0045] Figure 2 A block diagram schematically illustrates a magnetic flux model 200 of a preferred embodiment of the method according to the invention, which is particularly advantageous for synchronous motors and synchronous reluctance machines (SynRMs) with concealed magnets (IPMs).

[0046] As described above, the stator current I being measured at input terminal 116 is... UVW The flux is supplied to the magnetic flux model 200, and the stator voltage rating V is input to terminal 117. UVW These values ​​are fed into the magnetic flux model. They are sampled periodically, in particular.

[0047] The Clarke transformation is performed in calculation unit 201 to represent the stator voltage V in α / β coordinates. UVWAt the output terminal 202 of the calculation unit 201, the α component V for the stator voltage is provided as the first output parameter. α The rated value, and at output terminal 203, provides the β component V of the stator voltage as a second output parameter. β The rated value.

[0048] Furthermore, a Clarke transformation is performed in the calculation unit 220 to represent the stator current I in α / β coordinates. UVW At output 221, the calculation unit 220 provides the α component I for the stator current. α The actual value is used as the first output parameter, and the β component I for the stator current is provided at the output terminal 222. β The actual value is used as the second output parameter.

[0049] α component I of stator current α It is fed to the multiplication position 223, and there it is connected to the ohmic resistance R of the stator. S Multiplication. The product is fed to comparison position 204, where it is compared with the α component V of the stator voltage. α Subtract. Also consider the correction value from box 250, as further explained below.

[0050] In multiplication position 224, the β component I of the stator current β Ohmic resistance R of the stator S Multiply, and the product is taken from the β component V of the stator voltage at comparison position 206. β Subtract. Also consider the correction value from box 250, as further explained below.

[0051] The first integration of the comparison result of the correction of the α component of the stator voltage is performed in the first integrator 205, and the second integration of the comparison result of the correction of the β component of the stator voltage is performed in the second integrator 207.

[0052] The α component I provided by the stator current at the output terminal 221 α Furthermore, it is transported to the multiplication position 240, and there it is coupled with the q component L of the stator inductance. q Multiply. The product is subtracted from the α component of the integral from the stator voltage at comparison position 241.

[0053] The β component I provided by the stator current at the output terminal 222 β In the multiplication position 242, the q component L of the stator inductance is... q Multiply. In comparison position 243, the product is subtracted from the β component of the integral of the stator voltage.

[0054] In calculation unit 208, the arctangent function is used for the α component of the integral of the stator voltage and the β component of one of the integrals of the stator voltage. The component with respect to angle is provided at output terminal 210.

[0055] The angular component is provided at output 119 as the rotor attitude angle θ. In calculation unit 213, the rotor angular velocity, corresponding to the rotational speed of the three-phase motor 150, is determined by the time derivative of the rotor attitude angle θ and provided at output 118.

[0056] Within the scope of the invention, in block 250, at least one correction value for integrals 205 and 207 is determined during at least one third adjustment 259, 260, and integrals 205 and 207 are performed based on the at least one correction value.

[0057] At least one correction value is based on the measured n-phase stator current I of the three-phase motor 150 provided at input terminal 116. UVW And / or based on the stator voltage V of the three-phase motor 150 UVW The rated value provided at input terminal 117 is determined.

[0058] For this purpose, the α component I of the stator current α The β component I of the stator current is transmitted to the input terminal 226 of the computing unit 225. β The rotor attitude angle θ is fed to the calculation unit 225 at input terminal 227. In addition, the rotor attitude angle θ is fed to the calculation unit 225 at input terminal 228.

[0059] The calculation unit 225 performs the d-q conversion of the stator current and provides the d-component I of the stator current at the output terminal 229. d And the q component I of the stator current is provided at the output terminal 230. q .

[0060] d-component I of stator current d In the multiplication position 231, the d-component L of the stator inductance... d and q component L q The difference (L) d -L q Multiply by 1.

[0061] In a permanent magnet synchronous motor having a permanent magnet (IPM) embedded beneath the surface of the rotor, at position 232, the magnetic flux ψ M It is added to the product formed at multiplication position 231. Magnetic flux ψ M In particular, the rotor flux of the permanent magnet is determined in box 233.

[0062] The sum is sent to the calculation unit 251 at input terminal 252. The value 0 is sent to the input terminal 253 of the calculation unit 251 by unit 234. In addition, the rotor attitude angle θ is sent to the calculation unit 251 at input terminal 254.

[0063] The calculation unit 251 then performs the α / β coordinate transformation and provides the α component at the output terminal 255 and the β component of the product of the stator current and the stator inductance at the output terminal 256.

[0064] In comparison position 257, the α component of the integral of the stator voltage is subtracted from the α component of the product. This difference is adjusted in a regulator, for example configured as a PI regulator 259, to determine a first correction value for the first integral 205. This correction value is subtracted from the α component of the stator voltage in comparison position 204. Therefore, the first integral 205 is performed based on this first correction value.

[0065] In comparison position 258, the β component of the stator voltage integral is subtracted from the β component of the product. This difference is adjusted in a regulator, for example configured as a PI regulator 260, to determine a second correction value for the second integral 207. The correction value is subtracted from the β component of the stator voltage in comparison position 206, and the second integral 207 is performed based on the second correction value.

[0066] The flux model can be improved by adjusting the values, and simple optimization of transmitterless regulation of the three-phase motor 150 can be achieved. Drift of integrals 205 and 207 can be prevented by adjusting the values, and dynamic angle correction can be achieved in particular. Flux errors can be avoided even at very low speeds. The three-phase motor 150 can be regulated with high efficiency without a transmitter, especially at very low speeds.

[0067] The correction value is determined using parameters that are already known or exist, in particular based on the measured stator current I of the n-phase motor 150 provided at input terminal 116. UVW And based on the rated value V provided at input terminal 117 of the stator voltage of the three-phase motor 150. UVW Therefore, it is particularly unnecessary to detect additional parameters or parameters using measurement techniques to determine correction values, especially for stator voltage measurements, which could affect regulation dynamics. Thus, additional costs associated with measurement devices are also unnecessary.

[0068] exist Figure 2The diagram schematically illustrates a magnetic flux model 200b according to another preferred embodiment of the method according to the invention, which is particularly suitable for permanent magnet excitation synchronous motors having permanent magnets (IPMs) embedded below the surface of the rotor.

Claims

1. A method for operating a three-phase motor (150) without a transmitter by means of magnetic field-oriented adjustment, said three-phase motor having a stator and a rotor, in, The d-component of the stator current is adjusted during the first adjustment (103), and the q-component of the stator current is adjusted during the second adjustment (110), wherein the stator voltage of the n-phase is determined from the d-component and the q-component of the stator current (104, 114). The rotor attitude angle is determined by means of the magnetic flux model (200) based on the stator voltage of the n phases, and the integration of the stator voltage of the n phases (205, 207) is performed during this period. The first adjustment (103) of the d-component of the stator current and / or the second adjustment (110) of the q-component of the stator current are performed according to the determined rotor attitude angle. The invention is characterized in that, during at least one third adjustment (259, 260), at least one correction value for the integral (205, 207) is determined based on the measured stator current of the n-phase of the three-phase motor, wherein the integral (205, 207) is performed based on the at least one correction value, wherein, during the at least one third adjustment (259, 260), at least one correction value for the integral (205, 207) is determined based on the rotor flux of the permanent magnet.

2. The method according to claim 1, wherein, In the magnetic flux model (200), a first integral (205) of the α component of the stator voltage and a second integral (207) of the β component of the stator voltage are performed, wherein a first correction value for the integral (205) of the stator voltage and a second correction value for the integral (207) of the β component of the stator voltage are determined during the at least one third adjustment (259, 260).

3. The method according to claim 1 or 2, wherein, Determine (231) the product of the stator inductance and the measured stator current of the n-phase, wherein (257, 258) the difference formed by the integrated stator voltage of the n-phase and the product is determined, and wherein at least one correction value for the integration (205, 207) is determined based on the difference during the at least one third adjustment (259, 260).

4. The method according to claim 1 or 2, wherein, This method is used to operate a synchronous motor or magnetoresistive machine with embedded magnets without a transmitter by adjusting the magnetic field orientation.

5. The method according to claim 4, wherein, Determine (231) the product of the stator inductance and at least one component of the measured stator current of the n-phase, wherein the magnetic flux is added (232, 233) to the product, wherein determine (257) the difference formed by the integrated stator voltage of the n-phase and the sum, and wherein during the at least one third adjustment (259), at least one correction value for the integral (205) is determined based on the difference.

6. The method according to claim 1 or 2, wherein, The d-component and q-component of the measured n-phase stator current are determined (225, 229, 230), wherein the α-component and β-component of the measured n-phase stator current are determined (251, 255, 256) based on the d-component and the rotor attitude angle, and wherein at least one correction value is determined based on the α-component and β-component of the measured n-phase stator current during the at least one third adjustment (259, 260).

7. The method according to claim 1 or 2, wherein, The arctangent function is used for the integrated n-phase stator voltage (208), and during the at least one third adjustment (259, 260), at least one correction value for the integral (205, 207) is determined based on the angle-related component of the arctangent function.

8. A control device configured to perform the method according to any one of the preceding claims.

9. A computer program product comprising a computer program that, when implemented on a computing unit, causes the computing unit to perform the method according to any one of claims 1 to 7.

10. A machine-readable storage medium having a computer program stored thereon, wherein when the computer program is implemented on a computing unit, the computer program causes the computing unit to perform the method according to any one of claims 1 to 7.

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