An isolated power converter

Abstract

This invention discloses an isolated power converter. In this embodiment, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit. The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit controls the switching state of the secondary switching component in the secondary switching network based on an output feedback signal characterizing the output signal to achieve secondary leakage inductance excitation and freewheeling. Therefore, this embodiment can directly adjust the secondary switching component in the secondary switching network so that the secondary leakage inductance can act as a power storage element, achieving magnetic integration of the transformer. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs.

Description

Technical Field

[0001] This invention relates to the field of electronic power technology, and more specifically, to an isolated power converter. Background Technology

[0002] High-frequency transformers are indispensable in converter applications requiring electrical isolation. Traditional high-frequency transformers, due to their high-permeability cores that confine the magnetic field lines of the windings, have relatively high coupling coefficients (k>0.9) and low leakage inductance between the primary and secondary windings. However, in certain applications, such as wireless charging or high-power-density modular power supplies, the physical positions of the primary and secondary windings are relatively variable, or losses at ultra-high switching frequencies are unacceptable due to the core material. Therefore, the method of adding a core element to the primary and secondary windings to achieve a high coupling coefficient is no longer viable. Removing the core results in a loosely coupled transformer, significantly reducing the coupling coefficient (k<0.5). At this point, the leakage inductance between the primary and secondary windings increases dramatically, reaching orders of magnitude equal to the transformer's magnetizing inductance. Increased leakage inductance can cause serious problems for power electronic converters. On the one hand, it may cause an increase in the voltage stress of components, and on the other hand, it may deteriorate the output characteristics of the converter and reduce the conversion efficiency. Therefore, it is necessary to eliminate or compensate for it at the circuit and control levels. Summary of the Invention

[0003] In view of this, embodiments of the present invention provide an isolated power converter that achieves magnetic integration of the transformer by directly adjusting the secondary-side switching components so that the secondary-side leakage inductance can be used as a power storage element. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs.

[0004] In a first aspect, embodiments of the present invention provide an isolated power converter, the isolated power converter comprising:

[0005] The primary circuit is configured to generate an AC input signal;

[0006] A transformer, including a primary winding and a secondary winding, is configured to process the AC input signal to generate a secondary input signal;

[0007] The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit is configured to convert the secondary input signal to generate an output signal. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit is configured to control the switching state of the secondary switching component in the secondary switching network according to an output feedback signal characterizing the output signal.

[0008] In this embodiment of the invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit. The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit controls the switching state of the secondary switching component in the secondary switching network based on the output feedback signal characterizing the output signal to achieve secondary leakage inductance excitation and freewheeling. Therefore, this embodiment can directly adjust the secondary switching component in the secondary switching network so that the secondary leakage inductance can act as a power storage element, achieving magnetic integration of the transformer. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs. Attached Figure Description

[0009] The above and other objects, features and advantages of the present invention will become clearer from the following description of embodiments of the invention with reference to the accompanying drawings, in which:

[0010] Figure 1 This is a circuit diagram of an isolated power converter according to an embodiment of the present invention;

[0011] Figure 2 This is a circuit diagram of the secondary circuit of Embodiment 1 of the present invention;

[0012] Figure 3 This is a circuit diagram of the secondary circuit of Embodiment 2 of the present invention;

[0013] Figure 4 This is a schematic diagram of the VI curve characteristics of the secondary-side switch assembly according to an embodiment of the present invention;

[0014] Figure 5 This is a schematic diagram of the secondary-side switch assembly according to an embodiment of the present invention;

[0015] Figure 6 This is a schematic diagram of the drive signal of the secondary-side switch assembly in Embodiment 2 of the present invention;

[0016] Figures 7-10 This is an equivalent circuit diagram of the secondary circuit in Embodiment 2 of the present invention;

[0017] Figure 11 This is a circuit diagram of the secondary circuit of Embodiment 3 of the present invention;

[0018] Figure 12 This is a schematic diagram of the drive signal of the secondary-side switch assembly according to Embodiment 3 of the present invention;

[0019] Figures 13-14 This is an equivalent circuit diagram of the secondary circuit in Embodiment 2 of the present invention;

[0020] Figure 15This is a schematic diagram of the driving signal for another secondary-side switch assembly according to Embodiment 3 of the present invention;

[0021] Figure 16 This is a flowchart of the secondary-side voltage regulation control method according to an embodiment of the present invention. Detailed Implementation

[0022] The present invention is described below based on embodiments, but the invention is not limited to these embodiments. In the detailed description of the invention below, certain specific details are described in detail. Those skilled in the art will fully understand the invention even without these details. To avoid obscuring the essence of the invention, well-known methods, processes, flows, elements, and circuits are not described in detail.

[0023] Furthermore, those skilled in the art should understand that the accompanying drawings provided herein are for illustrative purposes only and are not necessarily drawn to scale.

[0024] Furthermore, it should be understood that in the following description, "circuit" refers to a conductive loop consisting of at least one element or sub-circuit connected by electrical or electromagnetic connections. When an element or circuit is said to be "connected" to another element or "connected" between two nodes, it can be directly coupled or connected to another element, or there may be intermediate elements. The connection between elements can be physical, logical, or a combination thereof. Conversely, when an element is said to be "directly coupled to" or "directly connected" to another element, it means that there are no intermediate elements between them.

[0025] Unless the context explicitly requires it, words such as "including" or "contains" in the instruction manual should be interpreted as including rather than exclusive or exhaustive; that is, meaning "including but not limited to".

[0026] In the description of this invention, it should be understood that the terms "first," "second," etc., are used for descriptive purposes only and should not be construed as indicating or implying relative importance. Furthermore, in the description of this invention, unless otherwise stated, "a plurality of" means two or more.

[0027] Existing leakage inductance compensation methods typically involve adding capacitors to both the primary and secondary sides to construct two independent LC resonant cavities. The impedance of these two cavities is then matched or mismatched by adjusting the converter switching frequency (PFM) to modulate the output energy. However, for applications with a wide input / output range, this method requires a wide range of switching frequency adjustments in pure PFM control mode to achieve voltage regulation. While combining PFM with pulse width modulation (PWM) can reduce the switching frequency adjustment range, it may cause the soft-switching characteristics of the switching transistor to be lost in certain situations, adversely affecting the switching device. Furthermore, the two additional resonant capacitors added to the primary and secondary sides are large in size because they serve as main power storage capacitors, which negatively impacts power density improvement to some extent.

[0028] Figure 1 This is a circuit diagram of an isolated power converter according to an embodiment of the present invention. In this embodiment, the isolated power converter 1 includes a primary circuit 11, a transformer 12, and a secondary circuit 13. The primary circuit 11 is configured to generate an AC input signal Ui. Optionally, the primary circuit 11 includes a primary impedance network 111. The primary impedance network 111 is an impedance network with the primary leakage inductance of the transformer. Optionally, the primary impedance network 111 may also include a resonant capacitor added to compensate for or adjust the overall impedance of the transformer. In other optional implementations, the primary impedance network 111 may not include an additional resonant capacitor to further reduce circuit complexity and cost. The primary and secondary windings of the transformer 12 can be an ideal transformer circuit, including only the portion where the primary and secondary linkage magnetic fluxes are fully coupled.

[0029] In this embodiment, the isolated power converter 1 is divided into the primary and secondary sides of the transformer 12. The primary inverter part of the isolated power converter 1 is used as a half-stage, and all the power devices on the secondary side of the power converter 1 form a complete converter. At this time, the leakage inductance of the secondary side of the transformer is functionalized as a power energy storage element of the subsequent converter, thus realizing the magnetic integration of the transformer.

[0030] In one alternative implementation, the primary-side circuit 11 may include an AC source to provide AC input. Optionally, the AC source may be a voltage source or a current source, and this embodiment is not limited to this.

[0031] In another optional implementation, the primary-side circuit 11 may include a DC input source and a DC-AC inverter network formed by the primary-side switching network 112. This DC-AC inverter network is used to convert the DC input source into an AC source. Optionally, the DC-AC inverter network can be configured as a forward converter, flyback converter, full-bridge, half-bridge, or push-pull structure, etc. It should be understood that this embodiment does not limit the structure of the DC-AC inverter network, as long as it can convert the DC source into an AC source. It should also be understood that this embodiment does not limit the AC input of the AC source; that is, the primary-side input of the isolated power converter 1 can be any AC waveform with unlimited amplitude and shape, and the pulse widths of the positive and negative half-cycles do not need to be equal. Since this invention uses closed-loop control of the secondary-side circuit, the primary-side switching network 112 in the primary-side circuit 11 can be controlled in an open-loop manner with a fixed duty cycle or a fixed frequency, without closed-loop modulation.

[0032] The secondary circuit 13 includes a buck converter circuit 131 and a feedback control circuit 132. The converter circuit 131 is configured to perform buck converter processing on the received secondary input signal Ui to generate an output signal Uo. It should be understood that a buck converter means the output signal is always less than the input signal.

[0033] In this embodiment, the buck converter circuit 131 is formed by the secondary-side switching network 131a and the secondary leakage inductance Lks of the secondary winding of the transformer 12. The buck converter circuit 131 achieves current conversion by energizing the secondary leakage inductance Lks of the secondary winding via the secondary-side switching components of the secondary-side switching network 131a. The feedback control circuit 132 is configured to control the switching state of the secondary-side switching components in the secondary-side switching network 131a according to the output feedback signal characterizing the output signal Uo, thereby adjusting the output signal Uo to tend towards a desired value. Therefore, the isolated power converter 1 in this embodiment can realize closed-loop control of the secondary side by utilizing the secondary side leakage inductance in the secondary side circuit. That is, the switching state of the secondary side switching components in the secondary side switching network is adjusted by negative feedback in the secondary side circuit to achieve secondary side self-regulation of voltage or current. At the same time, the primary side switching network in the primary side circuit can be open-loop controlled with a fixed duty cycle or fixed frequency. This enables the output signal to be adjusted without primary-secondary communication, thereby achieving primary-secondary decoupling and reducing the complexity and cost of the circuit structure.

[0034] In one optional implementation, this embodiment employs different frequencies with varying primary and secondary energy transfer times to control the primary-secondary decoupling strategy. Specifically, the primary-side switching frequency of the isolated power converter 1 in this embodiment is no greater than the secondary-side switching frequency, and the duty cycle of the primary-side switching components is independent of the duty cycle of the secondary-side switching components. Therefore, by setting the secondary-side switching frequency to be no less than the primary-side switching frequency, this embodiment achieves a difference in energy transfer times between the primary and secondary sides, enabling self-regulation of voltage and current on the secondary side. Simultaneously, the switching components in the primary-side circuit can be open-loop controlled with a fixed duty cycle or a fixed frequency, eliminating the need for feedback signal paths between the primary and secondary sides, thus achieving primary-secondary decoupling.

[0035] This embodiment adjusts the switching state of the secondary-side switching components based on negative feedback of the output voltage of the isolated power converter to achieve voltage or current regulation. In other words, this embodiment implements self-regulating voltage / current regulation on the secondary side of the isolated power converter without adjusting the switching frequency and duty cycle of the primary-side switching components, thus eliminating the need for feedback signal paths between the primary and secondary sides. This improves power density and reduces system control complexity and cost.

[0036] This embodiment primarily describes the use of PWM mode to control and adjust the duty cycle of the secondary-side switching component, taking the stabilization of the output voltage as an example. It should be understood that this embodiment does not limit the adjustment method of the secondary-side switching component.

[0037] Figure 2 This is a circuit diagram of the secondary circuit according to Embodiment 1 of the present invention. Figure 2 As shown, the secondary circuit 2 of the isolated power converter in this embodiment includes a buck converter circuit 21 and a feedback control circuit 22 formed by the secondary leakage inductance Lks and the secondary switching network 211. The secondary switching network 211 includes at least one secondary switching component. In this embodiment, at least one secondary switching component is controlled by a corresponding drive signal to adjust its switching state, thereby regulating the output voltage and achieving voltage / current regulation. The drive signal for the secondary switching component is determined based on the error signal corresponding to the output voltage. Optionally, the switching frequency of the secondary switching component is not less than the switching frequency of the primary switching component, resulting in a difference in energy transfer time between the primary and secondary sides. This achieves self-regulation of voltage / current on the secondary side, while eliminating the need for feedback signal paths between the primary and secondary sides, thus achieving decoupling between the primary and secondary sides.

[0038] In this embodiment, the feedback control circuit 22 is configured to control the switching state of the secondary-side switching components in the secondary-side switching network 211 according to the output feedback signal characterizing the output signal. Further, the feedback control circuit 22 adjusts the duty cycle of each secondary-side switching component based on the error between the output feedback signal and the reference signal, so that the output signal tends towards the desired value. In an optional implementation, the feedback control circuit 22 further includes an error acquisition circuit 221 and a driving circuit 222. The error acquisition circuit 221 is configured to acquire an error signal based on the error between the output feedback signal and the reference signal of the output voltage. Optionally, the error acquisition circuit 221 is used to acquire the output feedback signal Vf and the reference signal Vref sampled from the output terminal, and acquire the error signal Ve based on the difference between the output feedback signal Vf and the reference signal Vref. Optionally, in this embodiment, the output feedback signal Vf is acquired by sampling the voltage across the output equivalent resistance Ro. Further optionally, the secondary-side circuit 2 also includes an output capacitor Co connected in parallel with the output equivalent resistance Ro to further stabilize the output voltage, current, and other electrical signals. In this embodiment, the reference signal Vref can be the expected value of the output signal, or it can be other signals that can characterize the expected value of the output signal.

[0039] In one optional implementation, the drive signal 222 is configured to generate a PWM signal based on the error signal Ve, thereby determining the drive signal for each secondary-side switching component to control the switching state of the corresponding secondary-side switching component. Thus, this embodiment can adjust the output voltage based on negative feedback of the output voltage to achieve voltage / current regulation. Further optionally, the drive circuit 222 generates a PWM signal based on the error signal Ve and the ramp signal RAMP, thereby determining the drive signal for each secondary-side switching component. It should be understood that this embodiment is not limited to the ramp signal; other signals, such as a wave signal, can be used in the acquisition of the drive signal in this embodiment, and this embodiment does not impose any limitations on this. It should be understood that when the secondary-side switching component is controlled by the PWM signal, it is under high-frequency chopping control, and the switching frequency is determined by the frequency of the ramp signal.

[0040] In one alternative implementation, the buck converter circuit 21 can be either a buck full-wave rectifier circuit or a buck half-wave rectifier circuit. This embodiment does not limit the buck converter circuit, as long as it can achieve self-stabilization of the output electrical signal through buck conversion.

[0041] In one alternative implementation, if the switching frequency of the secondary-side switching component in the secondary-side switching network is an integer multiple of the switching frequency of the primary-side switching component, the ramp signal RAMP can be synchronized by detecting the transition edge (e.g., rising edge) of the secondary-side input signal, thereby keeping it synchronized with the switching frequency of the primary side, in order to further improve signal conditioning efficiency and reduce losses.

[0042] In one alternative implementation, during the operating cycle of the secondary-side switching network, at least one of the secondary-side switching components is chopper-controlled to magnetize the secondary-side leakage current, thereby achieving step-down conversion. Specifically, the feedback control circuit 22 is configured to chopper-control the corresponding control switching component during the positive or negative half-cycle of the secondary-side input signal, causing the secondary-side leakage current to magnetize, thus achieving step-down conversion and adjusting the output signal to tend towards the desired value, thereby achieving secondary-side self-regulation or current regulation.

[0043] In this embodiment of the invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit. The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit controls the switching state of the secondary switching component in the secondary switching network based on the output feedback signal characterizing the output signal to achieve secondary leakage inductance excitation and freewheeling. Therefore, this embodiment can directly adjust the secondary switching component in the secondary switching network so that the secondary leakage inductance can act as a power storage element, achieving magnetic integration of the transformer. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs.

[0044] Figure 3 This is a circuit diagram of the secondary-side circuit according to Embodiment 2 of the present invention. In this embodiment, the buck converter circuit is a buck full-wave rectifier circuit. Optionally, the switching frequency of the secondary-side switching component is not less than twice the switching frequency of the primary-side switching component, so as to make the energy transfer time of the primary and secondary sides different. Figure 3 As shown, in this embodiment, the secondary circuit 3 includes a step-down converter circuit formed by the secondary leakage inductance Lks and the secondary switching network. The secondary switching network includes secondary switching components S1-S4. Specifically, secondary switching component S1 is coupled between the first input terminal i1 of the secondary switching network and the first output terminal o1 of the secondary circuit 3; secondary switching component S2 is coupled between the first input terminal i1 of the secondary switching network and the second output terminal o2 of the secondary circuit 3; secondary switching component S3 is coupled between the second input terminal i2 of the secondary switching network and the first output terminal o1 of the secondary circuit 3; and secondary switching component S4 is coupled between the second input terminal i2 of the secondary switching network and the second output terminal o2 of the secondary circuit 3.

[0045] In one alternative implementation, the secondary-side switching components S1-S4 are active switching devices with VI curve characteristics similar to GaN. Figure 4 This is a schematic diagram of the VI curve characteristics of the secondary-side switching assembly according to an embodiment of the present invention. Figure 4 As shown, when the driving voltage V of the switching device GSWhen the voltage is high, the switching device has low internal resistance and can carry both forward and reverse current (region A). When the driving voltage V of the switching device is high... GS When the voltage is low or negative, the switching device behaves like a diode with a high voltage drop, exhibiting forward blocking capability while still allowing flow in reverse (region C). It should be understood that in other alternative implementations, the secondary-side switching components S1-S4 of this embodiment can also employ transistors with similar properties, such as combined MOSFETs, and this embodiment does not limit this.

[0046] Figure 5 This is a schematic diagram of the secondary-side switching assembly according to an embodiment of the present invention. In this embodiment, the secondary-side switching assemblies S1-S4 can be implemented not only using GaN-type switching devices, but also by combining a switching transistor with unidirectional current capability in series with an ideal diode, and then connecting it in parallel with a module containing multiple high-voltage drop diodes in series. For example... Figure 5 As shown, the secondary-side switching assembly 5 includes a switching transistor SQ, a diode DQ, and a diode series module D21-D2N, where N is greater than or equal to 1.

[0047] In this configuration, the switching transistor SQ can be any type of active switching transistor, the diode DQ is an ideal diode (with a forward voltage drop of approximately 0 and a reverse leakage current of approximately 0), and D21-D2N is a series diode module with a high on-state voltage drop. The switching transistor SQ and diode DQ are connected in series and then connected in parallel with the diode series module D21-D2N. When the driving voltage of the switching transistor SQ is high, SQ is turned on, and the impedance of branch 51 is close to zero, much smaller than that of branch 52. Therefore, all current flows through branch 51, and the branch voltage drop is approximately zero. When the driving voltage of the switching transistor SQ is low, SQ is turned off, and the impedance of branch 52 is smaller than that of branch 51. Therefore, all current flows through branch 52, and the branch voltage drop is the sum of the individual voltage drops of all diodes. This also fulfills the requirement for the secondary-side switching component in this embodiment. It should be understood that the high voltage drop during turn-off is precisely what enables the step-down function.

[0048] In one optional implementation, the feedback control circuit 30 in the secondary circuit 3 includes an error acquisition circuit 31 for acquiring an error signal corresponding to the output voltage. Further optionally, the error acquisition circuit 31 may include a sampling circuit 311 and a comparison circuit A1. The sampling circuit 311 samples and acquires an output feedback signal Vf characterizing the output voltage. The comparison circuit A1 compares the output feedback signal Vf with a reference signal Vref to acquire the error signal Ve. The reference signal Vref characterizes the desired value of the output voltage.

[0049] In one optional implementation, the feedback control circuit 30 in the secondary circuit 3 further includes a drive circuit 32 for obtaining drive signals for the secondary switching components based on the error signal. Further optionally, the drive circuit 32 includes a comparator A2 for generating a PWM signal SW based on the error signal Ve and a predetermined ramp signal RAMP to control the secondary switching components S1-S4. Specifically, the PWM signal SW is positively selected to obtain the drive signals SWA for the secondary switching components S1 and S4, and negatively selected to obtain the drive signals SWb for the secondary switching components S2 and S3. During the unselected period, the drive signals SWA and SWb remain at a low level.

[0050] In one alternative implementation, the secondary circuit 3 further includes an output capacitor Co and an output equivalent resistance Ro. The output capacitor Co can be used to filter out high-frequency ripple to further stabilize the output signal.

[0051] In one alternative implementation, the secondary circuit 3 further includes a positive and negative phase detection circuit. Figure 3 (Not shown in the diagram) is used to detect the positive and negative phases of the secondary-side input signal to select the operating secondary-side switching component, thereby switching between different operating modes. Specifically, when the secondary-side input signal is in the positive half-cycle, the drive signal controls the secondary-side switching component to be in the first operating mode; when the secondary-side input signal is in the negative half-cycle, the drive signal is switched to control the secondary-side switching component to be in the second operating mode.

[0052] Optionally, in this embodiment, during the positive half-cycle of the secondary input signal, the drive signal controls the secondary switching components to operate in a first operating mode, that is, the secondary leakage inductance Lks is alternately energized and releases energy via the first secondary switching component S1 and the fourth secondary switching component S4. During the negative half-cycle of the secondary input signal, the drive signal controls the secondary switching components to operate in a second operating mode, that is, the secondary leakage inductance Lks is alternately energized and releases energy via the third secondary switching component S3 and the second secondary switching component S2.

[0053] Optionally, the positive and negative phase detection circuit can be implemented through zero-crossing detection, and this embodiment does not limit this. Optionally, the positive and negative phase detection circuit can obtain the positive and negative phases of the input signal by detecting the signals at endpoints c1, c2, or c3, and this embodiment does not limit the phase detection points.

[0054] In one alternative implementation, if the switching frequency of the secondary-side switching component is an integer multiple of the switching frequency of the primary-side switching component, the ramp signal RAMP can be synchronized by detecting the transition edge (e.g., rising edge) of the secondary-side input signal, thereby keeping it synchronized with the switching frequency of the primary side to further improve the voltage regulation efficiency.

[0055] Figure 6 This is a schematic diagram of the driving signals of the secondary-side switching components according to Embodiment 2 of the present invention. Secondary-side switching components S1 and S4 are selected during the positive half-cycle of the secondary-side input signal and remain low during the negative half-cycle. Secondary-side switching components S2 and S3 are selected during the negative half-cycle of the secondary-side input signal and remain low during the positive half-cycle. Further, in this embodiment, the feedback control circuit 30 is further configured to perform chopping control on the first secondary-side switching component S1 and the fourth secondary-side switching component S4 during the positive half-cycle of the secondary-side input signal, and control the third secondary-side switching component S3 and the second secondary-side switching component S2 to remain off. During the negative half-cycle of the secondary-side input signal, it performs chopping control on the second secondary-side switching component S2 and the third secondary-side switching component S3, and controls the first secondary-side switching component S1 and the fourth secondary-side switching component S4 to remain off. Among them, the first secondary side switch assembly S1 and the fourth secondary side switch assembly S4 have the same drive signal, and the third secondary side switch assembly S3 and the second secondary side switch assembly S2 have the same drive signal.

[0056] like Figure 6 As shown, the gating signal SW on Pri is set to high in the positive half-cycle of the secondary input signal and high in the negative half-cycle of the secondary input signal, so that the secondary switching components S1 and S4 are selected in the positive half-cycle of the secondary input signal, and the secondary switching components S2 and S3 are selected in the negative half-cycle of the secondary input signal.

[0057] Figures 7-10 This is the equivalent circuit diagram of the secondary-side circuit in Embodiment 2 of the present invention. During the positive half-cycle of the secondary-side input signal, when secondary-side switching components S1 and S4 are on and secondary-side switching components S2 and S3 are off (e.g., at times t4-t41), the equivalent circuit diagram of the secondary-side voltage regulator circuit is as follows. Figure 7 As shown. During times t4-t41, with the drive signal SWA of secondary-side switching components S1 and S4 high, secondary-side switching components S1 and S4 are controlled to conduct. The secondary-side leakage inductance Lks, secondary-side switching component S1, output equivalent resistance Ro, secondary-side switching component S4, and secondary-side coil form a current loop. At this time, the voltage of the secondary-side leakage inductance Lks is higher on the left and lower on the right. During times t41-t42, with the drive signal SWA of secondary-side switching components S1 and S4 low, secondary-side switching components S1 and S4 are in a high reverse conduction voltage drop mode. The secondary-side leakage inductance Lks, secondary-side switching component S1, output equivalent resistance Ro, secondary-side switching component S4, and secondary-side coil form a current loop. At this time, the voltage of the secondary-side leakage inductance Lks is lower on the left and higher on the right. The equivalent circuit diagram of the secondary-side voltage regulator circuit is shown below. Figure 8 As shown.

[0058] During times t4-t41, the drive signal SWA of secondary-side switching components S1 and S4 is high, and secondary-side switching components S1 and S4 are turned on. The corresponding impedance is very small, and the voltage drop across the transistor is approximately zero. At this time, the secondary voltage of the transformer is greater than the output voltage, and the voltage across the secondary leakage inductance Lks is positive on the left and negative on the right, indicating the excitation state. Energy flows from the transformer to the output terminal. During times t41-t42, the drive signal SWA of secondary-side switching components S1 and S4 is low, and secondary-side switching components S1 and S4 are turned off. At this time, secondary-side switching components S1 and S4 have high reverse conduction voltage drop characteristics, resulting in larger corresponding voltage drops Vdsa and Vdsd. Consequently, (Vdsa + Vdsd + Vo) is greater than the secondary voltage of the transformer, and the voltage across Lks is negative on the left and positive on the right, indicating the demagnetization state. Here, Vo is the output voltage of the secondary-side voltage regulator circuit. Thus, the excitation and demagnetization states in this embodiment achieve step-down conversion, similar to a Buck circuit.

[0059] During the negative half-cycle of the secondary input signal, when secondary switching components S2 and S3 are on and secondary switching components S1 and S4 are off (e.g., at times t5-t51), the equivalent circuit diagram of the secondary voltage regulator circuit is as follows: Figure 9 As shown. During times t5-t51, with the drive signal SWb of secondary-side switching components S2 and S3 high, secondary-side switching components S2 and S3 are controlled to conduct. The secondary-side leakage inductance Lks, secondary-side coil, secondary-side switching component S3, output equivalent resistance Ro, and secondary-side switching component S2 form a current loop. At this time, the voltage of the secondary-side leakage inductance Lks is negative on the left and positive on the right. During times t51-t52, with the drive signal SWb of secondary-side switching components S2 and S3 low, secondary-side switching components S2 and S3 are in a high reverse conduction voltage drop mode. The secondary-side leakage inductance Lks, secondary-side coil, secondary-side switching component S3, output equivalent resistance Ro, and secondary-side switching component S2 form a current loop. At this time, the voltage of the secondary-side leakage inductance Lks is positive on the left and negative on the right. The equivalent circuit diagram of the secondary-side voltage regulator circuit is shown below. Figure 10 As shown.

[0060] During times t5-t51, the drive signal SWb of secondary-side switching components S2 and S3 is high, and secondary-side switching components S2 and S3 are turned on. The corresponding impedance is very small, and the voltage drop across the transistor is approximately zero. At this time, the absolute value of the transformer secondary voltage is greater than the output voltage, and the voltage across the secondary leakage inductance Lks is negative on the left and positive on the right, indicating the excitation state. Energy flows from the transformer to the output terminal. During times t51-t52, the drive signal SWb of secondary-side switching components S2 and S3 is low, and secondary-side switching components S2 and S3 are turned off. At this time, secondary-side switching components S2 and S3 have high reverse conduction voltage drop characteristics, making the corresponding voltage drops Vdsb and Vdsc relatively large. Therefore, |Vdsa + Vdsd + Vo| is greater than the transformer secondary voltage, and the voltage across Lks is positive on the left and negative on the right, indicating the demagnetization state. Here, Vo is the output voltage of the secondary-side voltage regulator circuit. Thus, the excitation and demagnetization states in this embodiment achieve step-down conversion, similar to a Buck circuit.

[0061] In summary, during the positive half-cycle of the secondary input signal, the secondary leakage inductance Lks is alternately energized and releases energy via secondary switching components S1 and S4. During the negative half-cycle of the secondary input signal, the secondary leakage inductance Lks is alternately energized and releases energy via secondary switching components S3 and S2. In other words, in this embodiment, the secondary leakage inductance Lks is energized and demagnetized via secondary switching components S1-S4 during both the positive and negative half-cycles of the secondary input signal, achieving full-wave step-down rectification and thus secondary-side self-regulation.

[0062] In one optional implementation, to ensure that the current of the secondary leakage inductor Lks does not cross zero prematurely when the primary phase changes, a current detection circuit can be set in the secondary voltage regulator circuit 3 to detect the zero-crossing point and avoid the aforementioned situation. In another optional implementation, a diode D_block or a switching transistor that acts as a synchronous rectifier is connected in series between the positive output terminal of the rectifier bridge formed by S1-S4 in the secondary voltage regulator circuit 3 and the output capacitor Co, to ensure that the current of the secondary leakage inductor Lks will never reverse when the primary phase does not change, thereby ensuring the effectiveness of the control method in this embodiment.

[0063] Figure 11 This is a circuit diagram of the secondary-side voltage regulator circuit according to Embodiment 3 of the present invention. In this embodiment, the buck converter circuit is a buck half-wave rectifier circuit. Figure 11 As shown, in this embodiment, the secondary circuit 11 includes a converter composed of a secondary leakage inductance Lks' and a secondary switching component SA in the secondary switching network. The secondary switching component SA is connected between the leakage inductance Lks' (i.e., the input terminal of the secondary switching network) and the output terminal of the secondary circuit 11.

[0064] In one optional implementation, the feedback control circuit 110 in the secondary-side voltage regulator circuit 11 of this embodiment includes an error acquisition circuit 111. The error acquisition circuit 111 may include a sampling circuit 111a and a comparison circuit A3. The sampling circuit 111a is used to sample and acquire an output feedback signal Vf characterizing the output voltage. The comparison circuit A3 is used to compare the output feedback signal Vf and a reference signal Vref to acquire an error signal Ve. The reference signal Vref is used to characterize the expected value of the output voltage.

[0065] In an optional implementation, the feedback control circuit 110 in the secondary-side voltage regulator circuit 11 of this embodiment further includes a drive circuit 112. The drive circuit 112 further includes a comparator A4, used to generate a PWM signal SW based on the error signal Ve and a predetermined ramp signal RAMP, and to obtain a drive signal SWA for the secondary-side switching component SA through the positive-phase gating PWM signal SW, thereby controlling the secondary-side switching component SA, wherein the secondary-side switching component SA remains off during the unselected period. That is, in this embodiment, the feedback control circuit 110 is further configured to perform chopping control on the secondary-side switching component SA during the positive half-cycle of the secondary-side input signal, and to control the secondary-side switching component SA to remain off during the negative half-cycle, wherein the secondary-side leakage inductance is energized when the secondary-side switching component SA is controlled to conduct.

[0066] In one alternative implementation, the secondary circuit 11 may further include a positive and negative phase detection circuit. Figure 11 (Not shown in the diagram) A secondary-side switching component is used to detect the positive and negative phases of the secondary-side input signal to select the operating secondary-side switching component, thereby switching between different operating modes. Specifically, when the secondary-side input signal is in the positive half-cycle, the drive signal controls the secondary-side switching component to operate in the first operating mode; when the secondary-side input signal is in the negative half-cycle, the drive signal is switched to control the secondary-side switching component to operate in the second operating mode. Optionally, in this embodiment, during the positive half-cycle of the secondary-side input signal, the drive signal controls the secondary-side switching component to operate in the first operating mode, that is, the secondary-side leakage inductance Lks is energized and releases energy through the secondary-side switching component SA. During the negative half-cycle of the secondary-side input signal, the drive signal controls the secondary-side switching component to operate in the second operating mode, that is, the secondary-side switching component SA is turned off, thereby achieving half-wave rectification.

[0067] Optionally, the positive and negative phase detection circuit can be implemented through zero-crossing detection, and this embodiment does not limit this. Optionally, the positive and negative phase detection circuit can obtain the positive and negative phases of the input signal by detecting the signals at endpoints d1, d2, or d3, and this embodiment does not limit the phase detection points.

[0068] In one alternative implementation, if the switching frequency of the secondary-side switching component is an integer multiple of the switching frequency of the primary-side switching component, the ramp signal RAMP can be synchronized by detecting the transition edge (e.g., rising edge) of the secondary-side input signal, thereby keeping it synchronized with the switching frequency of the primary side to further improve the voltage regulation efficiency.

[0069] Figure 12 This is a schematic diagram of the drive signal of the secondary-side switching component according to Embodiment 3 of the present invention. The PWM signal SW is selected during the positive half-cycle of the secondary-side input signal, and the drive signal SWA of the secondary-side switching component SA remains low during the negative half-cycle of the secondary-side input signal.

[0070] like Figure 12 As shown, the strobe signal SW on Pri is positively high during the positive half-cycle of the secondary input signal and negatively high during the negative half-cycle of the secondary input signal, so that the secondary switching component SA strobes the PWM signal during the positive half-cycle of the secondary input signal.

[0071] Figures 13-14 This is the equivalent circuit diagram of the secondary-side voltage regulator circuit of Embodiment 3 of the present invention. During the positive half-cycle of the secondary-side input signal, when the secondary-side switching component SA is turned on (e.g., at times t6-t61), the equivalent circuit diagram of the secondary-side circuit 11 is as follows. Figure 13 As shown in the diagram. The secondary leakage inductance Lks', secondary switching component SA, output equivalent resistance Ro', and secondary coil form a current loop. At this time, the voltage of the secondary leakage inductance Lks' is positive on the left and negative on the right. The secondary leakage inductance Lks' is energized by short-circuiting the secondary switching component SA, and the energizing time is determined by the conduction time of the secondary switching component SA. During times t61-t62, when the drive signal SWA of the secondary switching component SA is low, the secondary switching component SA is in a high reverse conduction voltage drop mode. The secondary leakage inductance Lks', secondary switching component SA, output equivalent resistance Ro', and secondary coil form a current loop. At this time, the voltage of the secondary leakage inductance Lks' is negative on the left and positive on the right. The equivalent circuit diagram of the secondary voltage regulator circuit is shown in the diagram. Figure 14 As shown.

[0072] During times t6-t61, the drive signal SWA of the secondary-side switching component SA is high, turning SA on. The corresponding impedance is very small, and the voltage drop across the transistor is approximately zero. At this time, the transformer secondary voltage is greater than the output voltage, and the voltage across the secondary leakage inductance Lks' is positive on the left and negative on the right, indicating the excitation state. Energy flows from the transformer to the output terminal. During times t61-t62, the drive signal SWA of the secondary-side switching component SA is low, turning SA off. At this time, SA has a high reverse conduction voltage drop characteristic, resulting in a large voltage drop VA. Consequently, (VA+Vo) is greater than the transformer secondary voltage, and the voltage across Lks' is negative on the left and positive on the right, indicating the demagnetization state. Here, Vo is the output voltage of the secondary-side voltage regulator circuit. Thus, the excitation and demagnetization states in this embodiment achieve step-down conversion, similar to a Buck circuit.

[0073] During the positive half-cycle of the secondary input signal, the magnetization and demagnetization of Lks' are achieved by controlling the on and off states of the secondary-side switching component SA. During the negative half-cycle of the secondary input signal, the secondary-side switching component SA is kept in positive-side cutoff, meaning the secondary side is not operational. This achieves step-down half-wave rectification, thereby stabilizing the voltage and reducing losses. This embodiment uses the example of half-wave rectification achieved by gating during the positive half-cycle of the secondary input signal. It should be understood that the control logic can also be applied to scenarios where half-wave rectification is achieved by gating during the negative half-cycle of the secondary input signal; this embodiment does not limit this application.

[0074] In one optional implementation, to ensure that the current of the secondary leakage inductance Lks' does not cross zero prematurely when the primary phase changes, a current detection circuit can be set in the secondary circuit 11 to detect the zero-crossing point and avoid the aforementioned situation. In another optional implementation, a diode D_block' or a switching transistor that acts as a synchronous rectifier is connected in series in the secondary circuit 11 to ensure that the current of the secondary leakage inductance Lks' will never reverse when the primary phase does not change, thereby ensuring the effectiveness of the control method in this embodiment. It should be understood that... Figure 11 The connection position of diode D_block' in this embodiment is merely exemplary, and it can also be set in other positions, such as between the secondary-side switching component SA and the output capacitor Co'. This embodiment does not limit this.

[0075] In the above control method, when the switching frequency of the secondary-side switching component SA is an integer multiple of the switching frequency of the primary-side switching component, the switching frequency of the secondary-side input signal can be synchronized with the phase of the ramp signal RAMP by detecting the rising edge of the secondary-side input signal. This synchronizes the switching frequency of the secondary-side switching component SA with that of the primary-side switching component, thereby further improving the voltage regulation efficiency. It should be understood that in other control methods, phase synchronization may not be necessary; the secondary-side switching component SA can maintain a predetermined switching frequency, and the duty cycle determined by the negative feedback loop can be used to adjust the switching state. The drive signal of the secondary-side switching component is as follows: Figure 15 As shown, its control logic is similar to that of the above embodiments, and will not be repeated here. This further simplifies the circuit and control strategy.

[0076] Optionally, in this embodiment, when the secondary-side switching frequency is much greater than the primary-side switching frequency—that is, when the ratio of the secondary-side switching frequency to the primary-side switching frequency (or the difference between the secondary-side switching frequency and the primary-side switching frequency) is greater than a predetermined value—it is unnecessary to detect the positive and negative phases of the secondary-side input signal and synchronize the drive signal of the secondary-side switching component. This avoids backflow of load energy and further improves regulation efficiency. Simultaneously, while ensuring voltage regulation, this further simplifies the control strategy. Since no phase detection circuit is required, the circuit is also further simplified, reducing costs.

[0077] In this embodiment of the invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit. The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit controls the switching state of the secondary switching component in the secondary switching network based on the output feedback signal characterizing the output signal to achieve secondary leakage inductance excitation and freewheeling. Therefore, this embodiment can directly adjust the secondary switching component in the secondary switching network so that the secondary leakage inductance can act as a power storage element, achieving magnetic integration of the transformer. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs.

[0078] In summary, this invention uses leakage inductance as an energy storage element and performs closed-loop modulation control on the secondary-side switching components. The primary-side switching components can then perform open-loop energy transfer with a fixed duty cycle or frequency. This avoids the problems of increased voltage stress on components and reduced converter efficiency caused by increased leakage inductance in high-frequency transformer applications. Furthermore, compared to traditional two-stage circuits (i.e., closed-loop control on the primary side and open-loop control on the secondary side), this invention's strategy of shifting the control target from the primary to the secondary side reduces the number of signal isolation devices on both sides, lowering system cost and complexity.

[0079] Figure 16This is a flowchart of the secondary-side voltage regulation control method according to an embodiment of the present invention. Figure 16 As shown, the secondary-side voltage regulation control method of this invention includes the following steps:

[0080] Step S110: Sample and acquire the output feedback signal. The output feedback signal is used to characterize the output signal.

[0081] Step S120: Compare the output feedback signal and the reference signal to obtain the error signal of the output signal. The reference signal is used to characterize the expected value of the output signal.

[0082] Step S130: Based on the error signal and the preset ramp signal, obtain the drive signal for the secondary-side switching component in the secondary-side converter circuit. The ramp signal is used to determine the switching frequency of the secondary-side switching component.

[0083] Step S140: Control the corresponding secondary switch assembly to adjust the switching state according to the drive signal, so as to regulate the output signal.

[0084] In this embodiment of the invention, the isolated power converter includes a primary circuit, a transformer, and a secondary circuit. The secondary circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit excites the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network to achieve buck conversion. The feedback control circuit controls the switching state of the secondary switching component in the secondary switching network based on the output feedback signal characterizing the output signal to achieve secondary leakage inductance excitation and freewheeling. Therefore, this embodiment can directly adjust the secondary switching component in the secondary switching network so that the secondary leakage inductance can act as a power storage element, achieving magnetic integration of the transformer. This eliminates the need for feedback signal paths between the primary and secondary sides, reducing control complexity and manufacturing costs.

[0085] In summary, this invention uses leakage inductance as an energy storage element and performs closed-loop modulation control on the secondary-side switching components. The primary-side switching components can then perform open-loop energy transfer with a fixed duty cycle or frequency. This avoids the problems of increased voltage stress on components and reduced converter efficiency caused by increased leakage inductance in high-frequency transformer applications. Furthermore, compared to traditional two-stage circuits (i.e., closed-loop control on the primary side and open-loop control on the secondary side), this invention's strategy of shifting the control target from the primary to the secondary side reduces the number of signal isolation devices on both sides, lowering system cost and complexity.

[0086] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. For those skilled in the art, the present invention can be modified and varied in various ways. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principle of the present invention should be included within the scope of protection of the present invention.

Claims

1. An isolated power converter, characterized in that, The isolated power converter includes: The primary circuit is configured to generate an AC input signal; A transformer, including a primary winding and a secondary winding, is configured to process the AC input signal to generate a secondary input signal; The secondary-side circuit includes a buck converter circuit and a feedback control circuit. The buck converter circuit is a buck full-wave rectifier circuit or a buck half-wave rectifier circuit. The buck converter circuit is configured to convert the secondary-side input signal to generate an output signal. The feedback control circuit is configured to control the switching state of the secondary-side switching components in the secondary-side switching network according to an output feedback signal characterizing the output signal. When the buck converter circuit is a full-wave rectifier circuit, the feedback control circuit performs buck chopping control on the secondary-side switching components in the secondary-side switching network during the positive and negative half-cycles of the secondary-side input signal, so that the secondary leakage inductance of the secondary winding alternately magnetizes and demagnetizes to achieve buck converter operation. When the buck converter circuit is a half-wave rectifier circuit, the feedback control circuit performs buck chopping control on the secondary-side switching components in the secondary-side switching network during either the positive or negative half-cycle of the secondary-side input signal, so that the secondary leakage inductance of the secondary winding alternately magnetizes and demagnetizes to achieve buck converter operation. The secondary-side switching component is a GaN-characteristic transistor or a switching module composed of an active switching transistor and multiple diodes. When the secondary-side switching assembly is a switching module composed of an active switching transistor and multiple diodes, the secondary-side switching assembly includes a first branch and a second branch connected in parallel. The first branch includes an active switching transistor and a diode connected in series, and the second branch includes a diode series module. The forward voltage drop of the diode in the first branch is basically zero, and the diode in the second branch has a high forward voltage drop.

2. The isolated power converter according to claim 1, characterized in that, The primary-side circuit includes a primary-side impedance network and a primary-side switching network, wherein the primary-side switching components in the primary-side switching network are subjected to open-loop control with a fixed duty cycle or a fixed frequency.

3. The isolated power converter according to claim 1, characterized in that, The switching frequency of the secondary-side switch assembly is not less than the switching frequency of the primary-side switch assembly.

4. The isolated power converter according to claim 1, characterized in that, The switching frequency of the secondary-side switch assembly is an integer multiple of the switching frequency of the primary-side switch assembly.

5. The isolated power converter according to claim 1, characterized in that, The feedback control circuit is further configured to adjust the duty cycle of each of the secondary-side switching components based on the error between the output feedback signal and the reference signal, so that the output signal tends toward the desired value.

6. The isolated power converter according to any one of claims 1-5, characterized in that, The step-down converter circuit is a step-down full-wave rectifier circuit, and the switching frequency of the secondary-side switching component is not less than twice the switching frequency of the primary-side switching component. The secondary-side switch network includes a first secondary-side switch assembly, a second secondary-side switch assembly, a third secondary-side switch assembly, and a fourth secondary-side switch assembly; Wherein, the first secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the first output terminal of the secondary-side circuit, the second secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit, the third secondary-side switch assembly is coupled between the second input terminal of the secondary-side switch network and the first output terminal of the secondary-side circuit, and the fourth secondary-side switch assembly is coupled between the second input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit.

7. The isolated power converter according to claim 6, characterized in that, During the positive half-cycle of the secondary input signal, the secondary leakage inductance is alternately energized and demagnetized via the first and fourth secondary switching components. During the negative half-cycle of the secondary input signal, the secondary leakage inductance is alternately energized and demagnetized via the third and second secondary switching components.

8. The isolated power converter according to claim 7, characterized in that, The feedback control circuit is further configured to perform chopping control on the first secondary-side switching component and the fourth secondary-side switching component during the positive half-cycle of the secondary-side input signal, and control the third secondary-side switching component and the second secondary-side switching component to remain in the off state; and to perform chopping control on the second secondary-side switching component and the third secondary-side switching component during the negative half-cycle of the secondary-side input signal, and control the first secondary-side switching component and the fourth secondary-side switching component to remain in the off state. The first and fourth secondary-side switch components have the same drive signal, and the third and second secondary-side switch components have the same drive signal.

9. The isolated power converter according to any one of claims 1-5, characterized in that, The step-down converter circuit is a step-down half-wave rectifier circuit, and the secondary-side switching network includes a fifth secondary-side switching component, which is coupled between the first input terminal of the secondary-side switching network and the first output terminal of the secondary-side circuit.

10. The isolated power converter according to claim 9, characterized in that, The feedback control circuit is further configured to perform chopping control on the fifth secondary-side switching component during the positive half-cycle of the secondary-side input signal, and to control the fifth secondary-side switching component to remain in the off state during the negative half-cycle, wherein the secondary-side leakage inductance is energized when the fifth secondary-side switching component is controlled to be turned on.

11. The isolated power converter according to claim 1, characterized in that, The secondary circuit also includes a diode connected to the output circuit to prevent current from flowing in reverse.

12. The isolated power converter according to claim 1, characterized in that, The feedback control circuit also includes: An error acquisition circuit is configured to acquire an error signal based on the difference between the feedback signal and a reference signal; and The driving circuit is configured to generate a PWM signal based on the error signal and the ramp signal for chopping control, thereby determining the driving signal for each of the secondary-side switching components and controlling the switching state of the corresponding secondary-side switching components according to the driving signal.

13. The isolated power converter according to claim 6, characterized in that, During the positive half-cycle of the secondary input signal, the third and second secondary switching components are disconnected, and the first and fourth secondary switching components are turned on and off according to the switching frequency of the secondary switching components to alternately excite and demagnetize the secondary leakage inductance of the secondary winding. During the negative half-cycle of the secondary input signal, the first and fourth secondary switching components are disconnected, and the third and second secondary switching components are turned on and off according to the switching frequency of the secondary switching components to excite and demagnetize the secondary leakage inductance of the secondary winding.

14. The isolated power converter according to claim 9, characterized in that, During the positive half-cycle of the secondary input signal, the fifth secondary-side switching component turns on and off according to its switching frequency to alternately energize and demagnetize the secondary leakage inductance of the secondary winding. Specifically, the secondary leakage inductance is energized when the fifth secondary-side switching component is on and demagnetized when it is off. During the negative half-cycle of the secondary input signal, the fifth secondary switch assembly remains in the off state.

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