Electro-acoustic transduction system, its mixed frequency wave impedance matching method and control system

Abstract

This invention discloses an electroacoustic transducer system and its mixer impedance matching method and control system. The impedance matching circuit includes a single-phase full-bridge inverter and an AC output filter inductor L. f Output filter inductor L f One end is connected to the inverter, and the other end is connected in parallel between the power amplifier and the load to compensate for the reactive power required by the load. The control method is based on instantaneous power theory. p -i q The algorithm calculates the inductive reactive component in each single-frequency current of the load, and superimposes the single-frequency reactive currents to obtain the total inductive reactive current of the load. Then, by controlling the impedance matching network, a capacitive reactive current in the opposite direction to this inductive reactive current is generated to achieve the impedance matching function. Under the condition of constant power amplifier capacity, the reactive power output of the power amplifier is reduced, the power factor of the power amplifier is improved, and it can output more active power. It also effectively reduces the current stress and switching loss of power devices.

Description

Technical Field

[0001] This invention relates to the field of power conversion technology in electrical engineering, and in particular to an electroacoustic transducer system and its mixing impedance matching method and control system. Background Technology

[0002] The giant magnetostrictive transducer is a crucial component of electroacoustic transduction systems and a key device for achieving electroacoustic energy conversion. Since the giant magnetostrictive transducer functions as an inductive load, the power amplifier transmits not only active power but also a significant amount of reactive power during the power supply process. This results in underutilization of the power amplifier's capacity and failure to meet the high output power and energy conversion efficiency requirements of the electroacoustic transducer. The most direct way to increase the power amplifier's active power output is to use a larger capacity amplifier, but this also proportionally increases the reactive power output, failing to reduce the reactive power and improve the power factor. Furthermore, increasing the power amplifier's capacity requires the switching devices to withstand higher voltages, larger currents, or abnormal thermal stresses. A better approach is to equip the power amplifier with an impedance matching network to improve the overall system efficiency, reduce reactive power transfer, decrease system heat generation, and enhance system reliability.

[0003] When a transducer emits only a single-frequency wave or a narrow-band acoustic wave, commonly used impedance matching networks are mostly composed of a single fixed capacitor, a capacitor array, or multiple sets of impedance matching circuits. CN108882108A discloses a capacitor bank impedance matching method, which can perform impedance matching according to the transducer's operating environment and load changes. However, it requires multiple single capacitors to be combined, thus having the disadvantages of large size and low matching accuracy. It can only achieve "stepped" adjustment, not "stepless" adjustment, i.e., continuous adjustment. In addition, the inrush current generated during capacitor switching must be considered to prevent the inrush current from affecting the system stability. CN113037230A discloses an impedance matching control method and system for an electroacoustic transducer system, which uses an impedance matching network to compensate for reactive voltage. When using this method for mixer impedance matching, the output current of the power amplifier is large, and the current stress on the power amplifier's power devices is large. This method requires identifying the equivalent resistance and inductance parameters of the load to obtain the reactive voltage during matching. For transducer loads with variable equivalent impedance, the fluctuation of the load parameter identification results affects the calculation accuracy of the reference reactive voltage. Under mixed-frequency waveforms, there may be certain errors, thus affecting the matching accuracy.

[0004] When the output frequency of a power amplifier changes or the equivalent inductance of the load changes, the load impedance also changes accordingly, causing problems such as poor matching accuracy and high matching difficulty in traditional fixed impedance matching systems. Furthermore, for impedance matching methods under mixed-frequency waves composed of multiple superimposed frequency components, neither a single capacitor nor a combination of capacitors and inductors can achieve perfect impedance matching. Moreover, the load impedance parameters are also changing in real time, making it difficult for conventional matching methods to achieve high-precision impedance matching for loads with a wide range of variations. Therefore, researching an adaptive impedance matching circuit under mixed-frequency waves is of great significance for improving the performance of electroacoustic transducer systems. Summary of the Invention

[0005] The technical problem to be solved by the present invention is to provide an electroacoustic transducer system and its mixing wave impedance matching method and control system, which are in response to the shortcomings of the prior art, thereby reducing the reactive power output of the power amplifier and the current stress of the power device, and improving the power factor and energy transmission efficiency of the power amplifier.

[0006] To solve the above-mentioned technical problems, the technical solution adopted by the present invention is: an electroacoustic transducer system, comprising a power amplifier and an impedance matching network; the power amplifier includes a DC-side capacitor and an inverter; the inverter includes two parallel bridge arms, which are connected in parallel with the DC-side capacitor; the power amplifier and the impedance matching network are connected in parallel and then connected to a load network; the impedance matching network includes a single-phase full-bridge inverter and an AC output filter inductor L. f The output filter inductor L f One end is connected to a single-phase full-bridge inverter, and the other end is connected in parallel between the power amplifier and the load network.

[0007] The actual current i output by the impedance matching network m With matching current reference value i m_ref The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with a high-frequency triangular wave to obtain the trigger pulse of the switching device in the single-phase full-bridge inverter.

[0008] Among them, the power amplifier output voltage u s Load current i L The output current i of the impedance matching network m Sampling is performed using i based on instantaneous reactive power theory. p -i q The algorithm calculates the load reactive current at each frequency, and then superimposes the reactive currents at each single frequency to obtain the total load reactive current i. Lq That is, matching current i m Reference value i m_ref .

[0009] This invention employs a parallel active impedance matching network for impedance matching. After impedance matching, the power amplifier current decreases, containing only active current and outputting only active power. This allows the power amplifier to provide the same active power to the load while reducing the reactive power output of the power amplifier and the current stress on the power devices, thereby improving the power factor and energy transfer efficiency of the power amplifier.

[0010] This invention also provides a method for impedance matching of a mixer wave in an electroacoustic transducer system. The electroacoustic transducer system includes a power amplifier and an impedance matching network; the power amplifier and the impedance matching network are connected in parallel and then connected to a load network; it also includes a controller; the controller includes:

[0011] The acquisition module is used to measure the output voltage u of the power amplifier. s Load current i L The output current i of the impedance matching network m Perform sampling;

[0012] The first calculation unit is used to calculate the output voltage u of the power amplifier. s Phase-locked operation is performed on the phases of each single-frequency voltage to obtain the phases of each single-frequency voltage. Then, a unit sine function sin(kωt) lagging behind the phases of each single-frequency voltage by π / 2 and a unit sine function 2sin(kωt) with twice the amplitude are constructed.

[0013] The second calculation unit is used to calculate the load current i L Multiplying it by the unit sine function sin(kωt) yields the instantaneous current i. qk The DC component is obtained after passing through a low-pass filter. The magnitude of the DC component is half the amplitude of the reactive current. Therefore, multiplying this DC component by a unit sine function 2sin(kωt) with twice the amplitude yields the reactive current components of each single-frequency load current. The total load reactive current i is obtained by superimposing the reactive components of each single-frequency load current. Lq That is, the matching current reference value i m_ref ;

[0014] Trigger unit, used to set the matching current reference value i m_ref The actual value of the current i output by the impedance matching network m The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with a high-frequency triangular wave to obtain the trigger pulse of the switching device in the single-phase full-bridge inverter.

[0015] The control system of the present invention further includes a power amplifier output voltage control module: used to compare a reference voltage and the actual output voltage of the power amplifier, input the error signal after comparison to a first PI controller, use the output of the first PI controller as a reference value of the power amplifier output filter inductor current, compare the reference value with the power amplifier output filter inductor current and input it to a second PI controller, and input the output of the second PI controller to a PWM module to obtain the trigger pulse required by the switching devices in the inverter, thereby controlling the output voltage of the power amplifier.

[0016] In this invention, the power amplifier output voltage control module is integrated into the controller; or, the power amplifier output voltage control module is separately located in another controller.

[0017] In this invention, the instantaneous current i q1 i q2 ... qk ... qn The calculation formula is:

[0018] i qk =i L ·sin(kωt)(k=1,2,3,...,n)

[0019] The DC component is obtained after passing a low-pass filter. The expressions for each DC component are as follows:

[0020]

[0021] The formulas for calculating the load reactive current at each frequency are as follows:

[0022]

[0023] The total load reactive current value or matching current reference value is:

[0024]

[0025] in, I is relative to the initial phase angle of each single-frequency current. k These are the effective values ​​of the single-frequency current for each cycle.

[0026] This invention employs a current compensation method. By detecting the reactive component in the load current, it uses a parallel active impedance matching network for impedance matching. After impedance matching, the power amplifier current decreases, outputting almost only active current and active power. This allows the power amplifier to provide the same active power to the load while reducing the reactive power output of the power amplifier and the current stress on the power devices, thereby improving the power amplifier's power factor and energy transfer efficiency. This invention eliminates the need for identifying the equivalent resistance and inductance parameters of the load to obtain the reactive voltage; it only requires measuring the voltage and current across the load to calculate the reactive current component for reactive compensation. Therefore, this invention achieves high impedance matching accuracy.

[0027] Compared with existing technologies, the beneficial effects of this invention are as follows: By detecting the reactive component in the load current and employing a parallel active impedance matching network for impedance matching, this invention reduces the reactive power output of the power amplifier and the current stress on the power devices, thereby improving the power factor and energy transfer efficiency of the power amplifier. Compared with traditional impedance matching devices, this invention can achieve high-precision dynamic impedance matching under varying waveforms of the power amplifier's output mixer and changes in load parameters. Attached Figure Description

[0028] Figure 1 This is an impedance matching circuit according to an embodiment of the present invention;

[0029] Figure 2 This is a block diagram of the impedance matching network control according to an embodiment of the present invention;

[0030] Figure 3 This is a block diagram of the power amplifier control according to an embodiment of the present invention;

[0031] Figure 4 The power amplifier output voltage and current, impedance matching network output current, and reactive power waveforms on the power amplifier side and the load side are obtained before and after impedance matching using the embodiments of the present invention.

[0032] Figure 5 The waveforms of the power amplifier side voltage and current, the load side voltage and current, the impedance matching network output current, and the reactive power on the power amplifier side and the reactive power on the load side are obtained by using the embodiments of the present invention to perform impedance matching under variable load. Detailed Implementation

[0033] like Figure 1As shown in Embodiment 1 of the present invention, the electroacoustic transducer system includes a power amplifier and an impedance matching network; the power amplifier includes a DC-side capacitor and an inverter; the inverter includes two parallel bridge arms, which are connected in parallel with the DC-side capacitor. The midpoint of one bridge arm is connected to one end of a filter inductor L0, the other end of the filter inductor L0 is connected to the positive terminal of a filter capacitor C0, and the negative terminal of the filter capacitor C0 is connected to the midpoint of the other bridge arm. In the power amplifier, each bridge arm includes two switching devices (Qx, x = 1, 2, 3, 4).

[0034] The impedance matching network includes a DC-side capacitor and two parallel bridge arms, each containing two switching devices (Sx, x = 1, 2, 3, 4). One of the bridge arms of the impedance matching network is connected to a filter inductor L. f .

[0035] The load network uses a series resistor R load and inductor L load Equivalent substitution.

[0036] In this embodiment of the invention, the power device of the bridge arm is a fully controllable power electronic device (IGBT or MOSFET).

[0037] Impedance matching network is connected in parallel between the power amplifier and the load through an output filter inductor. By controlling the bridge arm of the impedance matching network, the output current on the filter inductor is changed to achieve impedance matching.

[0038] like Figure 2 As shown, in Embodiment 2 of the present invention, the control process of the impedance matching network includes the following steps:

[0039] (1) Establishment of mathematical model

[0040] according to Figure 1 Given the voltage and current reference directions, for the impedance matching network, the system mathematical model is established as follows:

[0041]

[0042] Perform Laplace transform on it:

[0043] sL f i m (s)=u inv (s)-u s (s)

[0044] (2) Output matching current control

[0045] The reference value i of the matching network output current is calculated from the load voltage, load current, and system operating frequency. m_refBy using an inverter to compensate for the reactive current component in the load current, the reactive current component in the power amplifier is eliminated, thereby achieving impedance matching. This is achieved by using the matching network output current reference value i... m_ref The actual current i output by the matching network m The comparison is performed, and the comparison result is input into the PI controller. The output of the PI controller is compared with the high-frequency triangular carrier wave to generate the trigger pulse required by the inverter, thereby controlling the inverter (i.e., the two parallel bridge arms in the impedance matching network).

[0046] The specific process includes:

[0047] S1, Regarding the power amplifier output voltage u s Load current i L The output current i of the impedance matching network m Perform sampling;

[0048] S2, Regarding the power amplifier output voltage u s Phase-locked loop is performed on the phases of each single-frequency voltage to obtain the phases of each single-frequency voltage. Then, a unit sine function sin(kωt) lagging behind the phases of each single-frequency voltage by π / 2 and a unit sine function 2sin(kωt) with twice the amplitude are constructed.

[0049] S3, change the load current i L Multiplying it by the unit sine function sin(kωt) yields the instantaneous current i. qk The DC component is obtained after passing through a low-pass filter. The magnitude of the DC component is half the amplitude of the reactive current. Therefore, multiplying this DC component by a unit sine function 2sin(kωt) with twice the amplitude yields the reactive current components of each single-frequency load current. The total reactive current i is obtained by superimposing the reactive components of each single-frequency current. Lq That is, the matching current reference value i m_ref ;

[0050] S4. Set the total output current reference value i of the matching network. m_ref Actual value of output current i of impedance matching network m The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with the high-frequency triangular wave to obtain the trigger pulse of the inverter in the impedance matching network.

[0051] like Figure 3As shown, in Embodiment 3 of the present invention, the output voltage of the power amplifier is obtained by the following method: comparing the reference voltage and the actual output voltage of the power amplifier, inputting the error signal after comparison to the first PI controller, using the output of the first PI controller as the reference value of the output filter inductor current of the power amplifier, comparing the reference value with the output filter inductor current of the power amplifier and inputting it to the second PI controller, and inputting the output of the second PI controller to the PWM module to obtain the trigger pulse required by the switching devices in the inverter, thereby controlling the output voltage of the power amplifier.

[0052] The program instructions corresponding to Embodiments 2 and 3 above can be implemented by different controllers or executed by the same controller.

[0053] The following is used to verify that the embodiments of the present invention can effectively perform high-precision impedance matching and improve the power factor.

[0054] Power amplifier system parameters:

[0055] DC side voltage U of power amplifier dc_pa =600V, filter inductor is 1mH, filter capacitor is 50uf, and the output voltage is a mixed wave obtained by superimposing single-frequency voltages with amplitudes of 200V and frequencies of 100Hz and 200Hz respectively.

[0056] Impedance matching network system parameters:

[0057] DC voltage U dc =700V, filter inductor L f =3mH;

[0058] Load parameters:

[0059] Equivalent inductance value: L load =5mH~10mH, equivalent resistance value: R load =5Ω~8Ω.

[0060] The following are the simulation results for the above parameters:

[0061] Figure 4The simulation results are presented for a system where the power amplifier output voltage is a mixed wave obtained by superimposing single-frequency voltages with an amplitude of 200V and frequencies of 100Hz and 200Hz respectively, with an equivalent load resistance of 5Ω and an equivalent inductance of 5mH. The figure shows, from top to bottom, the power amplifier side voltage and current, the load side voltage and current, the matching network output current, the active and reactive power waveforms on the power amplifier side, and the reactive power waveform on the load side. Before impedance matching is initiated, due to the inductive nature of the load, the power amplifier side current lags behind the voltage, resulting in a low power factor. Its power factors at 100Hz and 200Hz are 0.84 and 0.62 respectively, significantly reducing the power amplifier's output efficiency. After impedance matching is initiated, the power amplifier side voltage and current quickly reach the same phase after a brief transient time, effectively improving the power factor. The power factors of the power amplifier at 100Hz and 200Hz are 0.99 and 0.99 respectively, effectively improving the power amplifier's output efficiency. As can be seen from the reactive power on the power amplifier side, after starting impedance matching, the reactive power output of the power amplifier gradually decreases to close to 0.

[0062] Figure 5 The simulation results are presented for a mixed-frequency waveform obtained by superimposing single-frequency voltages with an amplitude of 200V and frequencies of 100Hz and 200Hz respectively, when the equivalent load resistance changes from 6Ω to 8Ω and the equivalent load inductance changes from 8mH to 10mH. The figure shows, from top to bottom, the voltage and current on the power amplifier side, the voltage and current on the load side, the output current of the impedance matching network, the active and reactive power waveforms on the power amplifier side, and the reactive power waveform on the load side. The reactive power waveform on the load side shows that the load is inductive. Before the load change, the power factors on the load side at 100Hz and 200Hz were 0.76 and 0.51, respectively. After the load change, the power factors on the load side at 100Hz and 200Hz were 0.78 and 0.53, respectively. Therefore, there is a large amount of inductive reactive power on the load side. As can be seen from the voltage and current on the power amplifier side, before and after the load change, the voltage and current on the power amplifier side remain in the same phase, the output reactive power is still close to 0, and the power factor under single frequency reaches 0.99, which improves the power factor and output efficiency of the power amplifier.

[0063] The simulation results above verify that the method of the present invention can operate effectively in the mixing wave range, and prove that the impedance matching circuit proposed in this invention and the control method of impedance matching by compensating for the reactive current of the load can perform load impedance matching under the mixing wave, improve the power factor, and thus improve the operating efficiency of the electroacoustic transducer system.

Claims

1. An electroacoustic transducer system, comprising a power amplifier and an impedance matching network; the power amplifier comprising a DC-side capacitor and an inverter; the inverter comprising two parallel bridge arms, the two parallel bridge arms being connected in parallel with the DC-side capacitor; characterized in that, The power amplifier is connected in parallel with the impedance matching network and then connected to the load network. The impedance matching network includes a single-phase full-bridge inverter and an AC output filter inductor. One end of the output filter inductor is connected to the single-phase full-bridge inverter, and the other end is connected in parallel between the power amplifier and the load network. The actual current i output by the impedance matching network m With matching current reference value i m_ref The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with a high-frequency triangular wave to obtain the trigger pulse of the switching device in the single-phase full-bridge inverter. Among them, i is based on the theory of instantaneous reactive power. p -i q The algorithm calculates the load reactive current at each frequency, and then sums the reactive current values ​​at each frequency to obtain the total load reactive current i. Lq That is, to obtain the matching current reference value i m_ref i m_ref =i Lq ; The specific process for obtaining the load reactive current at each frequency includes: Load current i L Multiplying it by the unit sine function sin(kωt) yields the instantaneous current i. qk The DC component is obtained after passing through a low-pass filter. The magnitude of the DC component is half of the reactive current amplitude. Multiplying the DC component by the unit sine function 2sin(kωt) with twice the amplitude yields the reactive current components of each single-frequency load current. The specific process of obtaining the unit sine function 2sin(kωt) includes: calculating the output voltage u of the power amplifier... s Phase-locked loop is performed on the phases of each single-frequency voltage to obtain the phases of each single-frequency voltage. A unit sine function sin(kωt) lagging behind the phases of each single-frequency voltage by π / 2 and a unit sine function 2sin(kωt) with twice the amplitude are constructed.

2. A method for impedance matching of a mixer wave in an electroacoustic transducer system as described in claim 1, wherein the electroacoustic transducer system includes a power amplifier and an impedance matching network; the power amplifier and the impedance matching network are connected in parallel and then connected to a load network; the power amplifier includes a DC-side capacitor and an inverter; the inverter includes two parallel bridge arms, which are connected in parallel with the DC-side capacitor; the impedance matching network includes a single-phase full-bridge inverter and an AC output filter inductor L. f The output filter inductor L f One end is connected to a single-phase full-bridge inverter, and the other end is connected in parallel between the power amplifier and the load network; its characteristic is that... Includes the following steps: 1) Regarding the output voltage u of the power amplifier s Load current i L The output current i of the impedance matching network m Perform sampling; 2) i based on instantaneous reactive power theory p -i q The algorithm calculates the load reactive current at each frequency, and then sums the reactive current values ​​at each frequency to obtain the total load reactive current i. Lq That is, to obtain the matching current reference value i m_ref i m_ref =i Lq ; 3) Set the matching current reference value i m_ref The actual value of the current i output by the impedance matching network m The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with a high-frequency triangular wave to obtain the trigger pulse of the switching device in the single-phase full-bridge inverter.

3. The impedance matching method for the mixer wave of the electroacoustic transducer system according to claim 2, characterized in that, Step 2) specifically includes the following steps: adjusting the power amplifier output voltage u s Phase-locked loop is performed on the phases of each single-frequency voltage to obtain the phases of each single-frequency voltage. A unit sine function sin(kωt) lagging behind the phases of each single-frequency voltage by π / 2 is constructed, and the load current i is... L Multiplying the result by the unit sine function yields the instantaneous current i. qk The expressions for each instantaneous current are as follows: ； Where n is the number of times the maximum frequency is reached; instantaneous current i qk The DC component is obtained after passing a low-pass filter. The expressions for each DC component are as follows: ； Where, φ k I is relative to the initial phase angle of each single-frequency current. k The DC component is half the magnitude of the reactive current amplitude. Multiplying the DC component by a unit sine function 2sin(kωt) with twice the amplitude yields the reactive current component of the load's single-frequency current. ； The total load reactive current value or matching current reference value is: 。 4. A control system for an electroacoustic transducer system, the electroacoustic transducer system comprising a power amplifier and an impedance matching network; the power amplifier and the impedance matching network are connected in parallel and then connected to a load network; the impedance matching network comprises a single-phase full-bridge inverter and an AC output filter inductor L. f The output filter inductor L f One end is connected to a single-phase full-bridge inverter, and the other end is connected in parallel between the power amplifier and the load network; its characteristic is that... It also includes a controller; the controller includes: The acquisition module is used to measure the output voltage u of the power amplifier. s Load current i L The output current i of the impedance matching network m Perform sampling; The first calculation unit is used to calculate the output voltage u of the power amplifier. s Phase-locked operation is performed on the phases of each single-frequency voltage to obtain the phases of each single-frequency voltage. Then, a unit sine function sin(kωt) lagging behind the phases of each single-frequency voltage by π / 2 and a unit sine function 2sin(kωt) with twice the amplitude are constructed. The second calculation unit is used to calculate the load current i L Multiplying it by the unit sine function sin(kωt) yields the instantaneous current i. qk The DC component is obtained after passing through a low-pass filter. The magnitude of the DC component is half the amplitude of the reactive current. Multiplying the DC component by a unit sine function 2sin(kωt) with twice the amplitude yields the reactive current components of each single-frequency load current. Superimposing the reactive current components of each single-frequency current yields the matching current reference value i. m_ref ; Trigger unit, used to set the matching current reference value i m_ref The actual value of the current i output by the impedance matching network m The difference obtained after comparison is input to the PI controller, and the output of the PI controller is compared with a high-frequency triangular wave to obtain the trigger pulse of the switching device in the single-phase full-bridge inverter.

5. The control system of the electroacoustic transducer system according to claim 4, characterized in that, It also includes a power amplifier output voltage control module: used to compare the reference voltage and the actual output voltage of the power amplifier, input the error signal after comparison to the first PI controller, use the output of the first PI controller as the reference value of the power amplifier output filter inductor current, compare the reference value with the power amplifier output filter inductor current and input it to the second PI controller, and input the output of the second PI controller to the PWM module to obtain the trigger pulse required by the switching devices in the inverter, and control the output voltage of the power amplifier.

6. The control system of the electroacoustic transducer system according to claim 5, characterized in that, The power amplifier output voltage control module is integrated into the controller; or, the power amplifier output voltage control module is separately located in another controller.

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