switching sub-circuit

Abstract

A soft switching subcircuit forming part of or for use with a circuit. The soft switching subcircuit includes a bridge switching circuit. The soft switching subcircuit is configured and operable to provide a varying current output that tracks an output current from the bridge switching circuit to produce substantially zero current through at least one switching component of the bridge switching circuit to achieve soft switching.

Description

Technical Field

[0001] The present invention relates to a switching circuit that can be used independently, as part of another circuit, or in conjunction with other circuits. Background Technology

[0002] MOSFET

[0003] Switching devices such as MOSFETs, IGBTs, and BJTs are used in power electronic circuits to convert DC voltage to AC voltage, with MOSFETs being the preferred device for systems switching at frequencies above 20 kHz. The following explanation will be based on... Figure 15 The MOSFET switch shown (showing its parasitic capacitance and body diode).

[0004] A MOSFET has terminals G (gate), D (drain), and S (source), a body diode, and parasitic capacitance between its terminals, such as... Figure 15 As shown in the left figure, the body diode is inherent to the device structure and cannot be removed. It allows current to flow from source (S) to drain (D) regardless of whether the MOSFET is on or off. When the MOSFET is on, current can flow in both directions between drain and source.

[0005] MOSFET losses can occur during conduction, turn-on, and turn-off. Conduction losses occur when the MOSFET is in the ON state and depend on the values ​​of Rd and the MOSFET current. Depending on the direction of the MOSFET current, turn-on and turn-off switching losses can be categorized as hard switching losses and soft switching losses.

[0006] Hard switch

[0007] Hard switching occurs where the MOSFET current flows from drain (D) to source (S) during switching. For example... Figure 16 The dashed circle in the diagram illustrates the significant switching losses caused by the overlap between Vds and the MOSFET current, showing typical switching waveforms for hard turn-off and hard turn-on. More specifically, Figure 16 The diagram shows hard turn-off (left) and hard turn-on (right) (Vds = 400V, MOSFET current = 18A). The top graph represents Vg. The middle curve represents Vds and the MOSFET current (D to S). The last graph (red) is the product of V(D, S) and the MOSFET current or the instantaneous power loss of the MOSFET. For both switching losses, current flows from the drain to the source before switching, causing an overlap between the MOSFET voltage and current. It can be seen that the instantaneous switching losses can reach several kilowatts and last for hundreds of nanoseconds.

[0008] Hard-turn and hard-turn switching losses increase with higher Vds and / or higher MOSFET current. For the same Vds and MOSFET current, hard-turn losses are significantly higher than hard-turn losses. This is in Figure 16 The diagram shows that the peak instantaneous losses during hard turn-on and hard turn-off reach 12kW and 3.6kW, respectively. In this case, the energy loss during hard turn-on is more than three times higher than the energy loss during hard turn-off. (Switching loss diagram from MOSFET datasheet...) Figure 17 It shows the total switching energy (E) of the MOSFET C3M0065090D at 400V. 总 =E 接通 +E 关断 ), connect (E) 接通 ) and shutdown (E) 关断 The energy further confirms this observation, where E 接通 and E 关断 These are the energy losses from hard-on and hard-off states, respectively, and E 总 =E 接通 +E 关断 .

[0009] For this reason, most systems are intentionally designed to avoid hard turn-on and allow hard turn-off, since hard turn-on and hard turn-off are difficult to avoid.

[0010] Diode reverse recovery

[0011] Another reason to avoid hard-on is related to a phenomenon called diode reverse recovery, which occurs in half-bridge configurations operating in a hard-on state. Diode reverse recovery can significantly increase hard-on losses and generate electromagnetic interference (EMI) problems. It occurs in both the discrete diode and the body diode of the MOSFET.

[0012] When a diode is suddenly reverse-biased (turned off) while conducting current, it exhibits reverse recovery. The reverse recovery current flows from the cathode to the anode, such as... Figure 18 As shown ( Figure 18 A graphical version of the diode is shown, and a path back to the cathode must be found. If the return path forms a large loop, the reverse recovery current can cause electromagnetic interference (EMI). It is important to note that if the diode is turned off more quickly (with a higher rate of current change during turn-off) and / or the diode current is higher before turn-off, the peak reserve recovery current will be larger.

[0013] use Figure 19The half-bridge circuit (shown as half-bridge) illustrates diode reverse recovery, where S1 and S2 are driven by complementary signals. Between t0 and t1, S1 is ON, S2 is OFF, and it is assumed that the output current through inductor L1 flows from node C to B to A through D1 (the body diode of S1). The inductor resists sudden changes in its current, so I(output) cannot change abruptly. During this interval, the voltage at node B is Vdc + Vd (Vdc is the DC input voltage, and Vd is the voltage drop across diode D1). At t1, S1 is off, S2 is on, pulling the voltage at node B from Vdc + Vd to ground and applying a negative voltage across D1. The state of D1 suddenly changes from forward conduction to reverse blocking, which generates a large reverse recovery current flowing from the cathode of D1 to the anode of D1 and into the drain of S2, such as... Figure 20 As shown (it illustrates) Figure 19 (Key waveforms in the circuit). S2 undergoes a hard turn-on because the S2 turn-on current flows from the drain to the source and before turn-on, S2 Vds = Vdc + Vd. During the turn-on period, the S2 current is the sum of the inductor current and the reverse recovery current of D1, further increasing the turn-on losses. It can be seen that after S2 turns on, I(D1) decreases from 20A to 0A and reverses direction to -12A during the reverse recovery period. The negative diode current flows through S2 during the turn-on period, resulting in a peak current of approximately 32A for S2 (instead of the 20A inductor current), significantly increasing the turn-on losses of S2. In addition, the reverse recovery process also produces significant losses in D1.

[0014] exist Figure 20 In this example, the peak diode reverse recovery current is very high at 12A. Such a high current, combined with the high di / dt during the recovery process, can lead to serious EMI problems. The peak reverse recovery current can be reduced if the current charging rate (di / dt) during diode turn-off is slower and / or the diode forward current is smaller before turn-off. In practice, it is easier to slow down the di / dt of the diode current during turn-off to reduce the diode reverse recovery effect.

[0015] As can be clearly seen from the above background, for a half-bridge configuration, hard turn-on and diode reverse recovery occur simultaneously, resulting in excessive turn-on losses and potential EMI problems.

[0016] soft switching

[0017] As an alternative to hard switching, soft switching is another way to turn a MOSFET on or off. Soft switching is superior to hard switching because it requires less power loss during MOSFET switching. One option for soft switching is zero-voltage switching (ZVS). ZVS requires the MOSFET current to flow from the source to the drain during switching. In other words, the MOSFET current needs to flow through its body diode before it turns on. For ZVS turn-on and turn-off, the MOSFET current can flow through its body diode, the MOSFET channel from S to D, or a combination of both during switching. The current distribution depends on the equivalent resistance of these paths. However, regardless of the current path, Vds is clamped to a small negative voltage during switching, making the switching power loss negligible. Furthermore, diode reverse recovery is avoided because there is no sudden turn-off of the MOSFET body diode under ZVS.

[0018] Given the high switching losses and diode reverse recovery in half-bridge or full-bridge configurations associated with hard turn-on, some systems are designed to ensure ZVS during turn-on and allow hard turn-off.

[0019] The discussion now turns to existing technology that utilizes zero-voltage switching to convert a hard-on MOSFET in a half-bridge configuration to a soft-on configuration. For example... Figure 21 The auxiliary resonant commutated pole shown (taken from RWDe Doncker and J.Plyons, “The Auxiliary Resonant Commutated Pole Converter,” Conference record of the 1990 IEEE Industry Applications Society Annual Meeting, Seattle, Washington, 1990, pp. 1228-1235, Vol. 2) is a known circuit that uses an auxiliary AC switch and an inductor Lr to achieve zero-voltage turn-on of MOSFETs S1 and S2 in a half-bridge configuration. The AC switch includes switches S3 and S4 and diodes D3 and D4. Capacitors C1 and C2 can be external capacitors or switch parasitic capacitors Coss. Two series capacitors are used to generate Vdc / 2 at node A.

[0020] The typical operating waveforms of the resonant current I(Lr) and the output current I(output) are in Figure 22 The diagram shows a typical waveform for ARCP. Due to the large inductance Lf and the current flowing into the load (positive direction), the output current I(out) is assumed to be constant.

[0021] exist Figure 22Between t0 and t1, all switches are open, no current flows through Lr, the output current I(out) flows through Lf to the load and returns through D2, which makes the voltage at node C one diode voltage drop (Vd) below ground. The voltage of S1 is thus Vdc + Vd. Turning on S1 during this interval will result in a hard turn-on and cause the reverse recovery current of diode D2 to flow through S1. This is similar to the case of the half-bridge configuration with hard turn-on described previously.

[0022] Conversely, the ARCP solution proposes to turn on S3 at t1 and keep S1 off until a later time. This causes the current I(Lr) to linearly increase from zero between t1 and t2 at a constant slope of (Vdc / 2 + Vd) / Lr. This is because between t1 and t2, I(Lr) < I(out), and forces additional output current to flow through D2, which keeps the voltage of node C at -Vd. Therefore, the voltage of inductor Lr is Vdc / 2 + Vd during its interval.

[0023] At t2, I(Lr) catches up with I(out) and D2 turns off. After D2 turns off, the voltage at node C effectively "floats", and Lr forms a resonant circuit with C1 and C2. The resonant circuit causes I(Lr) to continue to increase beyond I(out) and reach a peak current of I(out) + Ires, where Ires is:

[0024]

[0025] Between t2 and t3, I(Lr) is greater than I(Lf), and the additional Lr current starts to flow through D1 and returns through the top DC bus capacitor S3 and D4. S1 is turned on with zero voltage switching (ZVS). As a result, ARCP converts the hard turn-on into a zero voltage turn-on.

[0026] When I(Lr) drops below I(out) and the voltage across S1 starts to increase due to D2 conduction, the zero voltage turn-on window of S1 closes after t3. The window for turning on S1 with zero voltage may become narrow, and a detection circuit is usually required to sense when to turn on S1 to achieve ZVS. Larger C1 and C2 capacitances can also be used to increase the time difference between t2 and t3 by making Ires larger.

[0027] The basic principle of ARCP is to create a small window of ZVS to achieve soft switching. To capture the small ZVS window, an accurate detection circuit is required. There are other existing technologies that improve different aspects of ARCP, however the principle of creating the ZVS window remains the same. Summary of the Invention

[0028] The object of the present invention is to provide an improved soft switch.

[0029] In one aspect, the invention is a soft-switching subcircuit that forms part of or is used with a circuit including a bridging switch circuit, the soft-switching subcircuit being configured and operable to provide a variable current output that tracks the output current from the bridging switch circuit to generate a substantially zero current through at least one switching element of the bridging switch circuit to enable soft switching.

[0030] Optionally, the bridging switch circuit is either a half-bridge switch circuit or a full-bridge switch circuit.

[0031] Optionally, the full-bridge switching circuit includes two or more half-bridge switching circuits, each half-bridge switching circuit having soft switching enabled by a soft-switching sub-circuit and optionally by one or more additional soft-switching sub-circuits.

[0032] Optionally, the bridging switch circuit can be connected to the load, and at least a portion of the output current from the bridging switch circuit passes through the load.

[0033] Alternatively, the soft-switching sub-circuit and the bridging switch circuit can be powered by a common power supply, such as a DC power supply with a reference voltage rating.

[0034] Optionally, the soft-switching sub-circuit includes an inductor and at least one switch.

[0035] Optionally, the soft-switching sub-circuit includes first and second switching components, the first switching component being used to power a first terminal of the inductor with a reference voltage or a voltage derived from the reference voltage, and the second switching component being used to power the first terminal of the inductor with a zero voltage or a voltage derived from the zero voltage.

[0036] Optionally, the first switching component is connected between the node that can be powered by the reference voltage and the first end of the inductor, and the second switching component is connected between the first end of the inductor and ground.

[0037] Optionally, the first switching component of the soft-switching sub-circuit is used for soft switching of the first switching component of the bridging switch circuit, and the second switching component of the soft-switching sub-circuit is used for soft switching of the second switching component of the bridging switch circuit.

[0038] Optionally, in the bridging switch circuit: a first switching element is connected between a node that can be powered by a reference voltage and a node between the two switching elements of the bridging switch circuit, and a second switching element is connected between a node between the two switching elements of the bridging switch circuit and ground.

[0039] Optionally, the bridging switch circuit is a half-bridge switch circuit, and the first and second switching components of the bridging switch circuit form a half-bridge switch circuit.

[0040] Optionally, the bridging switch circuit is a full-bridge switch circuit including first and second half-bridge switch circuits, and the first and second switching components of the bridging switch circuit form the first half-bridge switch circuit.

[0041] Optional variable current outputs include: an acceleration current output when the absolute value of the variable current output is less than the absolute value of the output current from the bridge switching circuit, and a deceleration current output when the absolute value of the variable current output is greater than the absolute value of the output current from the bridge switching circuit.

[0042] Optionally, the variable current output includes alternation between accelerating current output and decelerating current output.

[0043] Optionally, the inductor has a first end connected to the at least one switching component of the soft-switching sub-circuit and a second end for connecting to a node between the two switching components of the bridging switch circuit.

[0044] Optionally, the variable current output is provided by the operation of the inductor in the following manner: when the output current from the bridge switch circuit exceeds the variable current output of the soft-switching sub-circuit, the voltage difference applied across the inductor is increased, causing the current output of the soft-switching sub-circuit to exceed the output current of the bridge switch circuit; and when the variable current output of the soft-switching sub-circuit exceeds the output current of the bridge switch circuit, the voltage difference applied across the inductor is decreased, causing the output current from the bridge switch circuit to exceed the current output of the soft-switching sub-circuit.

[0045] Optionally, the soft-switching subcircuit includes a switching element connected to a first terminal of the inductor, the switching element being configured to provide a reference voltage or a voltage derived from a reference voltage to the first terminal of the inductor. The two switching elements of the bridging switching circuit are: a first switching element comprising a diode connecting a node between the two switching elements of the bridging switching circuit to a node that can be powered by a reference voltage, and a second switching element comprising a diode connecting ground to a node between the two switching elements of the bridging switching circuit.

[0046] Optionally, the forward bias conduction of the diode of the second switching component causes the voltage at the second terminal of the inductor to decrease toward zero voltage or the voltage derived from zero voltage, thereby increasing the voltage difference applied across the inductor and causing the current output of the soft-switching sub-circuit to exceed the output current from the bridge switching circuit. Conversely, the forward bias conduction of the diode of the first switching component causes the voltage at the second terminal of the inductor to increase toward the reference voltage or the voltage derived from the reference voltage, thereby decreasing the voltage difference applied across the inductor and causing the output current from the bridge switching circuit to exceed the current output of the soft-switching sub-circuit.

[0047] Optionally, the at least one switching component includes a switching component connected to the first terminal of the inductor component, the switching component being configured to provide a voltage of zero voltage or derived from zero voltage to the first terminal of the inductor component, and the two switching components of the bridging switching circuit are: a first switching component including a diode connecting a node between the two switching components of the bridging switching circuit to a node that can be powered by a reference voltage, and a second switching component including a diode connecting ground to a node between the two switching components of the bridging switching circuit.

[0048] Optionally, the forward bias conduction of the diode of the second switching component causes the voltage at the second terminal of the inductor to decrease toward zero voltage or the voltage derived from zero voltage, thereby reducing the voltage difference applied across the inductor and causing the output current from the bridge switching circuit to exceed the current output of the soft switching sub-circuit. Conversely, the forward bias conduction of the diode of the first switching component causes the voltage at the second terminal of the inductor to increase toward the reference voltage or the voltage derived from the reference voltage, thereby increasing the voltage difference applied across the inductor and causing the current output of the soft switching sub-circuit to exceed the output current from the bridge switching circuit.

[0049] Optionally, the variable current output includes an accelerating current output when the absolute value of the variable current output is less than the absolute value of the output current from the bridge switching circuit, and a decelerating current output when the absolute value of the variable current output is greater than the absolute value of the output current from the bridge switching circuit.

[0050] Optionally, the variable current output includes alternation between accelerating current output and decelerating current output.

[0051] Optionally, the alternation includes switching from accelerating current output to decelerating current output when the absolute value of the varying current output exceeds the absolute value of the output current of the bridge switching circuit, and switching from decelerating current output to accelerating current output when the absolute value of the output current of the bridge switching circuit exceeds the absolute value of the varying current output.

[0052] Optionally, the accelerating current output is generated by an increased voltage applied across the inductor, and the decelerating current output is generated by a decreased voltage applied across the inductor.

[0053] Optionally, the soft-switching subcircuit includes a switching element connected to a first terminal of the inductor, the switching element being configured to provide a reference voltage or a voltage derived from a reference voltage to the first terminal of the inductor. The two switching elements of the bridging switching circuit are: a first switching element comprising a diode connecting a node between the two switching elements of the bridging switching circuit to a node that can be powered by a reference voltage, and a second switching element comprising a diode connecting ground to a node between the two switching elements of the bridging switching circuit.

[0054] Optionally, the diodes of the first switching component and the second switching component are operable such that: the forward bias conduction of the diode of the first switching component generates a decelerating current output by increasing the voltage at the second terminal of the inductor, thereby reducing the voltage difference applied across the inductor; and the forward bias conduction of the diode of the second switching component generates an accelerating current output by decreasing the voltage at the second terminal of the inductor, thereby increasing the voltage difference applied across the inductor.

[0055] Optionally, the soft-switching subcircuit includes a switching element connected to a first terminal of the inductor, the switching element being configured to provide a voltage of zero or derived from zero to the first terminal of the inductor, and the two switching elements of the bridging switching circuit are: a first switching element comprising a diode connecting a node between the two switching elements of the bridging switching circuit to a node that can be powered by a reference voltage, and a second switching element comprising a diode connecting ground to a node between the two switching elements of the bridging switching circuit.

[0056] Optionally, the diodes of the first switching component and the second switching component are operable such that: the forward bias conduction of the diode of the first switching component generates an accelerating current output by increasing the voltage at the second terminal of the inductor, thereby reducing the voltage difference applied across the inductor; and the forward bias conduction of the diode of the second switching component generates a decelerating current output by decreasing the voltage at the second terminal of the inductor, thereby reducing the voltage difference applied across the inductor.

[0057] Optionally, the inductor component is adapted to reduce the rate of change of current in the diodes of the first and second switching components of the bridging switching circuit, thereby reducing the peak reverse recovery current of the diodes.

[0058] Optionally, the reduced peak reverse recovery current reduces the energy consumed by the reverse recovery current, thereby reducing electromagnetic interference and / or enabling soft switching of the diode.

[0059] Optionally, the inductive component is or includes an inductor.

[0060] Alternatively, the capacitor and the inductor can be connected in series.

[0061] In another aspect, the present invention is a circuit comprising a bridging switch circuit and a soft-switching sub-circuit according to any of the foregoing statements.

[0062] Optionally, the bridging switch circuit is either a half-bridge switch circuit or a full-bridge switch circuit.

[0063] Optionally, the full-bridge switching circuit includes two or more half-bridge switching circuits, each half-bridge switching circuit having soft switching enabled by a soft-switching sub-circuit and optionally by one or more additional soft-switching sub-circuits.

[0064] Optionally, the switching component is a transistor or includes a transistor.

[0065] Optionally, these switching components are one or more of the following, or consist of one or more of the following:

[0066] MOSFET

[0067] BJT

[0068] IGBT

[0069] Optional soft switch enable is or includes soft turn-on.

[0070] Soft switching can be optionally enabled when the absolute value of the output current from the bridge switching circuit increases.

[0071] In another aspect, the present invention is a soft-switching sub-circuit or circuit, substantially as shown herein with reference to Figures 1-14 and Appendix Figure 23-26 As described.

[0072] As used in this specification, the term "comprising" means "consisting of at least part of...". When interpreting each expression containing the term "comprising" in this specification, features other than that term or those beginning with that term may also exist. Related terms such as "comprise" and "comprises" will be interpreted in the same manner. Unless the context clearly requires otherwise, throughout the specification and claims, the words "comprise," "comprising," etc., shall be interpreted as inclusive, meaning "including but not limited to," contrary to the meaning of exclusion or exhaustive.

[0073] In this specification, which has already referenced patent specifications, other external documents, or other sources of information, this is generally to provide context for discussing the features of this disclosure. Unless otherwise expressly stated, references to such external documents should not be construed as an admission that such documents or sources are part of the prior art or common general knowledge in the art within any scope of the claims.

[0074] The invention can also be broadly defined as any part, element, or feature that is individually or jointly referred to or indicated in any or all combinations of two or more of the stated parts, elements, or features in the description of this application. Wherein the foregoing description refers to components in whole or having known equivalents thereof, which are incorporated herein as if described separately.

[0075] For those skilled in the art, numerous changes can be made to the construction and a wide variety of embodiments and applications of the invention without departing from the scope of the invention as defined by the appended claims. The disclosure and description herein are purely illustrative and not intended to be limiting in any sense. When a specific integer having a known equivalent in the field to which this invention pertains is referred to herein, such known equivalents are considered to be incorporated herein as if separately stated. The invention encompasses the foregoing, and the following constructions, given only as examples, are also contemplated. Attached Figure Description

[0076] The embodiments will be described with reference to the following figures, in which:

[0077] Figure 1A Each of the -B diagrams shows an electronic circuit, which illustrates a bridging switch circuit in operation.

[0078] Figure 2A -C illustrates a general form of electronic circuit that demonstrates soft switching.

[0079] Figure 3A -B indicates the "tracking" of the current used to implement soft switching.

[0080] Figure 4A -F forms a flowchart illustrating the changes in circuit operation to achieve soft switching.

[0081] Figure 5A -C illustrates an electronic circuit that demonstrates another general form of soft switching.

[0082] Figure 6A -B indicates the "tracking" of the current used to implement soft switching.

[0083] Figure 7A -F forms a flowchart illustrating the changes in circuit operation to achieve soft switching.

[0084] Figure 8 This is an electronic circuit that illustrates a specific form of soft switching.

[0085] Figure 9 Line graphs of voltage and current waveforms observed during circuit operation are shown.

[0086] Figure 10 An inductor is shown.

[0087] Figure 11 An enlarged line graph of the voltage and current waveforms observed during circuit operation is shown.

[0088] Figure 12 It is another line graph of the voltage and current waveforms observed during circuit operation.

[0089] Figure 13 This is an electronic circuit that illustrates another specific form of soft switching.

[0090] Figure 14 Line graphs of voltage and current waveforms observed during circuit operation are shown.

[0091] Figure 15 A MOSFET switch is shown.

[0092] Figure 16 Line graphs showing the voltage and current waveforms observed when the diode reverse recovers are shown.

[0093] Figure 17 This is a line graph showing the switching losses of a MOSFET switch.

[0094] Figure 18 It is a diode used in reverse recovery.

[0095] Figure 19 This shows a circuit with a hard switch.

[0096] Figure 20 This is a line graph showing the voltage and current waveforms observed during hard switching.

[0097] Figure 21 This shows a circuit for soft switching.

[0098] Figure 22 This is a line diagram showing the form of a soft switch.

[0099] Figure 23 This is a line diagram illustrating another form of soft switching.

[0100] Figure 24 This is a line diagram illustrating another form of soft switching.

[0101] Figure 25A -D is a line graph showing the changes in circuit operation to implement soft switching.

[0102] Figure 26 This is a line diagram illustrating how to implement soft switching of the diode, a switching component. Detailed Implementation

[0103] Some of the soft-switching embodiments described herein involve the use of zero-current switching (ZCS). ZCS requires that the current through the switching component (e.g., a MOSFET) be very close to 0A during switching. It can also significantly reduce switching losses because the transient losses during switching are very low.

[0104] Some of the soft-switching embodiments described herein involve the use of zero-voltage switching (ZVS). ZVS requires that the voltage across the switching component (e.g., a MOSFET) be very close to 0V during switching. It can also significantly reduce switching losses because the transient losses during switching are very low.

[0105] Given the high switching losses and high power losses associated with hard-turning of the bridged switching circuitry and the high diode reverse recovery current, the embodiments described herein are designed to provide soft turn-on by ZCS and / or ZVS. Soft turn-off can also be implemented in some embodiments described herein, but this is not required. More generally, hard turn-off is permissible in the described embodiments because the power loss from hard turn-off is typically smaller than the power loss from hard turn-on.

[0106] Overview

[0107] Figure 1A -B shows an overview of electronic circuit 1, including bridging switch circuit 2 and (soft) switching sub-circuit 4. The (soft) switching sub-circuit 4 is configured to be used in conjunction with bridging switch circuit 2 and to reduce the switching power loss experienced by bridging switch circuit 2.

[0108] Bridge (switch) circuit 2 in Figure 1A -B is depicted as a (full) bridge circuit (H-bridge) comprising two half-bridge switching circuits 6 and 8. It includes four switching components 12, 14, 16, and 18. More specifically, half-bridge switching circuit 6 includes switching components 12 and 14, and half-bridge switching circuit 8 includes switching components 16 and 18. The bridging (switching) circuit 2 can be connected to a load 20. By switching components 12 and 18, current 21 can flow from... Figure 1A As shown in 21a, the current flows from left to right through the load 20, and through the connecting parts 14 and 16, the current 21b can flow from... Figure 1B As shown in 21b, the bridge switch circuit 2 can be operated from right to left through the load 20.

[0109] Although the bridging (switching) circuit 2 is shown in this specification as a (full) bridge circuit including two half-bridges 6, 8, it will be apparent to those skilled in the art that the bridging switch circuit can simply be a half-bridge switch circuit for supplying power to the load 20; that is, the bridging (switching) circuit is half-bridge switch circuit 6 (so that only switching components 12, 14 are used), and the half-bridge switch circuit 8 is not required (so that switching components 16, 18 are not required). For the avoidance of doubt, in this specification, "switching component" may be used interchangeably with "switch component".

[0110] Although the specification and figures only refer to one switching subcircuit 4 used on one half 6 of the (full) bridging circuit 2, the switching subcircuit 4 can also be used for soft-switching of the other half 8 of the (full) bridging circuit 2. Alternatively, a duplicated second switching subcircuit can be optionally used on the other half (half-bridging switching circuit 8) of the (full) bridging circuit 2. Alternatively, the other half 8 of the (full) bridging circuit 2 can be switched using a different soft-switching mechanism, or alternatively, a hard switch. More generally, if there is a (full) bridging circuit 2 comprising two or more half-bridging switching circuits, a switching subcircuit 4 can be used with the first half-bridging switching circuit 6, and the remaining half-bridging switching circuits can each be switched using any combination of the following: the same switching subcircuit 4 (already used in the first half-bridging switching circuit 6), a duplicated switching subcircuit (separate from the switching subcircuit 4), a different soft-switching mechanism, and / or being hard-switched. The half-bridging switching circuits can be connected in series, in parallel, or a combination of both.

[0111] Those skilled in the art will also recognize that, although the bridge switching circuit 2 is shown as a DC-AC inverter in this specification, the described embodiments can be used on any other type of bridge switching circuit, such as a DC-DC buck-boost converter or other alternatives.

[0112] General Embodiments

[0113] General embodiments of the invention will now be described.

[0114] The switching sub-circuit 4 can be considered as part of, or used in conjunction with, circuit 1 including the bridging switching circuit 2, and is used to minimize the power loss experienced by the bridging switching circuit 2. The bridging switching circuit 2 includes two switching components 12 and 14 and a node 26. The two switching components 12 and 14 and the load 20 are all connected to each other via node 26. Switching component 12 is connected between a node that can be powered by a reference voltage 28 and node 26. Switching component 12 also includes a diode 12a connected between nodes 26 and 28. Switching component 14 is connected between node 26 and ground node 30. Switching component 14 also includes a diode 14a connected between nodes 30 and 26. When switching component 12 is turned on and switching component 14 is turned off, current 21a can flow as follows: Figure 1A As shown, the load 20 passes from left to right. When switch 12 is open and switch 14 is closed, current 21b can flow as follows: Figure 1B As shown, load 20 passes from right to left.

[0115] The (soft) switching subcircuit 4 can be integrated with the rest of the electronic circuit 1, or it can be a standalone module coupled to the rest of the electronic circuit 1. The (soft) switching subcircuit 4 includes an inductor 32, two switching components 34 and 36, and a node 38. The inductor 32 and the switching components 34 and 36 are all connected to each other via node 38. The inductor 32 is used for the connection between node 26 and node 38. The switching component 34 is connected between node 26, which can be powered by a reference voltage 28, and node 38. The switching component 36 is connected between node 38 and ground node 30. Either switching component (34 or 36) can be used – using both switching components 34 and 36 is not required.

[0116] refer to Figure 2A-2C The bridging switch circuit 2, along with 5A-5C, can be connected to load 20. Load 20 can be anything, including resistive reactance. The impedance of load 20 can include inductive or capacitive resistance and / or reactance. Examples of load 20 can include, but are not limited to, inductors, LC (resonant) boxes, etc. The attached diagram describes load 20 as part of electronic circuit 1; however, this is not necessary, and load 20 can be part of a separate electronic circuit. Load 20 can optionally be part of bridging switch circuit 2, but this is not necessary. Load 20 provides current output 40 of bridging switch circuit 2, and for the remainder of the detailed description, current output 40 of bridging switch circuit 2 can also be referred to as one or more of the following: for simplicity, "current 40 through load 20" or "i 负载 In some embodiments, the current output 40 is variable. More specifically, the current output 40 can be varied such that the absolute value of the current 40 increases. For example, Figures 3A-3B It shows in Figure 2A-2CThe circuit diagram shows a changing (positive) current of 40 flowing from left to right. Figures 6A-6B It shows in Figures 5A-5C The circuit diagram shows a changing (negative) current of 40 flowing from right to left. Figures 3A-3B Both the 6A and 6B models show a variable current output of 40. More specifically, Figures 3A-3B The 6A-6B models show a change in the absolute value of the current output 40.

[0117] (Soft) Switching subcircuit 4 should be configured as desired and be operable to produce a varying current output (i 开关_补偿 )42, which tracks the output current 40 from the bridge switching circuit 2 (to the load 20) to generate substantially zero current through at least one switching element 12, 14 of the bridge switching circuit 2, thereby achieving soft switching. More specifically, the switching sub-circuit element 4 should ideally generate a varying current output including one or more acceleration sections 44 and one or more deceleration sections 46. 开关_补偿 )42, which allows the variable current output 42 to "track" the variable current output 40 (i 负载 ),like Figure 3A and 6A As shown, this is to achieve soft-on. The acceleration section 44 should rapidly change the current output (i... 开关_补偿 The acceleration section 42 causes the current output 42 to "exceed" the current output 40. The acceleration section 44 is generated by increasing the voltage difference applied across the inductor 32. In this specification, the acceleration section 44 can be interchanged with "accelerated (current) output," "accelerated (current) component," or "accelerated (current) mode." Similarly, the deceleration section 46 should slowly change the current output so that the current output 40 "exceeds" the current output 42. The deceleration section 46 is generated by decreasing the voltage difference applied across the inductor 32. In this specification, the deceleration section 46 can be interchanged with "decelerated (current) output," "decelerated (current) component," or "decelerated (current) mode."

[0118] The alternation between the acceleration section 44 and the deceleration section 46 means that the current output 42 (i 开关_补偿 In absolute value and direction, the current output from the bridge switching circuit is 40 (i) 负载 Basic matching. When this occurs, the current entering node 26 is substantially equal to the current leaving node 26, resulting in no or negligible (i.e., substantially zero) current flowing through switching components 12, 14. When this occurs, a near-zero current state (“near-zero current window”) 45 is generated for switching components 12, 14, providing the conditions required for “soft-on” of switching components 12, 14, thereby reducing power losses in the bridged switching circuit 2. Figure 3B and 6BAs shown, the (near)zero current window 45 remains open, meaning that soft switching can be achieved without the need for precise detection circuitry to determine when the (near)zero current window 45 is open. Eliminating the need for precise detection circuitry simplifies circuit fabrication and operation.

[0119] although Figure 2A-2C Figures 5A-5C show a switching sub-circuit 4 including two switching components 34 and 36, but having two switching components is not necessary for the switching sub-circuit. That is, the switching sub-circuit 4 can have an inductor component 32 and only one switching component, i.e., the switching sub-circuit 4 can have a switching component 34 or a switching component 36.

[0120] The two near-zero current states used to achieve "soft turn-on" will now be described. The first near-zero current state describes the case where the (soft) switching subcircuit 4 produces a varying (positive) current output 42 that "tracks" the current 40 flowing out of the bridge switching circuit (i.e., "tracks" the positive current output 40), such as... Figure 3A As shown. The second (near) zero current state describes the situation where the (soft) switching subcircuit 4 generates a changing (negative) current output 42, which "tracks" the current 40 flowing out of the bridge switching circuit (i.e., "tracks" the negative current output 40), as... Figure 6A As shown.

[0121] First (near) zero current state

[0122] Reference Figure 2A-2C The 4A-4F describes the first (near) zero current state. Figure 2A-2C The sequence in which switching components 34 and 12 are turned on is shown, i.e., if all switching components are turned off (see...). Figure 2A If the switch component 34 is turned on first (see...), then the switch component 34 will be turned on first. Figure 2B This allows the switch component 12 to be "softly" switched on (see...). Figure 2C ). Figure 4A A flowchart is shown, from stage A to stage E, illustrating how the switching subcircuit 4 interacts with the bridging switching circuit 2 when the switching component 34 is turned on to produce a (near) zero current state for the "soft" turn-on of the switching component 12. Figure 4B-4F Expand on each stage AE separately.

[0123] exist Figure 2A In this configuration, all switching components 12, 14, 16, 18, 34, and 36 are open. The current output 42 is zero, i.e., i 开关_补偿 =0, and in this case, the current output 40 is positive and increases sinusoidally. This means that the current output 40 is greater than the current output 42, i.e., i 负载 >i 开关_补偿According to Kirchhoff's Current Law, the total current entering node 26 is equal to the total current leaving node 26. For node 26 to operate according to Kirchhoff's Current Law, the positive and increasing current output 40 forward-biased diode 14a, allowing current to flow from ground node 30 to node 26 through diode 14a, where the current through diode 14a matches the absolute value and direction of current output 40. That is, the current through diode 14a is also positive and increasing, such that the current through diode 14a corresponds to current output 40. The forward bias conduction of diode 14a implies the existence of V0. D2 The voltage drop makes the voltage at node 26 -V. D2 Preferably, the voltage drop across diode 14a should be negligible, such that the voltage at node 26 is essentially zero volts (i.e., -V). D2 ≈0).

[0124] like Figure 4A and 4B Phase A shown describes what happens once the switching component 34 is in operation. Figure 2B The diagram illustrates what happens when the switch is turned on. Turning on the switch 34 allows current to flow from node 28 through the switch 34 to node 38, and from node 38 through the inductor 32 to node 26. This is because the voltage across the switch 34 drops to zero and the voltage at node 28 is at the reference voltage V. 基准 Therefore, the voltage at node 38 is also at the reference voltage V. 基准 If it has already been established, the voltage at node 26 is -V. D1 This means that a Vt is applied across the inductor component 32. 基准 +V D2 The (non-zero) voltage difference applied across inductor 32 causes a (linear) increase in the current through inductor 34, which allows the switching subcircuit 4 to generate an acceleration portion 44 of the positive current output 42, see [link to diagram]. Figure 3A And allows a positive current output of 42 to be compared with a current output of 40 (i 负载 The rate of increase is greater than the rate of increase, see Figure 3A This means that current output 42 eventually "catches up" with current output 40. As the current 42 through inductor 32 increases, the current 48 through diode 14a decreases accordingly at the same rate, and as current output 42 gets closer to current output 40, current 48 through diode 14a approaches zero. At this point, we discuss transition phase B.

[0125] like Figure 4A and 4CPhase B, as shown, describes what happens after Phase A once current output 42 “catches up” with current output 40. When current output 42 finally “catches up” with current output 40, the current 48 through diode 14a reaches zero, and diode 14a turns off, stopping the forward bias conduction of diode 14a. The turn-off of diode 14a results in a reverse recovery current 50, in which diode 14a instantaneously conducts current from node 26 to ground node 30. According to Kirchhoff's current law, the reverse recovery current through diode 14a is compensated by the increase in current through inductor 32, making the total current entering node 26 equal to the total current leaving node 26. The increase in current through inductor 32 means that the current output 42 of switching subcircuit 4 “exceeds” the current output 40, making current output 42 more positive than current output 40, as... Figure 3A and 3B As shown.

[0126] As already discussed, breaking the forward bias conduction of diode 14a (from ground node 30 to node 26) causes a reverse recovery current 50, where diode 14a conducts current from node 26 to ground node 30 in the reverse bias direction. The reverse recovery current 50 undesirably generates electromagnetic interference and undesirably causes a “hard” disconnection of diode 14a (which would occur if switching element 14 were “hard” turned on). Inductor element 32 helps reduce unwanted electromagnetic interference and / or achieve a “softer” disconnection of diode 14a. This will now be explained. When the current through inductor element 32 changes, the rate of change cannot change instantaneously (due to the nature of inductor impedance). That is, the rate of change of current 42 through inductor element 32 is limited to a finite value. This means that in phase B, the current 42 through inductor element 32 changes gradually (as opposed to instantaneously). The rate of change of the current 42 through the inductor 32 also reduces the rate of change of the reverse recovery current 50 60a to the rate of change of the current 60b (according to Kirchhoff's current law). Figure 26 As shown. By reducing the variation of the reverse recovery current 50 to a gradual rate of change 60a,b, the peak reverse recovery current 62a is also reduced by 62b. The reduction of the peak reverse recovery currents 62a,b means that the energy consumed by the reverse recovery current 50 is reduced, such as by the energy consumed by the reverse recovery current 50. Figure 26 The reduced regions 64a and 64b covered by the curves are visually represented. The reduction in energy 64a and 64b consumed by the reverse recovery current 50 means less electromagnetic interference is generated, and a "softer" disconnection of diode 14a is achieved.

[0127] Furthermore, in stage B, current outputs 40 and 42 are substantially matched. Substantial matching of the current outputs means that, according to Kirchhoff's current law, there is (near) zero current flowing through switching element 12. Therefore, the substantial matching of the current outputs creates a (near) zero current switching window 45 for switching element 12, potentially allowing it to be soft-closed in stage B. As will be explained below in stage CE, the substantial matching of the output currents is maintained, thus maintaining a (near) zero current switching window 45 for switching element 12, potentially allowing it to be soft-closed in stage CE as well. That is, a (near) zero current switching window 45 is provided forward from stage B for the potential soft-close of switching element 12. The discussion now turns to stage C.

[0128] like Figure 4A and 4D Phase C, as shown, describes what happens after phase B once current output 42 “exceeds” current output 40. At this point, diode 12a is forward biased, causing the overcurrent from current output 42 to be conducted through diode 12a to node 28, causing node 26 to operate according to Kirchhoff's current law. The current conduction from node 26 to node 28 through diode 12a implies the presence of V0. D1 The voltage drop means that the voltage at node 26 is now from -V D2 Increase to V 基准 +V D1 The voltage at node 26 increases to V. 基准 +V D1 This means that the voltage drop across the inductor component changes from V 基准 +V D2 Reduce to -V D1 Since it is desirable to design the diode so that the voltage drop across it is negligible (i.e., essentially zero), it is reasonable for someone skilled in the art to assume that diode 12a (-V D1 The voltage drop across the terminals is (essentially) zero, which means that the voltage drop across the switching component 12 is also (essentially) zero. This implies the existence of a (near) zero voltage switching window 43 for "softly" switching on the switching component 12 during phase C, see [reference]. Figure 3B Furthermore, the (virtually) zero voltage drop across diode 12a means that the voltage drop across the inductor is also (virtually) zero, causing the current 42 through inductor 32 to "flatten" and "level off." This allows the decelerating portion 46 of the positive current output 42 of the switching subcircuit 4 to be generated, see... Figure 3A And allows the current output 40 to increase at a greater rate than the current output 42, see Figure 3BThe deceleration section 46 prevents the current output 42 from deviating from the current output 40, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides the switching unit 12 with a (near) zero current switching window 45, allowing for possible soft-switching during stage C. Eventually, the current output 40 "catches up" with and "exceeds" the current output 42. At this point, the discussion shifts to stage D.

[0129] like Figure 4A and 4E Phase D, as shown, describes what happens once the current output 40 "exceeds" the current output 42 after phase C. When the current output 40 exceeds the current output 42, diode 14a needs to be forward biased to provide additional current matching the current output 40. To forward bias diode 14a, the voltage at node 26 needs to be increased from V0... 基准 +V D1 Reduce to -V D2 However, once the voltage at node 26 begins to decrease to -V... D2 The voltage drop at node 26 means that the voltage difference across inductor 32 starts to drop from -V. D1 To V 基准 +V D2 The increase in the voltage difference across inductor 32 leads to an increase in the current 42 through inductor 32, which allows the accelerating portion 44 of the switching subcircuit 4 to generate a positive current output 42, see [link to relevant documentation]. Figure 3A And allows a positive current output of 42 to be compared with a current output of 40 (i 负载 The rate of increase is greater than the rate of increase, see Figure 3B Acceleration section 44 prevents current output 40 from deviating from current output 42, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides switching component 12 with a (near) zero current switching window 45, allowing for possible soft-switching during stage D. Eventually, current output 42 "catches up" with current output 40 and "exceeds" it. At this point, the discussion shifts to stage E.

[0130] like Figure 4A and 4F Phase E, as shown, describes what happens once the current output 42 "exceeds" the current output 40 after phase D. When the current output 42 exceeds the current output 40, diode 12a needs to be forward biased to provide additional current matching the current output 42. To forward bias diode 12a, the voltage at node 26 needs to be increased to V0. 基准 +V D1 Increase. However, once the voltage at node 26 starts to increase towards -V 基准 +V D1The increase in voltage at node 26 means that the voltage difference across inductor 32 begins to increase from V. 基准 +V D2 To-V D1 The voltage difference across inductor 32 decreases, causing the current 42 flowing through inductor 32 to "flatten," which allows the deceleration section 46 of the switching subcircuit 4 to generate a positive current output 42, see... Figure 3A And allows current output of 40 (i 负载 It increases at a rate greater than the rate at which the current output increases by 42, see Figure 3B The deceleration section 46 prevents the current output 42 from deviating from the current output 40, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides the switching unit 12 with a (near) zero current switching window 45, allowing it to be soft-switched during stage E. Eventually, the current output 42 "catches up" with and "exceeds" the current output 40. At this point, the discussion returns to stage D as described above.

[0131] The alternation between acceleration section 44 (stages AC and E) and deceleration section 46 (stage D) ensures that the current output 42 of the switching sub-circuit 4 can "track" the current output 40, making the difference between the current output 40 and the current output 42 essentially zero. See [link to relevant documentation]. Figure 3A , 3B And 4A. "Tracking" produces, for example Figure 3B The (near) zero current window 45 shown is used for switching component 12 to, as Figure 2C The minimum power loss shown is "soft" switched on. Furthermore, due to the (near) zero current window 45... Figure 3B As shown, it remains open, thus enabling soft-on without the need for precise detection circuitry to detect when the (near) zero current window 45 opens.

[0132] It will be apparent to those skilled in the art that it is sufficient for the switch sub-circuit 4 to have an inductor 32 and a switch 34. It is not necessary for the switch sub-circuit 4 to also have a switch 36.

[0133] Second (near) zero current state

[0134] Reference Figures 5A-5C 7A-7F describes the second (near) zero current state. Figures 5A-5C The sequence in which switching components 36 and 14 are turned on is shown, i.e., if all switching components are turned off (see...). Figure 5A If the switch component 36 is turned on first (see...), then the switch component 36 will be turned on first. Figure 5B This allows the switch component 14 to be "softly" switched on (see...). Figure 5C ). Figure 7AA flowchart is shown, from stage A to stage E, illustrating how the switching subcircuit 4 interacts with the bridging switching circuit 2 when the switching component 36 is turned on to produce a (near) zero current state for the "soft" turn-on of the switching component 14. Figure 7B-7F Expand on each stage AE separately.

[0135] exist Figure 5A In this configuration, all switching components 12, 14, 16, 18, 34, and 36 are open. The current output 42 is zero, i.e., i 开关_补偿 =0, and in this case, the current output 40 is negative (i.e., the current 40 through the load 20 is 0). Figure 5A (The current flows from right to left) and decreases sinusoidally. That is, current output 40 enters node 26. This means that current output 40 is more negative than current output 42, i.e., i 负载 >i 开关_补偿 According to Kirchhoff's Current Law, the total current entering node 26 is equal to the total current leaving node 26. To make node 26 operate according to Kirchhoff's Current Law, the negative and decreasing current output 40 forward-biased diode 12a, allowing current to flow from node 26 through diode 12a to node 28, where the current through diode 12a matches the absolute value and direction of current output 40. That is, the current through diode 12a is also negative and decreasing, such that the current through diode 12a corresponds to current output 40. The forward bias conduction of diode 12a implies the existence of V0. D1 The voltage drop makes the voltage at node 26 V. 基准 +V D1 Preferably, the voltage drop across diode 14a should be negligible, such that the voltage at node 26 is essentially at the reference voltage (i.e., V). 基准 +V D1 ≈V 基准 ).

[0136] like Figure 7A and 7B Phase A shown describes what happens once the switching component 36, as... Figure 5B This illustrates what happens when the circuit is switched on. The switching of component 36 allows current to flow from node 26 through inductor 32 to node 38, and from node 38 through switching component 36 to ground node 30. Since the voltage across switching component 36 drops to zero and the voltage at ground node 30 is zero volts, the voltage at node 38 is also grounded to zero volts. As already established, the voltage at node 26 is V. 基准 +V D1 This means that a Vt is applied across the inductor component 32. 基准 +V D1The (non-zero) voltage difference applied across inductor 32 causes a (linear) decrease in the current through inductor 36, which allows the switching subcircuit 4 to generate the accelerating portion 44 of the negative current output 42, see Figure 6A And allows negative current output of 42 to be compared with current output of 40 (i 负载 The rate of decrease is greater than the rate of decrease, see Figure 6B This means that current output 42 eventually "catches up" with current output 40. As the current 42 through inductor 32 decreases, the current 52 through diode 12a increases accordingly at the same rate, and as current output 42 gets closer to current output 40, current 52 through diode 12a approaches zero. At this point, we discuss transition phase B.

[0137] like Figure 7A and 7C Phase B, as shown, describes what happens after Phase A once current output 42 “catches up” with current output 40. When current output 42 finally “catches up” with current output 40, the current 48 through diode 12a reaches zero, and diode 12a turns off, stopping the forward bias conduction of diode 12a. The turn-off of diode 12a results in a reverse recovery current 54, which instantaneously conducts current from node 28 to node 26. According to Kirchhoff's current law, the reverse recovery current through diode 12a is compensated by the decrease in current through inductor 32, making the total current leaving node 26 equal to the total current entering node 26. The decrease in current through inductor 32 means that the current output 42 of the switching subcircuit 4 “exceeds” the current output 40, making current output 42 more negative than current output 40, as... Figure 6A and 6B As shown.

[0138] As already discussed, breaking the forward bias conduction of diode 12a (from node 26 to node 28) causes a reverse recovery current 54, where diode 12a conducts current from node 28 to node 26 in the reverse bias direction. This reverse recovery current 54 undesirably generates electromagnetic interference and undesirably causes a “hard” disconnection of diode 12a (which would occur if switching element 12 were “hard” turned on). Inductor element 32 helps reduce unwanted electromagnetic interference and / or achieve a “softer” disconnection of diode 12a. This will now be explained. When the current through inductor element 32 changes, the rate of change cannot change instantaneously (due to the nature of inductor impedance). That is, the rate of change of current 42 through inductor element 32 is limited to a finite value. This means that in phase B, the current 42 through inductor element 32 changes gradually (as opposed to instantaneously). The rate of change of the current 42 through the inductor 32 also reduces the rate of change of the reverse recovery current 54 60a to the rate of change of 60b (according to Kirchhoff's current law). Figure 26 As shown. By reducing the variation of the reverse recovery current 54 to a gradual rate of change 60a,b, the peak reverse recovery current 62a is also reduced by 62b. The reduction of the peak reverse recovery currents 62a,b means that the energy consumed by the reverse recovery current 54 is reduced, such as by the energy consumed by the reverse recovery current 54. Figure 26 The reduced regions 64a and 64b covered by the curves are visually represented. The reduction in energy 64a and 64b consumed by the reverse recovery current 54 means less electromagnetic interference is generated, and a "softer" disconnection of diode 12a is achieved.

[0139] Furthermore, in stage B, current outputs 40 and 42 are substantially matched. Substantial matching of the current outputs means that, according to Kirchhoff's current law, there is (near) zero current flowing through switching element 14. Therefore, the substantial matching of the current outputs creates a (near) zero current switching window 45 for switching element 14, potentially allowing it to be soft-closed in stage B. As will be explained below in stage CE, the substantial matching of the output currents is maintained, thus maintaining a (near) zero current switching window 45 for switching element 14, potentially allowing it to be soft-closed in stage CE as well. That is, a (near) zero current switching window 45 is provided forward from stage B for the potential soft-close of switching element 14. The discussion now turns to stage C.

[0140] like Figure 7A and 7D Phase C, as shown, describes what happens once current output 42 “exceeds” current output 40 after phase B. At this point, diode 14a is forward biased, causing overcurrent from current output 42 to conduct from ground node 30 through diode 14a, causing node 26 to operate according to Kirchhoff's current law. The current conduction from ground node 30 to node 26 through diode 14a implies the presence of V0.D2 The voltage drop means that the voltage at node 26 is now from V 基准 +V D1 Reduce to -V D2 The voltage at node 26 decreases to -V. D2 This means that the voltage drop across the inductor component changes from V 基准 +V D1 Reduce to -V D2 Since it is desirable to design the diode so that the voltage drop across the diode is negligible (i.e., essentially zero), it will be assumed to those skilled in the art that diode 14a (-V D2 The voltage drop across the terminals (which is essentially zero) is reasonable, meaning that the voltage drop across the switching element 14 is also (essentially) zero. This implies the existence of a (near) zero-voltage switching window 43 for "softly" switching on the switching element 14 during phase C, see [reference]. Figure 6B Furthermore, the (virtually) zero voltage drop across diode 14a means that the voltage drop across the inductor is also (virtually) zero, causing the current 42 through inductor 32 to "flatten" and "level off." This allows the decelerating portion 46 of the switching subcircuit 4 to generate a negative current output 42, see [link to relevant documentation]. Figure 6A And it allows the current output 40 to decrease at a greater rate than the rate at which the current output 42 decreases, see [reference]. Figure 6B The deceleration section 46 prevents the current output 42 from deviating from the current output 40, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides the switching unit 14 with a (near) zero current switching window 45, allowing for possible soft-switching during stage C. Eventually, the current output 40 "catches up" with and "exceeds" the current output 42. At this point, the discussion shifts to stage D.

[0141] like Figure 7A and 7E Phase D, as shown, describes what happens after phase C once current output 40 “exceeds” current output 42. When current output 40 exceeds current output 42, diode 12a needs to be forward biased to conduct in order to provide additional current matching current output 40. To forward bias diode 12a, the voltage at node 26 needs to be from -V... D2 Increase to V 基准 +V D1 However, once the voltage at node 26 begins to increase to V... 基准 +V D1 The voltage rise at node 26 means that the voltage difference across inductor 32 begins to rise from -V. D2 To V 基准 +V D1Increase. The (non-zero) voltage difference applied across inductor 32 causes a (linear) decrease in the current through inductor 36, which allows the switching subcircuit 4 to generate the accelerating portion 44 of the negative current output 42, see Figure 6A And allows negative current output of 42 to be compared with current output of 40 (i 负载 The rate of decrease is greater than the rate of decrease, see Figure 6B Acceleration section 44 prevents current output 40 from deviating from current output 42, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides switching component 14 with a (near) zero current switching window 45, allowing for possible soft-switching during stage D. Eventually, current output 42 "catches up" with current output 40 and "exceeds" it. At this point, the discussion shifts to stage E.

[0142] like Figure 4A and 4F Phase E, as shown, describes what happens once the current output 42 "exceeds" the current output 40 after phase D. When the current output 42 exceeds the current output 40, diode 14a needs to be forward biased to conduct in order to provide additional current matching the current output 42. To forward bias diode 14a, the voltage at node 26 needs to be shifted towards -V... D2 Decrease. However, once the voltage at node 26 starts to move towards -V D2 The decrease in voltage at node 26 means that the voltage difference across inductor 32 begins to decrease from V. 基准 +V D1 Towards -V D2 The voltage difference across inductor 32 decreases, causing the current 42 flowing through inductor 32 to "flatten," which allows the deceleration portion 46 of the switching subcircuit 4 to generate a negative current output 42, see [link to relevant documentation]. Figure 6A And allows current output of 40 (i 负载 It decreases at a rate greater than the rate at which the current output decreases by 42, see Figure 6B The deceleration section 46 prevents the current output 42 from deviating from the current output 40, ensuring that the current outputs remain substantially matched. This substantial matching of the current outputs provides the switching unit 14 with a (near) zero current switching window 45, allowing it to be soft-switched during stage E. Eventually, the current output 42 "catches up" with and "exceeds" the current output 40. At this point, the discussion returns to stage D as described above.

[0143] The alternation between the acceleration section 44 (stages AC and E) and the deceleration section (stage D) ensures that the current output 42 of the switching sub-circuit 4 can "track" the current output 40, making the difference between the current output 40 and the current output 42 essentially zero. See [link to relevant documentation]. Figure 6A , 6BAnd 7A. "Tracking" produces, for example Figure 6B The (near) zero current window 45 shown is used for switching component 14 to... Figure 5C The minimum power loss is shown as a "soft" turn-on. Furthermore, since the (near) zero current window 45 remains open, as... Figure 6B As shown, soft switching can be achieved without the need for precise detection circuitry to detect when the (near) zero current window 45 opens.

[0144] It will be apparent to those skilled in the art that it is sufficient for the switch sub-circuit 4 to have an inductor 32 and a switch 36. It is not necessary for the switch sub-circuit 4 to also have a switch 34.

[0145] Variations of the general embodiment

[0146] Switching components 12, 16, 14, 18, 34, and 36 are preferably MOSFETs, although other types of switches may be used, and the use of MOSFETs is not required. For example, BJTs and IGBTs may be used in any of the switching components 12, 16, 14, 18, 34, and 36.

[0147] Although diodes 12a and 14a are described as inherent diodes of switching components 12 and 14, those skilled in the art may also consider diodes 12a and 14a as representing diodes external to the switching components. That is, diodes 12a and 14a can be diodes external to switching components 12 and 14, respectively. Preferably, if diodes 12a and 14a are external, they are placed in parallel with switching components 12 and 14, respectively. Those skilled in the art will recognize that BJTs do not have intrinsic body diodes; therefore, if BJTs are chosen as switching components 12 and 14, diodes 12a and 14a will be external to BJT switches 12 and 14, respectively.

[0148] Inductor 32 is preferably an inductor, although its use is not mandatory. For example, those skilled in the art will recognize that any other circuit component can be used as a substitute, provided it can be modified to an equivalent inductor. Furthermore, any component capable of limiting the rate of change of current to a mathematically finite value can be considered inductor 32. That is, inductor 32 can be considered any component that, when operating as a single component, prevents instantaneous changes in current. Inductor 32 can also be any group of components that, when operating together, function to prevent instantaneous changes in current. Specific Implementation

[0150] Specific embodiments of the invention will now be described. These embodiments cover two specific circuit arrangements (“Circuit A” and “Circuit B”) and two different control methods (“Control Method A” and “Control Method B”), but all conform to the general principle of generating a (near) zero current state to achieve “soft-on” switching components 12, 14, as mentioned in the general embodiments described above. The specific embodiments differ from each other in how switching components 12, 14 are disconnected.

[0151] Three specific implementations will now be discussed:

[0152] First embodiment: This embodiment uses circuit A and control method A. In circuit A, a single inductor is used as inductor component 32. In control method A, switch component 34 is turned off before switch component 12 is turned off, and switch component 36 is turned off before switch component 14 is turned off.

[0153] Second embodiment: This embodiment uses circuit A and control method B. This second embodiment is similar to the first embodiment, except that it uses a different control method. That is, the difference from the first embodiment is that switch components 12 and 34 are (approximately) disconnected simultaneously, while switch components 14 and 36 are (approximately) disconnected simultaneously.

[0154] Third embodiment: This embodiment uses circuit B and control method B. This third embodiment is similar to the second embodiment, except that it uses a different circuit. That is, the difference from the third embodiment is the single capacitor (as inductor component 32) placed in series with a single inductor.

[0155] First Embodiment

[0156] Reference Figure 8 The circuit A shown here describes the first embodiment, and the circuit includes the following features:

[0157] Circuit 101, which corresponds to Circuit 1

[0158] • Bridge switch circuit 102, which corresponds to bridge switch circuit 2

[0159] • Switching sub-circuit 104 (which may be referred to as "auxiliary circuit") corresponds to switching sub-circuit 4

[0160] • The left half-bridge is 106, which corresponds to half-bridge circuit 6.

[0161] The right half-bridge is connected to 108, which corresponds to half-bridge circuit 8.

[0162] MOSFETs 112 (“(Main Switch) S1”), 114 (“(Main Switch) S2”), 116 (“S3”), 118 (“S4”), 134 (“(Auxiliary Switch) S5”), and 136 (“(Auxiliary Switch) S6”) correspond to switching components 12, 14, 16, 18, 34, and 36, respectively.

[0163] Diodes 112a (“(Diode)D1”) and 114a (“(Diode)D2”) correspond to diodes 12a and 14a, respectively.

[0164] • Resonant circuit 120, which corresponds to load 20

[0165] • Nodes 126 (“(Node)M”), 128, 130, and 138 (“(Node)K”) correspond to nodes 26, 28, 30, and 38, respectively.

[0166] • Inductor (“Ls”) 132, which corresponds to inductor component 32

[0167] • The “I (out)” current is 140, which corresponds to an output current of 40.

[0168] • “I(Ls)” current 142, which corresponds to output current 42.

[0169] • “I(D2)” current 148, which corresponds to current 48

[0170] • “I(D1)” current 152, which corresponds to current 52.

[0171] like Figure 8 As shown, adding an auxiliary circuit including a half-bridge (S5, S6) and an inductor Ls connected in parallel with the left half-bridge provides one or more of the following advantages:

[0172] 1) It converts the hard-on of the left half-bridge to near-zero current connection.

[0173] 2) The auxiliary switches (S5, S6) are turned on smoothly with zero current.

[0174] 3) It can reduce the Vds of the left half-bridge main switch in the part before it is turned on, which helps to further reduce the switching loss.

[0175] 4) Its control algorithm is very simple and does not require sensors or feedback.

[0176] 5) During a portion of the resonant cycle, conduction losses are distributed between the auxiliary and main switches, which helps to reduce the maximum switching temperature.

[0177] 6) It uses a shared inductor Ls to slow down the body diode turn-off current in the left half-bridge, thereby reducing the diode reverse recovery effect and improving EMI performance.

[0178] The unique characteristic of this circuit 201 is due to the sinusoidal load current I(out). Figure 9 The key operating waveforms of circuit A with control method A are shown, where Vdc = 450V and Ls = 6uH.

[0179] Before t0, S1, S5, and S6 are off, while S2 is on. The output current I(out) flowing toward the resonant circuit can flow through S2 (from S to D) and its body diode D2. The current distribution depends on the on-state resistance of the MOSFET and the forward voltage drop of D2.

[0180] At t0, S2 is off. The increasing I(out) is forced to flow through D2, still flowing towards the resonant circuit, which makes the voltage at node M equal to the voltage drop across the diode below ground. As a result, Vds of S1 is Vdc + Vd. S1 should not be turned on now, as this would result in a hard turn-on of S1 and an active diode reverse recovery of D2, leading to excessive turn-on losses and EMI problems for S1.

[0181] Further explanation requires understanding the relationship between the voltage and current of an inductor, which is described as follows:

[0182]

[0183] The reference directions for inductor voltage and current are as follows: Figure 10 As shown.

[0184] This equation states that the rate of change of inductor current is determined by v / L. For example, a positive inductor voltage causes its current to increase at a rate of v / L, and if its voltage is zero, the inductor current remains constant. This equation also implies that the inductor current cannot change instantaneously, as this would result in an impossible infinite inductor voltage.

[0185] At t1, S5 is turned on. Its current I(S5), which is also the inductor current I(Ls), rises linearly from zero at a constant rate of (Vdc + Vd) / Ls. Therefore, S5 is turned on with zero current. The ramp rate is due to the fact that when S5 is turned on, the voltage V(K) at the left terminal of Ls is at Vdc, and due to conduction from D2, the voltage V(M) at the right terminal of Ls is at -Vd. As a result, the voltage V(K, M) of Ls is Vdc + Vd, and I(Ls) rises according to the equation at (Vdc + Vd) / Ls.

[0186] The events that occur between t1 and t2 explain the operation of this embodiment, and in Figure 11 Magnification and reproduction Figure 11 It shows Figure 9 A magnified view.

[0187] In ta( Figure 11 When I(Ls) = I(out), the current of D2 becomes zero and is soft-turned off because the rate of decrease of the D2 current is controlled by the rate of increase of I(Ls), which is much slower than the sudden diode turn-off in the hard-on case. This slow diode turn-off reduces the peak reverse recovery current of D2 to a smaller value and significantly improves EMI performance.

[0188] The small reverse recovery current of D2 flows in the positive direction of I(S2) and through the loop consisting of D2, Cbus, S5, and Ls. As a result, it increases the existing I(Ls), causing I(Ls) to increase beyond I(out) after ta. Figure 11 ).

[0189] After ta, the inductor voltage V(K, M) begins to decrease from Vdc+Vd towards 0V and reaches 0V at tb, which makes I(Ls) remain constant after tb. The operation between ta and tb is explained as follows. During this interval, the inductor voltage decreases because the voltage V(M) at the right inductor terminal is driven to Vdc by the additional inductor current (due to diode reverse recovery), while the voltage V(K) at the left end of the inductor is fixed at Vdc (due to S5 being on). V(M) is driven to Vdc because the additional inductor current needs to flow through D1. When D1 conducts, V(M) is pulled towards Vdc. The slow rise of V(m) is due to S2Coss charging Vdc and S1 Coss discharging to 0V.

[0190] Between tb and tc, the inductor current is still higher than the output current, and D1 conducts. Theoretically, S1 can be turned on with zero voltage during this interval. However, in practice, this zero-voltage interval may be too short to turn on the MOSFET, especially when the output current rises rapidly. This is evident in... Figure 11 This is illustrated by comparing this interval with Vgs during the turn-on period of S1. It can be seen that the window between tb and tc is much shorter than the turn-on time of S1. Therefore, even if the detection circuit can sense when tb occurs, the MOSFET cannot be fully turned on at zero voltage. Furthermore, the window may depend heavily on how quickly the output current rises at ta, which is theoretically difficult to predict and can change significantly with operating conditions. In summary, zero-voltage turn-on is difficult to achieve for this type of operation where the output current rises rapidly during the MOSFET's conduction time.

[0191] However, this type of operation provides near-zero current turn-on for the main MOSFETs (S1 and S2). This is achieved by checking... Figure 11 This can be explained by what happened after TC.

[0192] After tc, the output current I(out) catches up with and exceeds the constant inductor current. The only path for the additional output current is through D2, since S1 remains off. However, before the additional output current can flow through D2, V(M) needs to be pulled to ground to forward bias D2. However, this is not easy to do because 1) pulling V(M) to ground requires the discharge of Coss of S2 and the charging of Coss of S1, which takes time and energy. 2) as V(M) begins to decrease from Vdc, the inductor voltage begins to increase, subsequently causing I(Ls) to increase again towards I(out). This reduces the difference between I(Ls) and I(out), which slows down the charging of Coss of S1, the discharging of Coss of S2, and the movement of V(m) towards ground. 3) Once I(Ls) exceeds I(out) again, the additional I(Ls) current reverses the process by starting to charge the Coss of S2 and discharging the Coss of S1, shifting V(m) toward Vdc. This mechanism behaves like negative feedback, causing I(Ls) to automatically track I(out) and preventing V(m) from falling completely to ground.

[0193] Because of this mechanism, as long as I(output) rises, the difference between I(Ls) and I(output) automatically remains very small. This condition creates a very large window for S1 to be switched on with near-zero current, and potentially eliminates the need for any sensing or control circuitry. This large zero-current window... Figure 9 The window is shown between t1 and t2. Theoretically, if S1 is not opened before t3, the window can be extended to approximately t3 ( Figure 9 ).

[0194] Due to the negative feedback mechanism, the voltage of S1 can be very low when it is turned on, which helps to reduce its switching losses.

[0195] Back Figure 9 MOSFET S1 turns on with near-zero current at t2. Between t2 and t3, both S5 and S1 are turned on, and the voltage across Ls is zero, which keeps its current essentially constant. Since the output current is still rising during this cycle, additional output current begins to flow through S1. S5 and S1 now share the output current during this interval, which helps to distribute conduction losses.

[0196] At t3, S5 is off. The positive inductor current is forced to flow through D6, and it circulates in a loop including Ls, Si, Cbus, and D6. This pulls the voltage at node K toward -Vd and applies -Vdc-Vd across Ls. This forces I(Ls) to decrease toward zero with a negative slope of -(Vdc+Vd) / Ls.

[0197] At t4, due to the fact that the current in S1 changes direction before turn-off and flows from the drain of S1 to its source, the main switch S1 turns off with zero voltage. As mentioned earlier, Vds of S1 is clamped by its body diode during turn-off. This zero-voltage turn-off characteristic is attributed to... Figure 8 The resonant characteristics of the circuit in the image.

[0198] This completes the full operating cycle. Since the voltage and current waveforms of S2 and S6 are similar and operate on the same principle, they will not be described here. S2 and S6 also turn on with near-zero current, while S2 turns off with zero voltage.

[0199] The value of Ls is a critical parameter and needs to meet two conditions: 1) It needs to be large enough to suppress diode reverse recovery by slowing down the diode turn-off current. 2) It should be small enough to achieve a sufficiently large di / dt for the Ls current so that I(Ls) can keep up with the output current, which is the condition required for zero-current switching of the left half-bridge MOSFET.

[0200] Second Embodiment

[0201] Reference Figure 8 Circuit A shown describes this second embodiment. This second embodiment is similar to the first embodiment, except that control method B is used instead of control method A. That is, the difference from this second embodiment is that switching components 12 and 34 are turned off (approximately) simultaneously, while switching components 14 and 36 are turned off (approximately) simultaneously. This difference will now be explained in more detail.

[0202] control Figure 8 Another way to control the circuit (i.e., control method B) is to turn off the auxiliary switch simultaneously with the main switch. This means that S1 and S5 turn off at (approximately) the same time, and S2 and S6 turn off at (approximately) the same time.

[0203] The second control method differs from the first control method in the following ways:

[0204] 1. The auxiliary switch can be turned on with zero voltage, while the auxiliary switch of control method A is turned on with zero current.

[0205] 2. Compared with control method A, this method also allows the output current to share a longer cycle between the auxiliary and main MOSFETs.

[0206] Figure 12 Some key operating waveforms of this control method are shown.

[0207] Before ta, S2 and S6 are turned on. The inductor current I(Ls) flows in the negative direction to node K and circulates in a loop including Ls, S6 and S2.

[0208] At ta, S2 and S6 are open, and auxiliary switch S5 is closed. Turning off S6 forces I(Ls), which is still flowing towards node K, to flow through D5. Therefore, S5 is turned on with zero voltage. Now I(Ls) still flows towards node K and circulates in a loop including S5, Cbus, D2, and Ls. The Ls voltage V(K, M) is at Vdc + Vd, because V(K) is at Vdc and V(M) is at -Vd. This positive inductor voltage will cause I(Ls) to increase at a rate of (Vdc + Vd) / Ls.

[0209] After ta, I(Ls), which is the same as I(S5), begins to slope linearly toward I(out) with a constant slope of (Vdc+Vd) / Ls. When I(Ls) = I(out), D2 is soft-turned off, and the small reverse recovery current of D2 makes I(Ls) slightly larger than I(out). From here, the negative feedback mechanism discussed earlier causes I(Ls) to track I(out), opening the near-zero current turn-on window of the main switch S1.

[0210] At tb, switch S1 is turned on with near-zero current. Since both S5 and S1 are now on, the voltage across Ls is zero. I(Ls) should theoretically remain constant and begin circulating in the loop including Ls, S1, and S5. However, resistive losses in the loop cause I(Ls) to decrease slightly over time, as... Figure 12 As shown, Figure 12 The waveform associated with circuit A in control method B is shown, where the auxiliary switch and the main switch are turned off simultaneously, and Ls = 6uH.

[0211] At tc, I(output) changes direction from positive to negative. Between tb and tc, and between S5 and S1, I(output) is shared, which helps distribute conduction losses. Compared to the first control method, I(output) is shared between the auxiliary and main MOSFETs for a longer period, resulting in a more even distribution of conduction losses.

[0212] After tc, I(out) changes direction and begins to flow toward the source of S1. Since I(Ls) also flows toward the source of S1, I(out) and I(Ls) now combine in the same direction, resulting in I(S1) being greater than I(out) by I(Ls).

[0213] At time td, S1 and S5 are simultaneously turned off. Because I(Ls) still must flow in the positive direction, it transfers from the body diode of S5 to S6, opening a window for zero-voltage turn-on of S6. S6 is briefly turned on at zero voltage.

[0214] After td, I(Ls) circulates in the loop including D1, Cbus, S6 / D6, and Ls, and a negative voltage (-Vdc-Vd) is applied to Ls, causing its current to slope down at a constant rate of -(Vdc+Vd) / Ls until it catches up with the negative I(out) when it soft-turns off due to the ramp-up of the inductor current. The reverse recovery current of D1 flows in the same direction as I(S1), causing I(Ls) to decrease slightly further, and the negative feedback mechanism kicks in, opening the window for zero-current turn-on of S2.

[0215] Third Embodiment

[0216] This third embodiment is similar to the second embodiment, except that circuit B is used instead of circuit A. That is, the difference from the third embodiment is the single capacitor (as inductor 32) placed in series with a single inductor. This difference will now be explained in more detail.

[0217] Reference Figure 13 The circuit B shown is used to describe this third embodiment. The circuit B includes the following features:

[0218] Circuit 201, which corresponds to Circuit 1

[0219] • Bridge switch circuit 202, which corresponds to bridge switch circuit 2

[0220] • Switching sub-circuit 204 (which may be referred to as "auxiliary circuit") corresponds to switching sub-circuit 4

[0221] • The left half-bridge is connected to 206, which corresponds to the half-bridge sub-circuit 6.

[0222] • The right half-bridge is connected to 208, which corresponds to the half-bridge sub-circuit 8.

[0223] MOSFETs 212 (“(Main Switch) S1”), 214 (“(Main Switch) S2”), 216 (“S3”), 218 (“S4”), 234 (“(Auxiliary Switch) S5”), and 236 (“(Auxiliary Switch) S6”) correspond to switching components 12, 14, 16, 18, 34, and 36, respectively.

[0224] Diodes 212a (“(Diode)D1”) and 214a (“(Diode)D2”) correspond to diodes 12a and 14a, respectively.

[0225] • Resonant circuit 220, which corresponds to load 20

[0226] • Nodes 226 (“(Node)M”), 228, 230, and 238 (“(Node)K”) correspond to nodes 26, 28, 30, and 38, respectively.

[0227] • Inductor (“Ls”) 232, which corresponds to inductor component 32

[0228] • The “I (out)” current is 240, which corresponds to an output current of 40.

[0229] • “I(Ls)” current 242, which corresponds to output current 42.

[0230] • “I(D2)” current 248, which corresponds to current 48

[0231] • “I(D1)” current 252, which corresponds to current 52.

[0232] Compared to circuit A, circuit B adds a capacitor connected in series with the shared inductor Ls, such as... Figure 13 As shown.

[0233] One or more of the main advantages are:

[0234] 1) Used to assist MOSFET zero-voltage turn-on

[0235] 2) Zero-current turn-on of the main MOSFET

[0236] 3) Compared to the second embodiment (circuit A with control method B), circuit B can reduce the high shutdown current of the main switch.

[0237] 4) Compared with circuit A, it can reduce the turn-off current of the auxiliary switch.

[0238] 5) Allows the sharing of output current between the main and auxiliary MOSFETs.

[0239] Circuit B uses control method B; S1 and S5 are simultaneously turned off, and S2 and S6 are simultaneously turned off. The inductor Ls and capacitor Cs form a series resonant network, causing the inductor current I(Ls) to follow a sinusoidal trajectory. The resonant frequency fs of Ls and Cs is chosen to be lower than the system's resonant frequency.

[0240]

[0241] Figure 14 The waveforms of the example design are shown, where fs = 63 kHz, Ls = 6 μH, Cs = 1 μF, and the system resonant frequency is 85 kHz. All parameters of this design were intentionally chosen to be the same as those of circuit A with control method B used for direct comparison. Cs is the only addition. The purpose is to illustrate its effectiveness in reducing the turn-off current of the auxiliary and main MOSFETs.

[0242] like Figure 14As shown, because fs is chosen to be below the system resonant frequency, when S5 is turned on, I(Ls) of I(S5) decreases more slowly than the output current I(output). The sinusoidal current shape of I(S5) allows S5 to turn off with a much smaller positive current, thus achieving zero-voltage turn-on of the auxiliary switch while minimizing the turn-off current of both the auxiliary and main MOSFETs to reduce turn-off losses. Figure 14 In this embodiment, the turn-off current of S5 is approximately 2A, which is significantly lower than that of the second embodiment (circuit A with control method B). Figure 12 The 7A S5 turn-off current.

[0243] The circuit operation is similar to that of the second embodiment (circuit A + control method B), and is briefly described below.

[0244] At ta, S2 and S6 are turned off, and S5 is turned on with zero voltage due to the small negative current in Ls.

[0245] After ta and before I(Ls) = I(out), I(Ls) (which is also I(S5)) begins to rise linearly at a rate of (Vdc + Vd) / Ls due to S5 being turned on and D2 conducting. When I(Ls) = I(out), diode D1 is soft-turned off, and its small reverse recovery current causes I(Ls) to slightly exceed I(out). Due to the negative feedback mechanism discussed earlier, I(Ls) begins to track I(out).

[0246] At tb, the main switch S1 is turned on with zero current. Since both S1 and S5 are now on, the series resonant circuit formed by Cs and Ls is effectively short-circuited, allowing I(Ls) to begin following a sinusoidal trajectory. In this example, the inductor current closely tracks the shape of I(out), and almost no current flows through the main MOSFET S1 for most of its on-cycle. I(S1) begins to increase in the negative direction as I(out) decreases faster than I(S5).

[0247] At tc, I(out) changes direction. I(out) and I(Ls) now add up and both flow to the source of S1. However, since I(S5) has already decreased significantly at tc, I(S1) is not significantly greater than I(out).

[0248] At time td, S1 and S5 are simultaneously turned off, and positive I(Ls) flows through D6, opening the zero-voltage turn-on window of S6. S6 is then turned on with zero voltage shortly thereafter.

[0249] Alternative situations

[0250] So far, a detailed description has been provided of the switching sub-circuit 4 that achieves soft switching of switching components 12, 14, 16, and 18 when the absolute value of the current 40 through load 20 increases. However, soft switching of the switching components can also be achieved in alternative cases where the absolute value of the current 40 through load 20 does not necessarily increase. Such soft switching of switching components 12, 14, 16, and 18 in these alternative cases will now be discussed.

[0251] Soft switching when the absolute value of the current decreases

[0252] Soft-switching of switching components 12 and 14 can be achieved when the absolute value of the current 40 through load 20 decreases (i.e., when the current 40 through the load decreases while being positive, or increases while being negative). This is achieved by operating the switching sub-circuit 4 to result in a (near) zero-voltage switching state on switching component 12, which will refer to Figure 2A-2C , Figure 23 and Figure 25A -D will be discussed later. Switching subcircuit 4 can also be operated to induce a (near) zero-voltage switching state across switching element 14, which will be discussed later. Figures 5A-5C Let's have a discussion.

[0253] Figure 2A-2C The sequence in which switching components 34 and 12 are turned on is shown, i.e., if all switching components are turned off (see...). Figure 2A If the switch component 34 is turned on first (see...), then the switch component 34 will be turned on first. Figure 2B This allows the switch component 12 to be "softly" switched on (see...). Figure 2C ). Figure 25A A flowchart is shown, from stage A to stage C, illustrating how the switching sub-circuit 4 interacts with the bridging switching circuit 2 when the switching component 34 is turned on to generate a zero-voltage state for the "soft" turn-on of the switching component 12. Figure 25B-25D Each phase (AC) is expanded separately.

[0254] exist Figure 2A In this configuration, all switching components 12, 14, 16, 18, 34, and 36 are open. The current output 42 is zero, i.e., i 开关_补偿 =0, and in this case, the current output 40 is positive (flowing from left to right) and decreases sinusoidally. This means that the current output 40 is greater than the current output 42, i.e., i 负载 >i 开关_补偿According to Kirchhoff's Current Law, the total current entering node 26 is equal to the total current leaving node 26. For node 26 to operate according to Kirchhoff's Current Law, the positive and decreasing current output 40 forward-biased diode 14a, allowing current to flow from ground node 30 to node 26 through diode 14a, where the current through diode 14a matches the absolute value and direction of current output 40. That is, the current through diode 14a is also positive and decreasing, such that the current through diode 14a corresponds to current output 40. The forward bias conduction of diode 14a implies the existence of V0. D2 The voltage drop makes the voltage at node 26 -V. D2 Preferably, the voltage drop across diode 14a should be negligible, such that the voltage at node 26 is essentially zero volts (i.e., -V). D2 ≈0).

[0255] like Figure 25A and 25B Phase A shown describes what happens once the switching component 34 is in operation. Figure 2B The diagram illustrates what happens when the switch is turned on. Turning on the switch 34 allows current to flow from node 28 through the switch 34 to node 38, and from node 38 through the inductor 32 to node 26. This is because the voltage across the switch 34 drops to zero and the voltage at node 28 is at the reference voltage V. 基准 Therefore, the voltage at node 38 is also at the reference voltage V. 基准 If it has already been established, the voltage at node 26 is -V. D1 This means that a Vt is applied across the inductor component 32. 基准 +V D2 The (non-zero) voltage difference applied across the inductor 32 causes a (linear) increase in the current through the switch 34, such as... Figure 23 As shown. Reference Figure 4B As the current 42 through the inductor 32 increases (linearly), the current 48 through the diode 14a decreases (linearly) accordingly. And as the current output 42 gets closer to the current output 40, the current 48 through the diode 14a approaches zero and is soft-turned off due to the inductor 32. This helps to minimize the diode reverse recovery energy by slowing down the diode turn-off di / dt.

[0256] like Figure 25A and 25CPhase B, as shown, describes what happens after Phase A once current output 42 “catches up” with current output 40. When current output 42 finally “catches up” with current output 40, the current 48 through diode 14a reaches zero, and diode 14a turns off, stopping the forward bias conduction of diode 14a. The turn-off of diode 14a results in a reverse recovery current 50, which instantaneously conducts current from node 26 to ground node 30. According to Kirchhoff's current law, the reverse recovery current through diode 14a is compensated by the increase in current through inductor 32, making the total current entering node 26 equal to the total current leaving node 26. The increase in current through inductor 32 means that the current output 42 of switching subcircuit 4 “exceeds” the current output 40, making current output 42 more positive than current output 40.

[0257] like Figure 25A and 25D Phase C, as shown, describes what happens after phase B, once the current output 42 “exceeds” the reduced current output 40, and the (near) zero-voltage turn-on window opens first. At this point, diode 12a is forward biased, causing the current difference between currents 42 and 40 to be conducted through diode 12a to node 28, causing node 26 to operate according to Kirchhoff's current law. The current conduction from node 26 to node 28 through diode 12a implies the presence of V D1 The voltage drop means that the voltage at node 26 is now from -V D2 Increase to V 基准 +V D1 The voltage at node 26 increases to V. 基准 +V D1 This means that the voltage drop across the inductor component changes from V 基准 +V D2 Reduce to -V D1 This causes the current 42 through the inductor 32 to "flatten" and "level out," thereby generating a substantially constant current 42 through the inductor 32, such as... Figure 23 As shown. At this time, the voltage drop across the switching component 12 is -V. D1 It has a (near) zero voltage drop because the voltage at node 28 is V. 基准 And the voltage at node 26 is V 基准 +V D1 This provides a near-zero voltage condition 43 across the switching element 12, enabling the switching element 12 to be soft-switched. Figure 23 As shown, as the current output 40 continues to decrease further, the conduction current through diode 12a increases towards current 42. Figure 23In the example shown, as long as the current 42 through the inductor 32 remains more positive than the current 40 through the load 20, the (near) zero-voltage switching window 43 for the switching element 12 (which allows for soft switching of the switching element 12) remains open.

[0258] Those skilled in the art will recognize that when the current 40 through load 20 is negative (i.e., the current 40 through load 20 is in Figure 5A The same principle applies when the switching component 14 is expected to "softly" turn on, in a situation where the current (flowing from right to left) increases sinusoidally, causing the absolute value of the current 40 to decrease over time. (See reference) Figure 5A When switching components 12, 14, 16, 18, 34, and 36 are open, and the current output 42 is zero, i.e. 开关_补偿 When = 0, switch component 36 can be turned on first (see Figure 5B This generates a (near) zero voltage condition applied across the switching element 14, allowing the switching element 14 to be "softly" turned on (see...). Figure 5C ).

[0259] Soft switching when current is constant

[0260] When the absolute value of the (positive or negative) current 40 through load 20 is substantially constant, soft switching of switching components 12 and 14 can be achieved. This is achieved by operating switching sub-circuit 4 to simultaneously induce a (near) zero voltage switching state across switching component 12 and a (near) zero current switching state through switching component 12, which will refer to Figure 2A-2C , Figure 24 and Figure 25A -D will be discussed further. Switching subcircuit 4 can also be operated to (simultaneously) induce a (near) zero voltage switching state across switching element 14 and a (near) zero current switching state through switching element 14, which will be discussed later. Figures 5A-5C Let's have a discussion.

[0261] Figure 2A-2C The sequence in which switching components 34 and 12 are turned on is shown, i.e., if all switching components are turned off (see...). Figure 2A If the switch component 34 is turned on first (see...), then the switch component 34 will be turned on first. Figure 2B This allows the switch component 12 to be "softly" switched on (see...). Figure 2C ). Figure 25A A flowchart is shown, from stage A to stage C, illustrating how the switching subcircuit 4 interacts with the bridging switching circuit 2 when the switching component 34 is turned on to generate a zero-voltage state and a zero-current state for the "soft" turn-on of the switching component 12. Figure 25B-25D Each phase (AC) is expanded separately.

[0262] exist Figure 2AIn this configuration, all switching components 12, 14, 16, 18, 34, and 36 are open. The current output 42 is zero, i.e., i 开关_补偿 =0, and in this case, the current output 40 is positive (flowing from left to right) and remains essentially constant. This means that the current output 40 is greater than the current output 42, i.e., i 负载 >i 开关_补偿 According to Kirchhoff's Current Law, the total current entering node 26 is equal to the total current leaving node 26. To make node 26 operate according to Kirchhoff's Current Law, a positive and constant current output 40 forward-biases diode 14a, allowing current to flow from ground node 30 through diode 14a to node 26, where the current through diode 14a matches the absolute value and direction of current output 40. That is, the current through diode 14a is also positive and constant, such that the current through diode 14a corresponds to current output 40. The forward bias conduction of diode 14a implies the existence of V0. D2 The voltage drop makes the voltage at node 26 -V. D2 Preferably, the voltage drop across diode 14a should be negligible, such that the voltage at node 26 is essentially zero volts (i.e., -V). D2 ≈0).

[0263] like Figure 25A and 25B Phase A shown describes what happens once the switching component 34 is in operation. Figure 2B The diagram illustrates what happens when the switch is turned on. Turning on the switch 34 allows current to flow from node 28 through the switch 34 to node 38, and from node 38 through the inductor 32 to node 26. This is because the voltage across the switch 34 drops to zero and the voltage at node 28 is at the reference voltage V. 基准 Therefore, the voltage at node 38 is also at the reference voltage V. 基准 If it has already been established, the voltage at node 26 is -V. D1 This means that a Vt is applied across the inductor component 32. 基准 +V D2 The (non-zero) voltage difference applied across the inductor 32 causes a (linear) increase in the current through the switch 34, such as... Figure 24 As shown. Reference Figure 4B As the current 42 through the inductor 32 increases (linearly), the current 48 through the diode 14a decreases (linearly) accordingly. And as the current output 42 gets closer to the current output 40, the current 48 through the diode 14a approaches zero and is soft-turned off due to the inductor 32. This helps to minimize the diode reverse recovery energy by slowing down the diode turn-off di / dt.

[0264] like Figure 25A and 25C Phase B, as shown, describes what happens after Phase A once current output 42 “catches up” with current output 40. When current output 42 finally “catches up” with current output 40, the current 48 through diode 14a reaches zero, and diode 14a turns off, stopping the forward bias conduction of diode 14a. The turn-off of diode 14a results in a reverse recovery current 50, which instantaneously conducts current from node 26 to ground node 30. According to Kirchhoff's current law, the reverse recovery current through diode 14a is compensated by the increase in current through inductor 32, making the total current entering node 26 equal to the total current leaving node 26. The increase in current through inductor 32 means that the current output 42 of switching subcircuit 4 “exceeds” the current output 40, making current output 42 more positive than current output 40.

[0265] like Figure 25A and 25D Phase C, as shown, describes what happens after phase B, once the current output 42 “exceeds” the reduced current output 40, and the (near) zero-voltage turn-on window opens first. At this point, diode 12a is forward biased, causing the current difference between currents 42 and 40 to be conducted through diode 12a to node 28, causing node 26 to operate according to Kirchhoff's current law. The current conduction from node 26 to node 28 through diode 12a implies the presence of V D1 The voltage drop means that the voltage at node 26 is now from -V D2 Increase to V 基准 +V D1 The voltage at node 26 increases to V. 基准 +V D1 This means that the voltage drop across the inductor component changes from V 基准 +VD2 decreases to -V D1 This causes the current 42 through the inductor 32 to "flatten" and "level out," thereby generating a substantially constant current 42 through the inductor 32, such as... Figure 24 As shown. At this time, the voltage drop across the switching component 12 is -V. D1 It has a (near) zero voltage drop because the voltage at node 28 is V. 基准 And the voltage at node 26 is V 基准 +V D1 This provides a near-zero voltage condition 43 across the switching element 12, enabling the switching element 12 to soft-switch. Figure 24In the example shown, the (near) zero-voltage switching window 43 for the switching component 12 (which allows for soft switching of the switching component 12) remains open as long as the current 42 through the inductor 32 remains more positive than the current 40 through the load 20. Furthermore, since the current 40 through the load 20 is substantially the same as the current 42 through the inductor 32 (in both absolute value and direction), a (near) zero current state is also generated through the switching component 12, because there is a (near) zero current through the switching component 12 at that point. The window for the (near) zero current state remains open as long as the current 40 through the load 20 is substantially the same as the current 42 through the inductor 32 (in both absolute value and direction).

[0266] In stages B and C, current outputs 40 and 42 are substantially matched. Substantial matching of the current outputs means that, according to Kirchhoff's current law, there is (near) zero current flowing through switching element 12. Therefore, the substantial matching of the current outputs produces a (near) zero current switching window 45, which can potentially allow switching element 12 to soft-switch forward from stage B, as... Figure 24 As shown.

[0267] Those skilled in the art will recognize that when the current 40 through load 20 is negative (i.e., the current 40 through load 20 is in Figure 5A The same principle applies when the switching element 14 is expected to be "softly" switched on, provided that the current (flowing from right to left) remains constant so that the absolute value of the current 40 is kept substantially the same. (See reference.) Figure 5A When switching components 12, 14, 16, 18, 34, and 36 are open, and the current output 42 is zero, i.e. 开关_补偿 When = 0, switch component 36 can be turned on first (see Figure 5B This simultaneously generates a near-zero voltage state applied across the switching element 14 and a near-zero current state through the switching element 14, allowing the switching element 14 to be "softly" turned on (see...). Figure 5C ).

[0268] Various examples of soft switches are described in the manual. While these examples involve soft turning on, at least some of the examples described above can also be used to implement soft turning off.

Claims

1. A soft-switching subcircuit, forming part of or used with a circuit including a bridging switch circuit, the soft-switching subcircuit being configured and operable to provide a varying current output that tracks the output current from the bridging switch circuit to generate a substantially zero current through at least one bridging switch element of the bridging switch circuit to enable soft switching. in, The variable current output includes: The acceleration current output when the first absolute value of the varying current output is less than the second absolute value of the output current from the bridge switching circuit, and The decelerated current output when the first absolute value of the variable current output is greater than the second absolute value of the output current from the bridge switching circuit.

2. The soft-switching sub-circuit according to claim 1, wherein, The variable current output includes alternation between the acceleration current output and the deceleration current output.

3. The soft-switching sub-circuit according to claim 1, wherein, The soft-switching sub-circuit includes an inductor and at least one switch.

4. The soft-switching sub-circuit according to claim 3, wherein, The soft-switching sub-circuit includes a first switching component and a second switching component. The first switching component is used to supply power to the first terminal of the inductor with a voltage that is a reference voltage or a voltage derived from the reference voltage, and the second switching component is used to supply power to the first terminal of the inductor with a voltage that is zero voltage or ground or a voltage derived from zero voltage or ground.

5. The soft-switching sub-circuit according to claim 4, wherein, The first switching component is connected between the node that can be powered by the reference voltage and the first end of the inductor, and the second switching component is connected between the first end of the inductor and zero voltage or ground.

6. The soft-switching sub-circuit according to claim 5, wherein, The first switching component of the soft-switching sub-circuit is used for soft switching of the first bridging switching component of the bridging switching circuit, and the second switching component of the soft-switching sub-circuit is used for soft switching of the second bridging switching component of the bridging switching circuit. In the bridge switch circuit: The first bridging switch component is connected between a first node that can be powered by the reference voltage and a second node between the first bridging switch component and the second bridging switch component. The second bridging switch component is connected between the second node and zero voltage or ground.

7. The soft-switching sub-circuit according to claim 3, wherein, The inductor has a first end connected to the at least one switching component of the soft-switching sub-circuit and a second end for connecting to a node between the two bridging switching components of the bridging switching circuit.

8. The soft-switching sub-circuit according to claim 7, wherein, The variable current output is provided by the operation of the inductor in the following manner: When the output current from the bridge switch circuit exceeds the changing current output of the soft-switching sub-circuit, the voltage difference applied across the inductor is increased, causing the changing current output of the soft-switching sub-circuit to exceed the output current of the bridge switch circuit. When the changing current output of the soft-switching sub-circuit exceeds the output current from the bridge switching circuit, the voltage difference applied across the inductor decreases, causing the output current from the bridge switching circuit to exceed the changing current output of the soft-switching sub-circuit.

9. The soft-switching sub-circuit according to claim 8, wherein: The soft-switching subcircuit includes a switching component connected to a first terminal of the inductor, the switching component being configured to provide a reference voltage or a voltage derived from a reference voltage to the first terminal of the inductor. The two bridging switch components in the bridging switch circuit are: A first bridging switch component includes a first diode that connects a node between two bridging switch components of the bridging switch circuit to another node capable of being powered by the reference voltage, and The second bridging switch component includes a second diode that connects to a node between the two bridging switch components of the bridging switch circuit with zero voltage or ground.

10. The soft-switching sub-circuit according to claim 9, wherein: The forward bias conduction of the second diode causes the voltage at the second terminal of the inductor to decrease toward the first voltage, which is either zero voltage or ground, or derived from that zero voltage or ground. This increases the voltage difference applied across the inductor, causing the changing current output of the soft-switching sub-circuit to exceed the output current from the bridge switching circuit. The forward bias conduction of the first diode causes the voltage at the second terminal of the inductor to increase toward the second voltage, which is a reference voltage or derived from the reference voltage, thereby reducing the voltage difference applied across the inductor and causing the output current from the bridge switching circuit to exceed the changing current output from the soft switching sub-circuit.

11. The soft-switching sub-circuit according to claim 9, wherein, The inductor component is adapted to reduce the rate of change of current through the first and second diodes of the first and second bridge switching components of the bridge switching circuit, thereby reducing the peak reverse recovery current of the first and second diodes.

12. The soft-switching sub-circuit according to any one of claims 8-11, wherein: The at least one switching component of the soft-switching subcircuit includes a switching component connected to the first terminal of the inductor component, the switching component being configured to provide a voltage to the first terminal of the inductor component, the voltage being zero voltage or ground, or derived from zero voltage or ground. The two bridging switch components in the bridging switch circuit are: The first bridging switch component includes a first diode that connects a node between the two bridging switch components of the bridging switch circuit to another node capable of being powered by a reference voltage, and The second bridging switch component includes a second diode that connects to a node between the two bridging switch components of the bridging switch circuit with zero voltage or ground.

13. The soft-switching sub-circuit according to claim 12, wherein: The forward bias conduction of the second diode causes the voltage at the second terminal of the inductor to decrease toward the first voltage, which is either zero voltage, ground, or derived from zero voltage or ground. This reduces the voltage difference applied across the inductor, causing the output current from the bridge switching circuit to exceed the changing current output of the soft-switching sub-circuit. The forward bias conduction of the first diode causes the voltage at the second terminal of the inductor to increase toward the second voltage, which is a reference voltage or derived from the reference voltage, thereby increasing the voltage difference applied across the inductor and causing the variable current output of the soft-switching sub-circuit to exceed the output current from the bridge switching circuit.

14. The soft-switching sub-circuit according to claim 7, wherein, The variable current output includes alternation between the acceleration current output and the deceleration current output.

15. The soft-switching sub-circuit according to claim 14, wherein, Alternation includes: When the first absolute value of the changing current output exceeds the second absolute value of the output current from the bridge switching circuit, the output changes from the accelerating current output to the decelerating current output. When the second absolute value of the output current from the bridge switch circuit exceeds the first absolute value of the changing current output, the output changes from decelerating current output to accelerating current output.

16. The soft-switching sub-circuit according to claim 7, wherein: The soft-switching subcircuit includes a switching component connected to a first terminal of the inductor, the switching component being configured to provide a reference voltage or a voltage derived from a reference voltage to the first terminal of the inductor. The two bridging switch components in the bridging switch circuit are: A first bridging switch component includes a first diode that connects a node between two bridging switch components of the bridging switch circuit to another node capable of being powered by the reference voltage, and The second bridging switch component includes a second diode that connects to a node between the two bridging switch components of the bridging switch circuit with zero voltage or ground.

17. The soft-switching sub-circuit according to claim 16, wherein, The first diode of the first bridging switch component and the second diode of the second bridging switch component are operational, such that: The forward bias conduction of the first diode generates the decelerating current output by increasing the voltage at the second terminal of the inductor, thereby reducing the voltage difference applied across the inductor. The forward bias conduction of the second diode generates the accelerating current output by reducing the voltage at the second terminal of the inductor, thereby increasing the voltage difference applied across the inductor.

18. The soft-switching sub-circuit according to claim 17, in, The inductive component is an inductor or includes an inductor, and The capacitor is connected in series with the inductor.

19. The soft-switching sub-circuit according to claim 7, wherein: The soft-switching subcircuit includes a switching component connected to a first terminal of the inductor, the switching component being configured to provide a voltage to the first terminal of the inductor that is zero voltage, ground, or derived from zero voltage or ground. The two switching components of the bridged switch circuit are: The first bridging switch component includes a first diode that connects a node between the two bridging switch components of the bridging switch circuit to another node capable of being powered by a reference voltage, and The second bridging switch component includes a second diode that connects to a node between the two bridging switch components of the bridging switch circuit with zero voltage or ground.

20. The soft-switching sub-circuit according to claim 19, wherein, The first diode of the first bridging switch component and the second diode of the second bridging switch component are operational, such that: The forward bias conduction of the first diode generates the accelerating current output by increasing the voltage at the second terminal of the inductor, thereby increasing the voltage difference applied across the inductor. The forward bias conduction of the second diode generates the deceleration current output by reducing the voltage at the second terminal of the inductor, thereby reducing the voltage difference applied across the inductor.

21. A soft-switching subcircuit, forming part of or for use with a circuit including a bridging switch circuit, the soft-switching subcircuit being configured and operable to provide a variable current output that tracks the output current from the bridging switch circuit to generate a substantially zero current through at least one bridging switch element of the bridging switch circuit to enable soft switching. in, The soft-switching sub-circuit includes an inductor and at least one switch. The inductor has a first end and a second end. The first end is connected to at least one switching component of the soft-switching sub-circuit, and the second end is used to connect to a node between two bridging switching components of the bridging switching circuit. The variable current output is provided by the operation of the inductor component in the following manner: When the output current from the bridge switch circuit exceeds the changing current output of the soft-switching sub-circuit, the voltage difference applied across the inductor increases, causing the changing current output of the soft-switching sub-circuit to exceed the output current from the bridge switch circuit. When the changing current output of the soft-switching sub-circuit exceeds the output current from the bridge switch circuit, the voltage difference applied across the inductor component decreases, causing the output current from the bridge switch circuit to exceed the changing current output of the soft-switching sub-circuit.

22. A soft-switching subcircuit, forming part of or for use with a circuit including a bridging switch circuit, the soft-switching subcircuit being configured and operable to provide a variable current output that tracks the output current from the bridging switch circuit to generate a substantially zero current through at least one bridging switch element of the bridging switch circuit to enable soft switching. in, The soft-switching sub-circuit includes an inductor and at least one switch. The inductor has a first terminal and a second terminal. The first terminal is connected to at least one switching component of the soft-switching sub-circuit, and the second terminal is used to connect to a node between two bridging switching components of the bridging switching circuit. The variable current output includes: The acceleration current output when the first absolute value of the varying current output is less than the second absolute value of the output current from the bridge switching circuit, and The decelerated current output when the first absolute value of the variable current output is greater than the second absolute value of the output current from the bridge switching circuit. The acceleration current output is generated by applying an increased voltage across the inductor, and the deceleration current output is generated by applying a decreased voltage across the inductor.

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