Radar communication integrated device and method based on resampling phase compensation

Abstract

The application relates to a radar-communication integrated device and method based on resampling phase compensation, which comprises a continuous wave laser, first and second polarization controllers, first and second Mach-Zehnder modulators, an arbitrary waveform generator, an optical band rejection filter, an erbium-doped fiber amplifier, an optical band pass filter, an optical photoelectric detector, an electrical band pass filter, an electrical amplifier, a power divider, first and second mixers, an antenna, first and second low noise amplifiers, a microwave signal source and a real-time oscilloscope. At the transmitting end, two Mach-Zehnder modulators and an optical band rejection filter are used to generate a four-fold frequency multi-level phase shift keying linear frequency modulation signal; at the receiving end, when the time delay of a return signal relative to a reference signal is close to or exceeds a communication symbol length, a resampling phase compensation method is adopted to break the limitation of radar detection performance by the communication symbol width. The application has compact structure, simple and effective, and good feasibility and application prospect.

Description

Technical Field

[0001] This invention belongs to the field of radar communication integration technology, specifically relating to a radar communication integration device and method based on resampling phase compensation. Background Technology

[0002] Both radar sensing systems and wireless communication systems belong to radio systems. Radar sensing systems illuminate targets by emitting electromagnetic waves and receive echoes to obtain information such as the target's distance, speed, and image. Wireless communication systems transmit electromagnetic waves at the transmitter and receive them at the receiver to obtain the communication data carried by the electromagnetic waves. These two systems respectively undertake the missions of target detection and information exchange, and have become widely equipped radio systems in both military and civilian fields.

[0003] To achieve high-speed communication and high-precision target detection, the operating bandwidth and frequency bands of communication and radar systems are continuously increasing, leading to a growing shortage of spectrum resources. In both military and civilian fields, unmanned aerial vehicle (UAV) warfare and urban air traffic have become cutting-edge hot topics. These systems not only require all-weather, complex environmental perception but also need to exchange information between different devices. The increasing number of electromagnetic devices results in bulky systems and severe electromagnetic interference. To reduce system size, mitigate electromagnetic interference, and improve spectrum utilization efficiency, integrated radar-communication systems have been extensively researched. High-speed communication and high-precision target detection necessitate the generation of high-frequency, broadband, and tunable integrated radar-communication waveforms in integrated radar-communication systems. However, limited by electronic bottlenecks, integrated radar-communication signals generated using traditional electronic technologies suffer from low electronic device speeds, small operating bandwidths, and poor reconfigurability, failing to meet the application requirements of next-generation integrated radar-communication systems.

[0004] Microwave photonics technology has been widely studied due to its advantages such as large bandwidth, low loss, and easy tuning, and is used to overcome some bottleneck problems in traditional electronic technology. In recent years, some radar-communication integrated schemes based on microwave photonics technology have also been reported for radar-communication integrated systems. The radar-communication integrated signals are usually divided into two categories: one is signal multiplexing, that is, radar signals and communication signals occupy different time, space and other dimensions, and the two signals are independent of each other and do not interfere with each other; the other is signal sharing, that is, radar and communication use the same waveform. The orthogonal frequency division multiplexing signal and the linear frequency modulation signal are allocated to different time segments, and after microwave photonic frequency conversion, the radar-communication integrated signal in the W band is generated. The working bandwidth of the integrated signal is 10 GHz. The system can achieve centimeter-level distance resolution and high-speed data communication. Its disadvantage is that it is necessary to generate two signals separately in the electric domain and fuse them through time division multiplexing. The signal generation difficulty and complexity are high (Opt. Lett., 46(24): 6103-6106, 2021). To reduce the difficulty of integrated signal generation and improve spectral efficiency, orthogonal frequency division multiplexing (Opt. Exp., 29(14): 22442-22454, 2021) and orthogonal phase shift keying (QPSK) signals based on optoelectronic oscillators (J. Lightw. Technol., 40(13): 4101-4109, 2022) were used to realize data communication and range measurement, respectively. On the one hand, both of the above schemes realize range measurement through cross-correlation algorithms, and the radar receiver needs to obtain the complete echo signal waveform, which puts higher requirements on the operating bandwidth of the radar receiver; on the other hand, the relevant experiments did not obtain target imaging information. To acquire target imaging information and reduce the bandwidth requirements of radar receivers, a radar-communication integrated system based on quadrature phase shift keying linear frequency modulation (QPSK-LFM) signals has been proposed. The ISAR imaging resolution is 14.99 cm × 3.25 cm, and the communication rate is 105.26 Mbps or 210.52 Mbps (Appl. Optics, 60(16): 4752-4760, 2022). However, in the aforementioned radar-communication integrated scheme based on QPSK-LFM signals, the radar detection performance is limited by the communication rate due to the random phase jumps of the descrambling signal. Specifically, range measurement and ISAR imaging cannot be achieved when the time delay of the echo signal relative to the reference signal is close to or exceeds the length of one communication symbol. Therefore, further research on radar-communication integrated systems based on multi-level phase shift keying linear frequency modulation (MPSK-LFM) signals and the proposal of solutions to overcome the limitation of radar detection performance on the length of communication symbols is an important research topic. The advancement of this topic is of great practical significance for the realization of a new generation of radar-communication integrated systems. Summary of the Invention

[0005] To address the technical problems existing in the background art, this invention proposes an integrated radar-communication device and method based on resampling phase compensation. By reasonably setting the precoded vector signal type and carrier frequency, it is possible to generate a fourth-harmonic MPSK-LFM signal. After adopting the resampling phase compensation method, the radar receiver can effectively overcome the limitation of communication symbol length in target range measurement and ISAR imaging. This invention, while realizing the integrated generation of high-frequency, broadband radar-communication signals, increases the system's tunability and feasibility, and has significant strategic importance and application value.

[0006] The present invention adopts the following solution to solve its technical problem:

[0007] An integrated radar-communication device based on resampling phase compensation is characterized by comprising a continuous-wave laser, a first polarization controller, a first Mach-Zehnder modulator, a second polarization controller, a second Mach-Zehnder modulator, an arbitrary waveform generator, an optical bandstop filter, an erbium-doped fiber amplifier, an optical bandpass filter, a photodetector, an electrical bandpass filter, an electrical amplifier, a power divider, a first mixer, an antenna, a first low-noise amplifier, a second low-noise amplifier, a second mixer, a microwave signal source, and a real-time oscilloscope; the output port of the continuous-wave laser is connected to one port of the first polarization controller. The other port of the first polarization controller is connected to the input port of the first Mach-Zehnder modulator. The output port of the first Mach-Zehnder modulator is connected to one port of the second polarization controller. The other port of the second polarization controller is connected to the input port of the second Mach-Zehnder modulator. The two output ports of the arbitrary waveform generator are respectively connected to the RF input ports of the first and second Mach-Zehnder modulators. The output port of the second Mach-Zehnder modulator is connected to the input port of the optical bandstop filter. The output port of the optical bandstop filter is connected to the input port of the erbium-doped fiber amplifier. The erbium-doped fiber amplifier's output port is connected to the input port of an optical bandpass filter, which in turn is connected to the input port of a photodetector. The photodetector's output port is connected to the input port of an electrical bandpass filter, which in turn is connected to the input port of an electrical amplifier. The electrical amplifier's output port is connected to the input port of a power divider, and the two output ports of the power divider are respectively connected to the antenna and the local oscillator port of the first mixer. The echo signal reflected from the target is received by the antenna and connected to the input port of a first low-noise amplifier. The output port of the first low-noise amplifier is connected to the RF input port of the first mixer, and the intermediate frequency output port of the first mixer is connected to a real-time oscilloscope. The MPSK-LFM signal received at the communication terminal is connected to the input port of a second low-noise amplifier, whose output port is connected to the RF port of a second mixer. The output port of the microwave signal source is connected to the local oscillator port of the second mixer, and the intermediate frequency output port of the second mixer is connected to the input port of the real-time oscilloscope. The required transmitted data can be obtained by digitally demodulating the data sampled by the real-time oscilloscope. Wherein:

[0008] The carrier frequency of the precoded vector signal loaded onto the first Mach-Zehnder modulator is the same as the center frequency of the linear frequency modulated signal loaded onto the second Mach-Zehnder modulator, and the center frequency of the optical bandstop filter is the same as the frequency of the output optical signal of the continuous wave laser.

[0009] In the device, the phase of the precoded vector signal is half that of the conventional multi-level phase shift keying signal. Changing the modulation format of the precoded vector signal can realize the generation of radar communication integrated signals of different formats.

[0010] A radar-communication integration method based on resampling phase compensation using the above-mentioned device, the method comprising the following steps:

[0011] 1) The output frequency of the continuous wave laser is f C The optical carrier is injected into the first Mach-Zehnder modulator after passing through the first polarization controller. A precoded vector signal output by the arbitrary waveform generator is used to drive the first Mach-Zehnder modulator. The bias voltage of the first Mach-Zehnder modulator is adjusted to make it work at the minimum bias point in order to suppress carrier double-sideband modulation.

[0012] 2) The suppressed carrier double-sideband signal output from the first Mach-Zehnder modulator is injected into the second Mach-Zehnder modulator after passing through the second polarization controller. Another linear frequency modulation signal output from the arbitrary waveform generator is used to drive the second Mach-Zehnder modulator. The bias voltage of the second Mach-Zehnder modulator is adjusted so that it works at the minimum bias point. At this time, the output optical signal of the second Mach-Zehnder modulator consists of three frequency components: one is the optical sideband near the optical carrier, and the other two are two broadband optical sidebands far away from the optical carrier.

[0013] 3) The optical signal output from the second Mach-Zehnder modulator is passed through an optical bandstop filter to eliminate the influence of the optical sidebands near the optical carrier on the generation of the MPSK-LFM signal. Erbium-doped fiber amplifiers are used to amplify the two broadband optical sidebands far from the optical carrier. An optical bandpass filter is used to suppress the spontaneous emission noise introduced by the erbium-doped fiber amplifier.

[0014] 4) After photoelectric detection, the signal passes through an electrical bandpass filter and an electrical amplifier to generate a fourth-harmonic MPSK-LFM signal, which is used as an integrated radar and communication signal.

[0015] 5) The MPSK-LFM signal output by the amplifier is split into two paths by a power divider. One path is used as a reference signal for the radar receiver, and the other path is fed to the transmitting antenna for communication and target detection.

[0016] 6) The echo signal reflected by the target is collected by the receiving antenna and then connected to the radio frequency port of the first mixer after passing through the first low noise amplifier. The intermediate frequency output port of the first mixer is sampled by a real-time oscilloscope. Then, the resampling phase compensation method is used to overcome the problem that the time delay of the echo signal relative to the reference signal is close to or exceeds the communication symbol length, which makes it impossible to realize target distance measurement and ISAR imaging.

[0017] 7) The communication receiver first amplifies the signal received by the antenna using a second low-noise amplifier, then uses a microwave signal source to downconvert the amplified MPSK-LFM signal, then uses a real-time oscilloscope to sample the downconverted signal, and finally obtains the communication data carried by the MPSK-LFM signal through digital domain coherent demodulation.

[0018] The resampling phase compensation method described in step 6) is based on the fact that the resampling rate and the symbol rate are numerically the same. The phase to be compensated is determined according to the time delay of the echo signal relative to the reference signal, the resampling position, and the transmitted communication data, thereby eliminating the influence of the additional phase contained in the descrambling signal on target distance measurement and ISAR imaging.

[0019] This invention proposes an integrated radar-communication device and method based on resampling phase compensation. By rationally setting the precoded vector signal type and carrier frequency, it is possible to generate a fourth-harmonic MPSK-LFM signal. After employing the resampling phase compensation method, the radar receiver can effectively overcome the limitations imposed by communication symbol length on target range measurement and ISAR imaging. This invention achieves integrated high-frequency, broadband radar-communication signal generation while increasing the system's tunability and feasibility, possessing significant strategic importance and application value. Attached Figure Description

[0020] Figure 1 This is a schematic diagram of the device structure of the present invention;

[0021] Figure 2 This is a schematic diagram of the resampling phase compensation method of the present invention;

[0022] Figure 3 The image shows the ISAR imaging results obtained using this invention.

[0023] Figure 4 A graph showing the distance measurement results obtained using this invention;

[0024] Figure 5 The waveform of the baseband signal obtained by demodulating the QPSK-LFM signal using this invention is shown.

[0025] Figure 6 This invention provides a constellation diagram obtained by demodulating a QPSK-LFM signal. Detailed Implementation

[0026] The embodiments of the present invention will now be described in detail with reference to the accompanying drawings. These embodiments are implemented based on the technical solution of the present invention, providing detailed implementation methods and specific operating procedures. However, the scope of protection of the present invention is not limited to the following embodiments.

[0027] See Figure 1The device of the present invention includes: a continuous wave laser 1, a first polarization controller 2, a first Mach-Zehnder modulator 3, a second polarization controller 4, a second Mach-Zehnder modulator 5, an arbitrary waveform generator 6, an optical bandstop filter 7, an erbium-doped fiber amplifier 8, an optical bandpass filter 9, a photodetector 10, an electrical bandpass filter 11, an electrical amplifier 12, a power divider 13, a first mixer 14, an antenna 15, a first low-noise amplifier 16, a real-time oscilloscope 17, a second low-noise amplifier 18, a second mixer 19, and a microwave signal source 20.

[0028] The output port of the continuous wave laser 1 is connected to one port of the first polarization controller 2, the other port of the first polarization controller 2 is connected to the input port of the first Mach-Zehnder modulator 3, the output port of the first Mach-Zehnder modulator 3 is connected to one port of the second polarization controller 4, and the other port of the second polarization controller 4 is connected to the input port of the second Mach-Zehnder modulator 5. The two output ports of the arbitrary waveform generator 6 are respectively connected to the RF input ports of the first Mach-Zehnder modulator 3 and the second Mach-Zehnder modulator 5. The output port of the second Mach-Zehnder modulator 5 is connected to the input port of the optical bandstop filter 7, the output port of the optical bandstop filter 7 is connected to the input port of the erbium-doped fiber amplifier 8, the output port of the erbium-doped fiber amplifier 8 is connected to the input port of the optical bandpass filter 9, and the output port of the optical bandpass filter 9 is connected to the input port of the photodetector 10. The output port of the photodetector 10 is connected to the input port of the electrical bandpass filter 10. The input port of filter 11 is connected to the input port of electric bandpass filter 11, the output port of electric amplifier 12 is connected to the input port of power divider 13, and the two output ports of power divider 13 are respectively connected to the local oscillator port of first mixer 14 and antenna 15; the echo signal reflected by the target is received by antenna 15 and connected to the input port of first low noise amplifier 16, the output port of first low noise amplifier 16 is connected to the RF input port of first mixer 14, and the IF output port of first mixer 14 is connected to real-time oscilloscope 17; the MPSK-LFM signal received by the communication terminal is connected to the input port of second low noise amplifier 18, the output port of second low noise amplifier 18 is connected to the RF port of second mixer 19, the output port of microwave signal source 20 is connected to the local oscillator port of second mixer 19, and the IF output port of second mixer 19 is connected to the input port of real-time oscilloscope 17.

[0029] This invention achieves integrated radar and communication, and the specific steps are as follows:

[0030] Step 1: The output frequency of the continuous wave laser is... f CThe optical carrier is injected into the first Mach-Zehnder modulator after passing through the first polarization controller. A precoded vector signal output by the arbitrary waveform generator is used to drive the first Mach-Zehnder modulator. The bias voltage of the first Mach-Zehnder modulator is adjusted to make it work at the minimum bias point in order to suppress carrier double-sideband modulation.

[0031] Step 2: The suppressed carrier double-sideband signal output from the first Mach-Zehnder modulator is injected into the second Mach-Zehnder modulator after passing through the second polarization controller. Another linear frequency modulation signal output from the arbitrary waveform generator is used to drive the second Mach-Zehnder modulator. The bias voltage of the second Mach-Zehnder modulator is adjusted so that it operates at the minimum bias point. At this time, the output optical signal of the second Mach-Zehnder modulator consists of three frequency components: one is the optical sideband near the optical carrier, and the other two are two broadband optical sidebands far away from the optical carrier.

[0032] Step 3: The optical signal output from the second Mach-Zehnder modulator is passed through an optical bandstop filter to eliminate the influence of the optical sidebands near the optical carrier on the generation of the MPSK-LFM signal. An erbium-doped fiber amplifier is used to amplify the two broadband optical sidebands far from the optical carrier. An optical bandpass filter is used to suppress the spontaneous emission noise introduced by the erbium-doped fiber amplifier.

[0033] Step 4: After photoelectric detection, the signal passes through an electrical bandpass filter and an electrical amplifier to generate a fourth-harmonic MPSK-LFM signal, which is used as the integrated radar and communication signal.

[0034] Step 5: The MPSK-LFM signal output by the amplifier is split into two paths by a power divider. One path is used as a reference signal for the radar receiver, and the other path is fed to the transmitting antenna for communication and target detection.

[0035] Step 6: The echo signal reflected by the target is collected by the receiving antenna and then connected to the radio frequency port of the first mixer after passing through the first low noise amplifier. The intermediate frequency output port of the first mixer is sampled by a real-time oscilloscope. Then, the resampling phase compensation method is used to overcome the problem that the time delay of the echo signal relative to the reference signal is close to or exceeds the communication symbol length, which makes it impossible to realize target distance measurement and ISAR imaging.

[0036] Step 7: The communication receiver first amplifies the signal received by the antenna using a second low-noise amplifier, then downconverts the amplified MPSK-LFM signal using a microwave signal source, then samples the downconverted signal using a real-time oscilloscope, and finally obtains the communication data carried by the MPSK-LFM signal through digital domain coherent demodulation.

[0037] Figure 2This is a schematic diagram of the resampling phase compensation method of the present invention. Compared with the result of de-chewing the linear frequency modulated signal, the MPSK-LFM signal after de-chewing contains an additional phase, which is related to the communication symbol. When the time delay ∆ of the echo signal relative to the reference signal... τ Much smaller than the length of a communication symbol T B At this time, the effect of this additional phase on target range measurement and ISAR imaging can be ignored. When the time delay ∆ of the echo signal relative to the reference signal... τ Approximate communication symbol length T B In such cases, a resampling phase compensation method is required to accurately obtain target distance information and ISAR imaging results. The principle of resampling phase compensation is as follows: Figure 2 As shown in (a). It should be noted that the resampling rate and the symbol rate are numerically the same, that is, one sampling point is taken for each communication symbol. Assume that the first... n The phase corresponding to each communication symbol is φ n In a communication symbol length T B The additional phase of the descrambling signal introduced by the communication symbol. φ n There are two cases: one is that the phase is zero, i.e., ∆ φ n =0, or the phase difference ∆ between two adjacent symbols. φ n = φ n+1 - φ n If the sampling point of each communication symbol is a point with zero phase, then the additional phase contained in the de-scratching signal is zero. In this case, no phase compensation is needed to accurately obtain the target's range information and ISAR imaging results. If the sampling point of each communication symbol is the phase difference ∆ between adjacent symbols... φ n = φ n+1 - φ n At this point, phase compensation needs to be performed on each sampling point, and the compensated phase is -∆ φ n = φ n - φ n+1 This eliminates the impact of the additional phase contained in the descrambling signal on target range measurement and ISAR imaging.

[0038] When the echo signal is delayed relative to the reference signal by ∆ τ Exceeding the length of communication symbols TB At that time, exceeding the length of a communication symbol ( T B <∆ τ <2 T B Taking (e.g.) as an example, the principle of resampling phase compensation is as follows: Figure 2 As shown in (b). Within a communication symbol length T B The additional phase of the descrambling signal introduced by the communication symbol. φ n There are also two possibilities, one is ∆ φ n = φ n+1 - φ n Another one is ∆ φ n = φ n+2 - φ n In both cases, phase compensation is required to obtain target information. If the sampling point of each communication symbol belongs to the first case, then the compensated phase is -∆. φ n = φ n - φ n+1 Otherwise, the compensated phase is -∆ φ n = φ n - φ n+2 Then, the target distance information and ISAR imaging results can also be accurately obtained using the compensated de-slant signal. Example

[0039] The specific implementation process of this embodiment is as follows:

[0040] Step 1: The continuous wave laser outputs an optical carrier with a working wavelength of 1550 nm, which is injected into the first Mach-Zehnder modulator after passing through the first polarization controller. The arbitrary waveform generator outputs a precoded QPSK signal with a carrier frequency of 3 GHz to drive the first Mach-Zehnder modulator. The bias voltage of the first Mach-Zehnder modulator is adjusted to make it work at the minimum bias point to achieve suppressed carrier double-sideband modulation.

[0041] Step 2: The suppressed carrier double-sideband signal output from the first Mach-Zehnder modulator is injected into the second Mach-Zehnder modulator after passing through the second polarization controller. The arbitrary waveform generator outputs another linear frequency modulated signal with a center frequency of 3 GHz, a bandwidth of 0.5 GHz, a pulse width of 3.8 µs, and a pulse period of 4 µs. This linear frequency modulated signal is used to perform suppressed carrier double-sideband modulation on the optical signal output from the first Mach-Zehnder modulator. At this time, the optical signal output from the second Mach-Zehnder modulator consists of three frequency components: one is the optical sideband near the optical carrier, and the other two are two broadband optical sidebands far away from the optical carrier.

[0042] Step 3: The optical signal output from the second Mach-Zehnder modulator is passed through an optical bandstop filter with a center wavelength of 1550 nm and a 3dB bandwidth of 10 GHz. The purpose is to eliminate the influence of the optical sidebands near the optical carrier on the generation of the QPSK-LFM signal. Erbium-doped fiber amplifiers are used to amplify the two broadband optical sidebands far from the optical carrier, and optical bandpass filters are used to suppress the spontaneous emission noise introduced by the erbium-doped fiber amplifier.

[0043] Step 4: After photoelectric detection, the signal passes through an electrical bandpass filter (10.4-14.1 GHz) and an electrical amplifier (5.85-14.5 GHz) to generate a fourth-harmonic QPSK-LFM signal, which is used as the integrated radar communication signal.

[0044] Step 5: The QPSK-LFM signal output by the amplifier is split into two paths by a power divider (2-18 GHz). One path is used as a reference signal for the radar receiver, and the other path is fed to the transmitting antenna for communication and target detection.

[0045] Step 6: The echo signal reflected by the target is collected by the receiving antenna and then connected to the RF port of the first mixer after passing through the first low-noise amplifier (5.85-14.5 GHz). The intermediate frequency output port of the first mixer is sampled by a real-time oscilloscope. Then, the resampling phase compensation method is used to overcome the problem that the time delay of the echo signal relative to the reference signal is close to or exceeds the communication symbol length, which makes it impossible to realize target distance measurement and ISAR imaging.

[0046] Step 7: The communication receiver first amplifies the signal received by the antenna using a second low-noise amplifier, then downconverts the amplified QPSK-LFM signal using a local oscillator signal with a frequency of 11 GHz generated by a microwave signal source, then samples the downconverted signal using a real-time oscilloscope, and finally obtains the communication data carried by the QPSK-LFM signal through digital domain coherent demodulation.

[0047] The communication rate carried by the precoded QPSK signal was set to 200 Mbps, and ISAR imaging tests were performed on the system. In the experiment, the turntable rotation period was 24.56 s, and the distance between the turntable center and the antenna pair along the radar line of sight was 1.55 m. Due to experimental limitations, the time delay of the echo signal relative to the reference signal was adjusted by modifying the cable length between the radar receiving antenna and the first low-noise amplifier. L accomplish. Figure 3 (a) and Figure 3 (b) The ISAR imaging results before and after resampling phase compensation are shown respectively. As can be seen from the figure, the target imaging results can be accurately obtained after using the resampling phase compensation method.

[0048] The communication rate carried by the precoded QPSK signal was set to 400 Mbps, and target distance was measured using the system. In the experiment, the distance between a single stationary target and the antenna pair along the radar line of sight was 1.28 m. The time delay of the echo signal relative to the reference signal was adjusted by varying the cable length between the radar receiving antenna and the first low-noise amplifier. L accomplish. Figure 4 (a) to Figure 4 (d) shows the cable length at the receiving end. L The distance measurement results before and after resampling phase compensation at 0 m, 1 m, 2 m and 3 m are shown in the figure. The dashed line represents the distance measurement result before compensation and the solid line represents the distance measurement result after compensation. As can be seen from the figure, the target distance cannot be determined without compensation. After using the resampling phase compensation method, the target distance information can be accurately obtained.

[0049] The communication rate carried by the precoded QPSK signal was set to 200 Mbps. The baseband signal waveform was obtained by demodulating the QPSK-LFM signal received by the communication receiver to verify the communication performance. Figure 5 (a) and Figure 5 (b) The solid lines show the baseband signals recovered from the I and Q channels, while the dashed lines represent the original baseband signals. Compared to the original baseband signals, the device can accurately recover the transmitted communication data.

[0050] The communication rate carried by the precoded QPSK signal was set to 200 Mbps or 400 Mbps. The QPSK-LFM signal received by the communication receiver was demodulated and a constellation diagram was drawn to verify the demodulation effect. Figure 6 (a) and Figure 6 (b) Constellation diagrams were drawn for communication rates of 200 Mbps and 400 Mbps. As can be seen from the diagrams, data demodulation can be achieved without errors under appropriate decision thresholds.

[0051] In summary, this invention utilizes microwave photonics quadruple harmonic technology to generate integrated high-frequency, broadband radar communication signals. By employing a resampling phase compensation method, the radar receiver can effectively overcome the limitations imposed by communication symbol length on target range measurement and ISAR imaging. This invention features a simple and compact structure, excellent system tunability, and significant strategic and application value.

[0052] In summary, the above-described embodiments are merely one example of the present invention and are not intended to limit the scope of protection of the present invention. It should be noted that those skilled in the art can make several equivalent modifications and substitutions based on the content disclosed in the present invention. For example, using a binary phase shift keying linear frequency modulation (BPSK-LFM) signal as the integrated radar and communication signal, the radar and communication integrated device and method based on resampling phase compensation proposed in this invention only requires slight modifications to achieve radar and communication functions. These equivalent modifications and substitutions, as well as adjustments to the signal format, should also be considered within the scope of protection of this invention.

Claims

1. A radar-communication integrated device based on resampling phase compensation, characterized in that, The device includes a continuous-wave laser, a first polarization controller, a first Mach-Zehnder modulator, a second polarization controller, a second Mach-Zehnder modulator, an arbitrary waveform generator, an optical bandstop filter, an erbium-doped fiber amplifier, an optical bandpass filter, a photodetector, an electrical bandpass filter, an electrical amplifier, a power divider, a first mixer, an antenna, a first low-noise amplifier, a second low-noise amplifier, a second mixer, a microwave signal source, and a real-time oscilloscope. The output port of the continuous-wave laser is connected to one port of the first polarization controller, and the other port of the first polarization controller is connected to the first Mach-Zehnder modulator. The input port of the first Mach-Zehnder modulator is connected to the input port of the second polarization controller. The output port of the first Mach-Zehnder modulator is connected to one port of the second polarization controller, and the other port of the second polarization controller is connected to the input port of the second Mach-Zehnder modulator. The two output ports of the arbitrary waveform generator are respectively connected to the RF input ports of the first and second Mach-Zehnder modulators. The output port of the second Mach-Zehnder modulator is connected to the input port of the optical bandstop filter. The output port of the optical bandstop filter is connected to the input port of the erbium-doped fiber amplifier. The output port of the erbium-doped fiber amplifier is connected to the input port of the erbium-doped fiber amplifier. The input port of the optical bandpass filter is connected to the input port of the photodetector; the output port of the photodetector is connected to the input port of the electrical bandpass filter; the output port of the electrical bandpass filter is connected to the input port of the electrical amplifier; the output port of the electrical amplifier is connected to the input port of the power divider; the two output ports of the power divider are respectively connected to the antenna and the local oscillator port of the first mixer; the echo signal reflected from the target is received by the antenna and connected to the input port of the first low-noise amplifier; the output port of the first low-noise amplifier is connected to the RF input port of the first mixer; the intermediate frequency output port of the first mixer is connected to the real-time oscilloscope; the multi-level phase-shift keying linear frequency modulation signal (MPSK-LFM signal) received at the communication terminal is connected to the input port of the second low-noise amplifier; the output port of the second low-noise amplifier is connected to the RF port of the second mixer; the output port of the microwave signal source is connected to the local oscillator port of the second mixer; the intermediate frequency output port of the second mixer is connected to the input port of the real-time oscilloscope; the data to be transmitted is obtained by digitally demodulating the data sampled by the real-time oscilloscope; wherein: The carrier frequency of the precoded vector signal loaded onto the first Mach-Zehnder modulator is the same as the center frequency of the linear frequency modulated signal loaded onto the second Mach-Zehnder modulator, and the center frequency of the optical bandstop filter is the same as the frequency of the output optical signal of the continuous wave laser.

2. The radar-communication integrated device based on resampling phase compensation according to claim 1, characterized in that, The phase of the precoded vector signal is half that of the conventional multi-level phase shift keying signal. Changing the modulation format of the precoded vector signal can realize the generation of integrated radar communication signals in different formats.

3. A radar-communication integrated method based on resampling phase compensation using the device as described in claim 1, characterized in that, The method includes the following steps: 1) The output frequency of the continuous wave laser is f C The optical carrier is injected into the first Mach-Zehnder modulator after passing through the first polarization controller. A precoded vector signal output by the arbitrary waveform generator is used to drive the first Mach-Zehnder modulator. The bias voltage of the first Mach-Zehnder modulator is adjusted to make it work at the minimum bias point in order to suppress carrier double-sideband modulation. 2) The suppressed carrier double-sideband signal output from the first Mach-Zehnder modulator is injected into the second Mach-Zehnder modulator after passing through the second polarization controller. Another linear frequency modulation signal output from the arbitrary waveform generator is used to drive the second Mach-Zehnder modulator. The bias voltage of the second Mach-Zehnder modulator is adjusted so that it operates at the minimum bias point. At this time, the output optical signal of the second Mach-Zehnder modulator consists of three frequency components: one is the optical sideband near the optical carrier, and the other two are two broadband optical sidebands far away from the optical carrier. 3) The optical signal output from the second Mach-Zehnder modulator is passed through an optical bandstop filter to eliminate the influence of the optical sidebands near the optical carrier on the generation of the MPSK-LFM signal. Erbium-doped fiber amplifiers are used to amplify the two broadband optical sidebands far from the optical carrier. An optical bandpass filter is used to suppress the spontaneous emission noise introduced by the erbium-doped fiber amplifier. 4) After photoelectric detection, the signal passes through an electrical bandpass filter and an electrical amplifier to generate a fourth-harmonic MPSK-LFM signal, which is used as the integrated radar communication signal. 5) The MPSK-LFM signal output by the amplifier is split into two paths by a power divider. One path is used as a reference signal for the radar receiver, and the other path is fed to the transmitting antenna for communication and target detection. 6) The echo signal reflected by the target is collected by the receiving antenna and then connected to the radio frequency port of the first mixer after passing through the first low noise amplifier. The intermediate frequency output port of the first mixer is sampled by a real-time oscilloscope. Then, the resampling phase compensation method is used to overcome the problem that the time delay of the echo signal relative to the reference signal is close to or exceeds the communication symbol length, which makes it impossible to realize target distance measurement and inverse synthetic aperture radar (ISAR) imaging. 7) The communication receiver first amplifies the signal received by the antenna using a second low-noise amplifier, then uses a microwave signal source to downconvert the amplified MPSK-LFM signal, then uses a real-time oscilloscope to sample the downconverted signal, and finally obtains the communication data carried by the MPSK-LFM signal through digital domain coherent demodulation.

4. The radar-communication integrated method based on resampling phase compensation according to claim 3, characterized in that, Step 6) The resampling phase compensation method is as follows: the resampling rate and the symbol rate are numerically the same. The phase to be compensated is determined based on the time delay of the echo signal relative to the reference signal, the resampling position, and the transmitted communication data, thereby eliminating the influence of the additional phase contained in the descrambling signal on target distance measurement and ISAR imaging.

Images