A wideband control method of a large-capacity low-switching-frequency harmonic treatment device

Abstract

The application relates to a wide-band control method of a large-capacity low-switching-frequency harmonic treatment device, belongs to the technical field of power electronics, and solves the problem of compensation bandwidth limitation of the large-capacity low-switching-frequency harmonic treatment device; the technical scheme of the application is to reduce the harmonic extraction error caused by high delay of the low-switching-frequency system, to sample by using a phase-locked loop of a SOGI filter, to design a two-order generalized integrator with phase correction to extract each harmonic of a load, to compensate the phase error of each harmonic, to design a combined control strategy of a current loop controller composed of a proportional control and a multiple quasi-proportional-resonance controller with phase compensation in current closed-loop control, to compensate the control gain of high-frequency harmonics through high-frequency harmonic phase compensation, and to effectively expand the control bandwidth of the large-capacity low-switching-frequency harmonic treatment device; the control mode of the application is simple and easy to realize, and the harmonic tracking and compensation effect is obviously superior to that of a traditional method without prediction and phase correction.

Description

Technical Field

[0001] This invention belongs to the field of power electronics technology and relates to a wideband control method for a large-capacity, low-switching-frequency harmonic mitigation device. Background Technology

[0002] As the harmonic power capacity in microgrids continues to increase, the installed capacity of harmonic mitigation equipment is also rapidly expanding. Traditional compensation equipment has a relatively small capacity; large-capacity harmonic mitigation systems (greater than 200A) typically employ a parallel configuration of multiple smaller-capacity devices. Compared to multiple parallel configurations, single-unit large-capacity low-switching-frequency harmonic mitigation devices offer advantages such as lower cost, simpler control, and less likelihood of resonance with the grid. However, single-unit large-capacity harmonic mitigation devices usually require operation at lower switching frequencies, and their capacity and compensation frequency range cannot meet the demands of megawatt-level microgrid systems. Controlling large-capacity low-switching-frequency harmonic mitigation devices has always been a challenge, requiring compromises between control bandwidth and switching losses for high-current devices. Traditional large-capacity low-switching-frequency harmonic mitigation devices typically use switching frequencies above 10kHz to compensate for high-frequency harmonics, but high switching frequencies significantly increase switching losses, and reducing the switching cycle necessitates lower computational load on the current controller. Because the inductive electromotive force generated by high current is relatively large, high-current systems using high switching frequencies require faster turn-on and turn-off processes, placing higher demands on the devices due to the resulting transient voltages. Because low switching frequencies correspond to large sampling delays, the cutoff frequency of the entire system is low, making it impossible to effectively implement high proportional feedback control and resulting in poor system stability. Therefore, wideband control of large-capacity low-switching-frequency harmonic mitigation devices is a significant technical challenge. Summary of the Invention

[0003] The technical problem to be solved by this invention is how to design a wideband control method for a large-capacity, low-switching-frequency harmonic mitigation device, so as to solve the problem of the compensation bandwidth limitation of the large-capacity, low-switching-frequency harmonic mitigation device.

[0004] The present invention solves the above-mentioned technical problems through the following technical solutions:

[0005] A broadband control method for a high-capacity, low-switching-frequency harmonic mitigation device includes the following steps:

[0006] S1. Collect the grid connection point voltage of the harmonic mitigation device, and extract the frequency and phase of the fundamental voltage using a phase-locked loop with an SOGI filter; collect the DC side voltage of the inverter, and input the difference between the rated DC value and the actual value into the PI DC controller for closed-loop control. The output of the PI DC controller is used as the fundamental current amplitude reference; multiply the fundamental current amplitude reference by the sine value of the fundamental voltage phase extracted by the phase-locked loop of the SOGI filter to obtain the fundamental current reference.

[0007] S2. The load current is collected and the load harmonics are extracted by a second-order generalized integrator with phase correction. The extracted harmonics are accumulated to obtain the harmonic current reference. The fundamental current reference and the harmonic current reference are added together to obtain the complete current reference signal.

[0008] S3. Acquire the output inverter current and perform first-order prediction interpolation on the instantaneous values ​​of the output inverter current and the grid connection point voltage. The difference between the interpolated value of the output inverter current and the current reference is controlled by a current loop controller composed of a proportional control and a multi-quasi-proportional resonant controller with phase compensation. The output result of the current loop controller plus the interpolated value of the grid connection point voltage is used to obtain the modulation wave signal. The modulation wave signal is input to the PWM modulator to modulate it into a control signal.

[0009] Furthermore, the calculation formula for the fundamental current reference mentioned in step S1 is as follows:

[0010]

[0011] in, As a reference for the fundamental current, As a reference for the fundamental current amplitude, θ PCC The fundamental voltage phase extracted by the phase-locked loop of the SOGI filter.

[0012] Furthermore, the calculation formula for the fundamental current amplitude reference is as follows:

[0013]

[0014] Among them, G DC (s) is the transfer function of the PI DC controller, k p,DC k is the proportional gain of the PI DC controller. i,DC The integral coefficient of the PI DC controller is . The design rating for the DC side voltage is V. DC,pre This is the actual value of the DC side voltage.

[0015] Furthermore, the calculation formula for extracting each harmonic of the load using a second-order generalized integrator with phase correction described in step S2 is as follows:

[0016] I load,n,p =PSOGI n (s)·I load

[0017] Among them, I load,n,p To extract each harmonic, PSOGI n (s) is the transfer function of a second-order generalized integrator with phase correction, I load This is the load current.

[0018] Furthermore, the transfer function of the second-order generalized integrator with phase correction is:

[0019]

[0020] Among them, T s ω is the switching period, n is the harmonic order, N is the multiple of the estimated delay relative to the switching period. r ω is the frequency of the fundamental voltage. b The bandwidth coefficient for adjusting the tracking rate of a second-order generalized integrator with phase correction.

[0021] Furthermore, the complete formula for calculating the current reference is as follows:

[0022]

[0023] in, As a reference for harmonic current, For complete current reference.

[0024] Furthermore, the calculation formula for first-order prediction extrapolation of the instantaneous values ​​of the output inverter current and the grid connection point voltage described in step S3 is as follows:

[0025] I L1,pre =H feed (s)·I L1 V PCC,pre =H feed (s)·V PCC

[0026] H feed (s)=y(n) / x(n),y(n)=2x(n)-x(n-1)

[0027] Among them, I L1,pre V is the interpolated value of the inverter output current. PCC,pre I is the interpolated value of the grid-connected node voltage. L1 V is the instantaneous value of the output inverter current. PCC This is the instantaneous value of the voltage at the grid connection point.

[0028] Furthermore, the transfer function of the current loop controller composed of the proportional control and the multiple quasi-proportional resonant controller with phase compensation described in step S3 is:

[0029]

[0030] Where, k pi and k i,n These represent the ratio and resonance coefficient of the current loop, respectively, ω c For the bandwidth of the current resonant controller, ω r To extract the frequency of the fundamental voltage, is the phase correction value, and n is the harmonic order.

[0031] Furthermore, the calculation formula for the modulated wave signal is as follows:

[0032]

[0033] in, It is a modulated wave signal.

[0034] The advantages of this invention are:

[0035] The technical solution of this invention aims to reduce the harmonic extraction error caused by high delay in low switching frequency systems. It employs a phase-locked loop (PLL) with an SOGI filter for sampling and designs a second-order generalized integrator with phase correction to extract each harmonic of the load, compensating for the phase error of each harmonic. In the current closed-loop control, a joint control strategy is designed, consisting of a proportional control and a multiple quasi-proportional resonant controller with phase compensation. High-frequency harmonic phase compensation compensates for the control gain of high-frequency harmonics, effectively expanding the control bandwidth of large-capacity low switching frequency harmonic mitigation devices. The control method of this invention is simple and easy to implement, and its harmonic tracking and compensation effects are significantly superior to traditional methods without prediction or phase correction. Attached Figure Description

[0036] Figure 1 This is a topology diagram of a high-power three-phase four-wire large-capacity low-switching-frequency harmonic mitigation device in a microgrid according to an embodiment of the present invention.

[0037] Figure 2 The circuit topology and control block diagram of the control system of the large-capacity low-switching-frequency harmonic mitigation device in the microgrid according to an embodiment of the present invention are shown below.

[0038] Figure 3 This is a structural analysis diagram of the LCL according to an embodiment of the present invention;

[0039] Figure 4 This is a functional structure decomposition diagram of the method proposed in this invention;

[0040] Figure 5 This is a control structure diagram of PCSOGI according to an embodiment of the present invention;

[0041] Figure 6 The Bode plot of the system closed-loop control transfer function of the large-capacity low-switching-frequency harmonic mitigation device according to an embodiment of the present invention is shown.

[0042] Figure 7 This is a diagram showing the output voltage and current of each branch of the grid-connected system without harmonic phase compensation according to an embodiment of the present invention.

[0043] Figure 8The embodiments of the present invention provide the output voltage and current of each unit in the grid-connected system with phase compensation.

[0044] Figure 9 This is a diagram showing the THD results of the load harmonic current requiring compensation in an embodiment of the present invention.

[0045] Figure 10 The diagram shows the THD results of the low switching frequency, high capacity harmonic mitigation device according to an embodiment of the present invention, using control without phase compensation and control with phase compensation. Detailed Implementation

[0046] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0047] The technical solution of the present invention will be further described below with reference to the accompanying drawings and specific embodiments:

[0048] Example 1

[0049] 1. Large-capacity low-switching-frequency harmonic mitigation device

[0050] like Figure 1 The diagram shows the topology of a high-power three-phase four-wire, high-capacity, low-switching-frequency harmonic mitigation device in a microgrid. The device consists of IGBT modules and a passive filter. The passive filter is a low-impedance LCL filter with a switching frequency of 3kHz, and oversampling is used for sampling.

[0051] The microgrid nonlinear load and the large-capacity low-switching-frequency harmonic mitigation device in grid-connected mode are connected to the main grid via AC bus. The large-capacity low-switching-frequency harmonic mitigation device adopts a "1"-shaped three-level structure, which has advantages such as large output capacity, high output voltage, and low current harmonic content. The nonlinear load adopts a typical three-phase uncontrolled rectifier load. Unlike a three-phase three-wire system where only two phases are freely adjustable, each phase of a three-phase four-wire system is a relatively independent system. Furthermore, the DC bus voltage control has little impact on the harmonic frequency current control of the large-capacity low-switching-frequency harmonic mitigation device. Therefore, the three-phase four-wire system can be decomposed into a single-phase system for analysis. The output of the large-capacity low-switching-frequency harmonic mitigation device is connected to the system PCC node via an LCL filter. To simplify the analysis, the single-phase system topology of the large-capacity low-switching-frequency harmonic mitigation device is taken as follows: Figure 2 As shown.

[0052] Figure 3 This is an exploded view of the LCL circuit. Each phase's LCL filter consists of an inverter-side inductor L1, a grid-side inductor L2, and a parallel filter capacitor C1. To reduce switching ripple, passive damping (using damping resistor R) is added to the capacitor branch of the LCL filter. f (This is combined with a parallel capacitor C2 in series) to reduce high-frequency resonance. The impedance after adding passive damping with the parallel capacitor is Z. C (s), DC capacitor voltage is V DC The physical signals to be collected for phase l (l = a, b, c) include the grid current I. grid,l The output current I of the high-capacity low-switching-frequency harmonic mitigation device line,l Inverter-side current I L1,l and load current I load,l To simplify modeling and better demonstrate the universality of the three-phase control principle, I... line,l I L1,l and I load,l and PCC voltage V PCC,l After sampling, it is abbreviated as I. line I L1 and I load and PCC voltage V PCC .

[0053] 2. Wideband Control Method for Large-Capacity Low-Switching-Frequency Harmonic Mitigation Devices

[0054] (1) Due to the long system sampling delay, it is difficult to achieve system stability by directly using proportional control. This invention proposes to correct the sampling signal, I line I L1 and I load and PCC voltage V PCC The signal is I after sampling and correction. line,pre I L1,pre and I load,pre and V PCC,pre This facilitates system stability when adjusting the proportional controller.

[0055] Voltage V at the grid connection point of the harmonic mitigation device PCC The frequency ω of the fundamental voltage is extracted using a phase-locked loop with an SOGI filter. r and phase θ PCC ;

[0056] Collect the DC side voltage of the inverter and convert it to the DC design rating. Compared with the actual value V DC,pre The difference input to the PI DC controller G DC (s) performs closed-loop control, PI DC controller G DC The output of (s) is used as a reference for the fundamental current amplitude:

[0057]

[0058] in, As a reference for the fundamental current amplitude, k p,DC For PI DC controller G DC The proportionality coefficient of (s), k i,DC For PI DC controller G DC The integral coefficient of (s);

[0059] Multiplying the fundamental current amplitude reference by the sine of the fundamental voltage phase extracted by the phase-locked loop of the SOGI filter yields the fundamental current reference:

[0060]

[0061] in, As a reference for the fundamental current, θ PCC The fundamental voltage phase extracted by the phase-locked loop of the SOGI filter.

[0062] (2) The load current is acquired, and the load harmonic components are extracted using a second-order generalized integrator with phase correction (PCSOGI) algorithm. The harmonic frequencies are integer multiples of the fundamental frequency reference value obtained in step 2. The harmonic current reference is obtained by summing the extraction results of the 3rd, 5th, and 7th harmonics. The complete current reference signal is obtained by adding the fundamental current reference and the harmonic current reference. The calculation formula for the harmonic current reference is as follows:

[0063] I load,n,p =PSOGI n (s)·I load

[0064] Among them, I load,n,p PSOGI ​​serves as a harmonic current reference. n (s) is the transfer function of a second-order generalized integrator with phase correction, I load This is the load current.

[0065] Figure 4 The diagram shows the functional structure decomposition of the method proposed in this invention. The harmonic extraction part employs a phase correction method, the current loop sampling signal uses a predictive correction method, and the current loop controller uses a resonant controller with phase compensation. Firstly, because the sampling and output delay times are both large at low switching frequencies, this invention proposes using a second-order generalized integrator with phase correction (Phase Correction SOGI, PCSOGI) to adjust the current loop signal. load Harmonics in the signal can simultaneously achieve signal extraction and phase correction.

[0066] Figure 5The transfer function of the second-order generalized integrator with phase correction shown is:

[0067]

[0068] Among them, T s ω is the switching period, n is the harmonic order, N is the multiple of the estimated delay relative to the switching period. r ω is the frequency of the fundamental voltage. b The bandwidth coefficient for adjusting the tracking rate of a second-order generalized integrator with phase correction.

[0069] Target reference current The fundamental reference current Harmonic reference current sum:

[0070]

[0071] (3) The inverter current is acquired, and then the instantaneous values ​​of the inverter current signal and the grid connection point voltage signal are subjected to first-order prediction interpolation. The difference between the interpolated value of the inverter current and the current reference signal is controlled by a current loop controller of a multi-quasi-proportional resonant controller group with phase compensation. The output of the current loop controller is added to the interpolated value of the grid connection point voltage to obtain the modulation signal. The modulation signal is input to the PWM modulator, and the modulation result is the control signal of the IGBT module.

[0072] The inverter-side inductor current signal is acquired, and then the instantaneous values ​​of the inverter current and the grid connection point voltage are subjected to first-order prediction interpolation. The specific calculation formula is as follows:

[0073] I L1,pre =H feed (s)·I L1 V PCC,pre =H feed (s)·V PCC

[0074] H feed (s)=y(n) / x(n),y(n)=2x(n)-x(n-1)

[0075] Among them, I L1,pre V is the interpolated value of the inverter output current. PCC,pre I is the interpolated value of the grid-connected node voltage. L1 V is the instantaneous value of the output inverter current. PCC This is the instantaneous value of the voltage at the grid connection point;

[0076] The transfer function of the current loop controller composed of the proportional control plus the phase-compensated multiple quasi-proportional resonant controller is:

[0077]

[0078] Where, k pi and k i,n These represent the ratio and resonance coefficient of the current loop, respectively, ω c For the bandwidth of the current resonant controller, ω r To extract the frequency of the fundamental voltage, is the phase correction value, and n is the harmonic order.

[0079] The formula for calculating the modulated wave signal is as follows:

[0080]

[0081] in, It is a modulated wave signal.

[0082] The modulation signal is input to the PWM modulator, and the modulation result is the control signal for the IGBT module.

[0083] 3. Simulation verification

[0084] The algorithm for a high-capacity harmonic mitigation device with low switching frequency was verified using the system and control parameters in Table 1. A fully discretized simulation of the control process was performed using MATLAB / SIMULINK. The switching frequency was 3kHz, and the sampling frequency was 24kHz. The sampling process of the physical system was simulated by updating the acquisition signal through a counter interruption method. The modulation process of the system was simulated using a fully discretized method with a unified clock and control modulation algorithm, where the clock was 0.6MHz. To avoid injecting current into uncompensated harmonic frequencies, the resonant bandwidth was designed to be 10rad / s. The modulation stage adopted a double-edge counter comparison method. The power supply was a standard 380V three-phase four-wire system, with rated power of 1.2MW and 2.2MW for the three-phase uncontrollable loads. The peak output of the high-capacity low switching frequency harmonic mitigation device compensated for a single phase exceeding 600A, far exceeding the output capacity of traditional high-capacity low switching frequency harmonic mitigation devices.

[0085] Table 1. Key parameters of the experimental platform

[0086]

[0087]

[0088] 4. Conclusion

[0089] Figure 6 This is the Bode plot of the closed-loop transfer function of the control method proposed in this invention. Clearly, through phase compensation, the system achieves precise closed-loop control of high-frequency harmonics, which is impossible with traditional methods.

[0090] It should be noted that without predictive control, direct proportional control cannot stabilize a low-switching-frequency system. While adding predictive control allows for stable operation, it only allows for the incorporation of 5th and 7th harmonics, failing to incorporate higher-order harmonic control. Without phase compensation, the high-frequency resonant coefficient is small, resulting in poor harmonic compensation performance. The system is stable with a smaller bandwidth; even under frequently changing operating conditions, delay compensation minimizes overcompensation or undercompensation errors, preventing severe harmonic injection and thus promoting safer system operation. Traditional discretization methods generate severe amplitude attenuation and phase errors at 7th and higher harmonic frequencies, preventing higher-order harmonic compensation. Therefore, simulations without phase compensation only achieve 7th harmonic compensation with insufficient accuracy. In contrast, resonant phase compensation achieves 25th harmonic compensation, demonstrating a significant improvement.

[0091] Figure 7 The proposed control method is used to address the issue of high-capacity, low-switching-frequency harmonic mitigation in a large-capacity system with low-switching-frequency harmonics, resulting in a reduced PCC node voltage V. PCC Grid current I grid Load current I load and output current I line The voltage at the PCC node is close to the ideal voltage, while the load current exhibits severe waveform distortion.

[0092] Figure 8 The system employs a phase-less harmonic compensation algorithm (compensation range cutoff at the 7th order) to measure the grid current and output current of a large-capacity, low-switching-frequency harmonic mitigation device. Clearly, harmonic phase compensation significantly improves the system's harmonic compensation range and accuracy.

[0093] Figure 9 The load current THD was 18.31%. With the proposed harmonic phase compensation algorithm, the grid current THD decreased to 2.16%, while the grid current THD without harmonic phase compensation remained above 11%. The THD results are as follows: Figure 10 As shown in the figure. In summary, the simulation results fully demonstrate that the proposed control algorithm can realize harmonic compensation of a large-capacity low-switching-frequency harmonic mitigation device under low switching frequencies. The phase compensation control technology of high-frequency resonance can significantly extend the control bandwidth and help the system obtain good transient and steady-state performance.

[0094] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A wideband control method of a large-capacity low-switching-frequency harmonic management device, characterized by, Includes the following steps: S1. Collect the grid connection point voltage of the harmonic mitigation device, and extract the frequency and phase of the fundamental voltage using a phase-locked loop with an SOGI filter; collect the DC side voltage of the inverter, and input the difference between the rated DC value and the actual value into the PI DC controller for closed-loop control. The output of the PI DC controller is used as the fundamental current amplitude reference; multiply the fundamental current amplitude reference by the sine value of the fundamental voltage phase extracted by the phase-locked loop of the SOGI filter to obtain the fundamental current reference. S2. The load current is collected and the load harmonics are extracted by a second-order generalized integrator with phase correction. The extracted harmonics are accumulated to obtain the harmonic current reference. The fundamental current reference and the harmonic current reference are added together to obtain the complete current reference signal. The transfer function of the second-order generalized integrator with phase correction is: in, For the switching cycle, For harmonic order, To estimate the multiple of the delay relative to the switching period, The frequency of the fundamental voltage. The bandwidth coefficient for adjusting the tracking rate of a second-order generalized integrator with phase correction; S3. Acquire the output inverter current and perform first-order prediction interpolation on the instantaneous values ​​of the output inverter current and the grid connection point voltage. The difference between the interpolated value of the output inverter current and the complete current reference is obtained by a current loop controller composed of a proportional control and a multi-quasi-proportional resonant controller with phase compensation. The output result of the current loop controller plus the interpolated value of the grid connection point voltage is used to obtain the modulation wave signal. The modulation wave signal is input to the PWM modulator to modulate it into a control signal. The calculation formula for first-order prediction extrapolation of the instantaneous values ​​of the output inverter current and the grid connection point voltage is as follows: in, This is the interpolated value of the inverter output current. This is the interpolated value of the grid-connected node voltage. The instantaneous value of the output inverter current. This is the instantaneous value of the voltage at the grid connection point.

2. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 1, characterized in that, The formula for calculating the fundamental current reference mentioned in step S1 is as follows: in, As a reference for the fundamental current, As a reference for the fundamental current amplitude, The fundamental voltage phase extracted by the phase-locked loop of the SOGI filter.

3. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 2, characterized in that, The formula for calculating the fundamental current amplitude reference is as follows: in, This is the transfer function of the PI DC controller. The proportional gain of the PI DC controller. The integral coefficient of the PI DC controller is . Design ratings for DC-side voltage. This is the actual value of the DC side voltage.

4. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 3, characterized in that, The calculation formula for extracting the load harmonics using a second-order generalized integrator with phase correction described in step S2 is as follows: in, For the extracted harmonics, Let the transfer function be a second-order generalized integrator with phase correction. This is the load current.

5. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 4, characterized in that, The complete formula for calculating the current reference is as follows: in, As a reference for harmonic current, For complete current reference.

6. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 5, characterized in that, The transfer function of the current loop controller composed of proportional control and a multi-quasi-proportional resonant controller with phase compensation described in step S3 is: in, and These are the ratio and resonance coefficient of the current loop, respectively. For the bandwidth of the current resonant controller, To extract the frequency of the fundamental voltage, is the phase correction value, and n is the harmonic order.

7. The broadband control method for a large-capacity, low-switching-frequency harmonic mitigation device according to claim 6, characterized in that, The formula for calculating the modulated wave signal is as follows: in, It is a modulated wave signal.

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