Hybrid modulation method for stacked bridge based dual active bridge converter

Abstract

This invention discloses a hybrid modulation method for a dual active bridge converter based on a stacked bridge. The invention achieves primary-secondary voltage matching over a wide voltage range by modulating the duty cycle of the stacked bridge, and controls the power flow of the primary and secondary sides through phase-shift modulation. This control method not only expands the soft-switching range but also reduces the effective value of the current, conduction losses, and current stress. Furthermore, this control method is simple, reduces the computational optimization requirements of the controller, and is more beneficial for industrial applications. Moreover, the converter proposed in this invention can achieve a large voltage gain across the dual active bridge with a smaller transformer turns ratio, reducing transformer losses in high step-up / step-down ratio applications.

Description

Technical Field

[0001] This invention relates to a modulation method for a dual active bridge (DAB) converter. Background Technology

[0002] The DAB converter is a widely used interface converter between the DC bus and the battery in microgrids. In the DAB converter, phase-shift modulation can flexibly regulate the bidirectional power flow. However, due to the wide voltage range of the battery, severe voltage mismatch may occur. This mismatch leads to a reduction in the zero-voltage switching (ZVS) range and an increase in the effective and maximum current values. To increase the soft-switching range and reduce the effective current value, researchers have proposed modulation methods such as dual phase-shift (DPS), extended phase-shift (EPS), and triple phase-shift (TPS) to optimize the control strategy of dual active bridge converters, and continuously calculate the most efficient operating method under different operating conditions through optimized algorithms. However, various performance objectives usually cannot be achieved simultaneously. Due to the extreme complexity of its objective function, obtaining the optimal solution is often very complicated, making it difficult to achieve efficiency optimization and apply it in practical engineering. Furthermore, when the DC bus voltage is much higher than the battery voltage, a high turns ratio transformer is required, resulting in higher transformer losses and increasing the design difficulty and cost. In high-voltage applications, expensive power semiconductors are typically required to withstand high-voltage stress, which also increases circuit costs. Summary of the Invention

[0003] The technical problem to be solved by this invention is that in DAB converters, phase shift modulation will cause severe voltage mismatch, and existing solutions are difficult to apply in engineering practice and have high circuit costs.

[0004] To address the aforementioned technical problems, the present invention provides a hybrid modulation method for a dual active bridge converter based on a stacked bridge, characterized in that the circuit topology of the dual active bridge converter based on the stacked bridge includes a primary-side stacked bridge and a transformer T. r The primary-side stacked bridge circuit includes four switches S1 to S4 connected in series. The drain of switch S1 is connected to the positive terminal of the converter input, and the source of switch S4 is connected to the reference ground of the converter input. Support capacitors C1 and C2 are connected in series between the positive and negative terminals of the primary-side DC bus. Support capacitor C1 is connected to the positive terminal of the primary-side DC bus, and support capacitor C2 is connected to the negative terminal of the primary-side DC bus. The connection point of support capacitors C1 and C2 is connected to the connection point of switches S2 and S3. The capacitance values ​​of support capacitors C1 and C2 are the same. The connection point of switches S1 and S2 is connected to the connection point of support capacitor C1 and C2. p The first end is connected to the supporting capacitor C. p The second terminal is connected to the inductor Ls The first terminal is connected, inductor L s The second end is connected to transformer T r The first end of the primary winding of transformer T r The second end of the primary winding is connected to the connection point of switching transistors S3 and S4; the secondary full-bridge circuit includes four switching transistors S5 to S8, with S5 and S6 connected in series between the positive and negative terminals of the converter's output, and S7 and S8 also connected in series between the positive and negative terminals of the converter's output; transformer T r The first end of the secondary winding is connected to the center point of switching transistors S5 and S6. Transformer T r If the second end of the secondary winding is connected to the center point of switching transistors S7 and S8, then the hybrid modulation method includes the following steps:

[0005] Step 1: Input voltage V p and output voltage V s The voltage conversion ratio M is obtained by sampling and calculation;

[0006] Step 2: Calculate the duty cycle D of the primary active bridge based on the voltage conversion ratio M, where D = 1 - 2M;

[0007] Step 3: Adjust the output voltage V s and output current i s Sampling is performed, and the calculated power value P is compared with the reference power value P. ref The phase shift angles of the primary and secondary sides are compared and obtained through closed-loop adjustment. Where P = V s i s Phase shift angle The difference between the conduction times of switching transistors S1 and S5 is the power flow direction on the primary and secondary sides, determined by the phase shift angle. adjust;

[0008] Step 4: Based on the duty cycle D and phase shift angle Control signals are generated for switching transistors S1 to S8 to control their on / off states.

[0009] Preferably, the voltage conversion ratio M is calculated using the following formula:

[0010] M = 2nV s / V p

[0011] In the formula, n:1 represents the transformer T. r The turns ratio.

[0012] Preferably, in step four, based on the phase shift angle Carriers are generated for input to the primary-side PWM generator and the secondary-side PWM generator, respectively. The primary-side PWM generator generates control signals for switching transistors S1 to S4, and the secondary-side PWM generator generates control signals for switching transistors S5 to S8. The carriers input to the primary-side PWM generator and the carriers input to the secondary-side PWM generator are phase-differential.

[0013] Preferably, the drive signals of switching transistors S1 and S2 are complementary, and the drive signals of switching transistors S3 and S4 are complementary. Therefore, based on the operating state of the primary side, the primary-side switching transistors S1 to S4 have two sets and four states:

[0014] State 1: Switch S1 and switch S4 are both turned on, while switch S2 and switch S3 are turned off.

[0015] State 2: Switch S2 and switch S4 are both turned on, while switch S1 and switch S3 are turned off.

[0016] State 3: Switch S2 and switch S3 are both turned on, while switch S1 and switch S4 are turned off.

[0017] State 4: Switch S1 and switch S3 are both turned on, while switch S2 and switch S4 are turned off.

[0018] In state one, the duty cycle is always 50% in one switching cycle. In states two and four, the duty cycle is the same in one switching cycle and this duty cycle is called D. In state three, the duty cycle is 1-D in one switching cycle.

[0019] The first group of switches operates in three phases: state one with a 50% duty cycle, state two with a duty cycle of D, and state three with a duty cycle of 1-D.

[0020] The second set of switches operates in cycles including state one with a 50% duty cycle, state four with a duty cycle of D, and state three with a duty cycle of 1-D.

[0021] The alternation of the first and second sets of switching states constitutes the switching state of the primary side. Based on the switching state of the primary-side switching transistor, the primary side can generate three voltage levels.

[0022] Preferably, the duty cycle of the secondary side switches S5 to S8 is 50%, and switches S5 and S6 are complementary in conduction, and switches S7 and S8 are complementary in conduction; switches S5 and S8 are simultaneously in conduction, and switches S6 and S7 are simultaneously in conduction; the secondary side can generate a two-level square wave.

[0023] This invention achieves primary-secondary voltage matching over a wide voltage range by modulating the duty cycle of the stacked bridge, and controls the power flow of the primary and secondary sides through phase-shift modulation. This control method not only expands the soft-switching range but also reduces the effective value of the current, conduction losses, and current stress. Furthermore, this control method is simple, reduces the computational optimization requirements of the controller, and is more beneficial for industrial applications. Moreover, the converter proposed in this invention can achieve a large voltage gain across the dual active bridge with a smaller transformer turns ratio, reducing transformer losses in high step-up / step-down ratio applications. Attached Figure Description

[0024] Figure 1 This is a circuit schematic diagram of a dual active bridge converter according to an embodiment of the present invention;

[0025] Figure 2 This is a waveform diagram of the dual active bridge converter operating in one mode according to an embodiment of the present invention;

[0026] Figures 3(a)-3(d) Four mode diagrams of the stacked bridge of a dual active bridge converter according to an embodiment of the present invention;

[0027] Figures 4(a)-4(d) According to an embodiment of the present invention Figure 1 The working waveform diagrams of the four modes;

[0028] Figure 5 This is a control logic diagram according to an embodiment of the present invention;

[0029] Figure 6 This is a schematic diagram of the primary and secondary voltages and inductor current in an embodiment of the present invention when power is transmitted in the forward direction at 800V on the primary side and 200V on the secondary side.

[0030] Figure 7 This is a schematic diagram of the waveforms of the primary and secondary voltages and the inductor current when the power is transmitted in the forward direction at 800V on the primary side and 100V on the secondary side in an embodiment of the present invention.

[0031] Figure 8 This is a schematic diagram of the waveforms of the primary and secondary voltages and the inductor current when the power is transmitted in the forward direction at 800V on the primary side and 150V on the secondary side in an embodiment of the present invention.

[0032] Figure 9 This is a schematic diagram of the waveforms of the primary and secondary voltages and the inductor current when power is transmitted in reverse at 800V on the primary side and 150V on the secondary side in an embodiment of the present invention. Detailed Implementation

[0033] The present invention will be further illustrated below with reference to specific embodiments. It should be understood that these embodiments are for illustrative purposes only and are not intended to limit the scope of the invention. Furthermore, it should be understood that after reading the teachings of this invention, those skilled in the art can make various alterations or modifications to the invention, and these equivalent forms also fall within the scope defined by the appended claims.

[0034] Figure 1 The circuit schematic of the dual active bridge based on stacked bridges provided in this embodiment is shown below. Figure 1 As shown, the dual active bridge converter includes a stacked bridge on the primary side, a full-bridge circuit on the secondary side, and an inductor L. s Transformer T r And supporting capacitors C1, C2, C p The primary-side stacked bridge includes four switching transistors S1 to S4, and the secondary-side full-bridge circuit includes four switching transistors S5 to S8. Switches S1 to S4 are connected in series. The drain of switch S1 is connected to the positive terminal of the converter input, and the source of switch S4 is connected to the negative terminal of the converter input.

[0035] Support capacitors C1 and C2 are connected in series between the positive and negative terminals of the primary DC bus, with C1 connected to the positive terminal and C2 connected to the negative terminal. The connection point of support capacitors C1 and C2 is connected to the connection point of switching transistors S2 and S3. Since support capacitors C1 and C2 have the same capacitance, their voltages are equal in steady state, both equal to the input voltage V. in Half of it.

[0036] The connection point of switching transistors S1 and S2 and the supporting capacitor C p The first end is connected to the supporting capacitor C. p The second terminal is connected to the inductor L s The first terminal is connected, inductor L s The second end is connected to transformer T r The first end of the primary winding is connected to the first terminal, and the second end of the primary winding is connected to the connection point of switching transistors S3 and S4. Support capacitor C p The capacitance value is large enough to support the DC bias voltage. In steady state, the supporting capacitor C... p The terminal voltage is assumed to remain constant.

[0037] In the secondary-side full-bridge circuit, switches S5 and S6 are connected in series between the positive and negative terminals of the converter's output, and switches S7 and S8 are also connected in series between the positive and negative terminals of the converter's output. Transformer T r The first end of the secondary winding is connected to the center point of switching transistors S5 and S6. Transformer T r The second end of the secondary winding is connected to the center point of the switching transistors S7 and S8.

[0038] Figure 2 The waveform shown is for steady-state conditions. In steady state, the voltage across capacitors C1 and C2 is equal to V. in / 2. As shown in the figure, the primary-side stacked bridge uses an asymmetric duty cycle modulation method, and the primary-side output voltage V op A three-level voltage can be modulated based on the states of the four primary-side switching transistors S1 to S4.

[0039] The specific working method of the primary-side stacked bridge is as follows: Figures 3(a) to 3(d) As shown in Figure 3(a), in state one, switches S1 and S4 are simultaneously turned on, while switches S2 and S3 are turned off. The output voltage V of the stacked bridge is shown in Figure 3(a). op Equal to input voltage V in .

[0040] In state two, switches S2 and S4 are simultaneously turned on, while switches S1 and S3 are turned off, as shown in Figure 3(b). The output voltage V of the stacked bridge is... op equals V in / 2.

[0041] In state three, switches S2 and S3 are simultaneously turned on, while switches S1 and S4 are turned off, as shown in Figure 3(c). The output voltage V of the stacked bridge is... op It equals 0.

[0042] In state four, switches S1 and S3 are simultaneously turned on, while switches S2 and S4 are turned off, as shown in Figure 3(d). The output voltage V of the stacked bridge is... op equals V in / 2.

[0043] In steady state, switches S1 and S2 conduct complementaryly, and switches S4 and S3 conduct complementaryly. Switches S1 and S4 are always on simultaneously, with no phase shift adjustment. The switching state of the primary-side stacked bridge consists of two sets of switching cycles: In the first cycle, the drive signals for switches S1 and S2 are complementary square waves with a 50% duty cycle, the drive signal for switch S4 is a square wave with a duty cycle greater than or equal to 50%, and the drive signal for switch S3 is a complementary square wave to the drive signal for switch S4. Therefore, the primary-side output voltage V... op For a group including V in V in / 2, 0 three-level voltage; (Switching cycle two) The drive signals for switches S3 and S4 are complementary square waves with a 50% duty cycle. The drive signal for switch S1 can be a square wave with a duty cycle greater than or equal to 50%, and the duty cycle of switch S1 is the same as that of switch S4 in switching cycle one. The drive signal for switch S2 is complementary to the drive signal for switch S1 and is the same as in switching cycle one. The output voltage V on the primary side in switching cycle two is... op Also for a group including V in V in / 2, 0 three-level voltage.

[0044] V can be generated in both sets of cycles. in The primary-side output voltage is 2 / 2, but it is achieved through switching state 2 in switching cycle 1, and through switching state 4 in switching cycle 2. The two sets of cycles alternate to keep the voltage of capacitors C1 and C2 the same.

[0045] The portion of switch S4 or switch S1 with a duty cycle exceeding 50% is defined as D.

[0046] The duty cycle of the secondary-side switches S5-S8 is 50%, and switches S5 and S6 are complementary in conduction, as are switches S7 and S8. Switches S5 and S8 can be turned on simultaneously, as can switches S6 and S7. A square wave voltage V is generated on the secondary side. cd like Figure 2 As shown.

[0047] To maintain the volt-second balance between the series inductor and the transformer, the primary and secondary voltages V must be kept constant. ab V cd Since the average value is the same in both the positive and negative half-cycles, the following relationship can be obtained:

[0048] 0.5T s (V p -V cp )+DT s (0.5V p -V cp )+(0.5-D)T s (-V cp ) = 0

[0049] Therefore, further derivation yields the following:

[0050] V cp =0.5(1+D)V p

[0051] V ab The average value during the positive half-cycle is V. ab,pos =0.5(1-D)Vp

[0052] To achieve voltage matching between the primary and secondary sides, we have V ab,pos =0.5(1-D)V p =nV s

[0053] Therefore, D = 1 - 2nV s / V p =1-2M.

[0054] Therefore, voltage matching can be achieved within a wide voltage range: 0.25≤M≤0.5, corresponding to 0≤D≤0.5.

[0055] Based on the phase shift angle of the primary and secondary sides The mathematical relationship with D may have four working modes, as shown in Figures 4(a) to (d).

[0056] when At that time, the steady-state waveform is shown in Figure 4(a). During the time interval t1 to t2, primary and secondary voltage matching is achieved, and the inductor L... s The current remains constant.

[0057] when At that time, the steady-state waveform is shown in Figure 4(b). The primary and secondary voltages are matched during the time period t1 to t2, and the current of inductor Ls remains unchanged.

[0058] when At that time, the steady-state waveform is shown in Figure 4(c). The primary and secondary voltages are matched during the time period from t0 to t1, and the current in the inductor Ls remains unchanged.

[0059] when At that time, the steady-state waveform is shown in Figure 4(d). The primary and secondary voltages are matched during the time period from t0 to t1, and the current in the inductor Ls remains unchanged.

[0060] By maintaining the matching of the primary and secondary voltages, the effective value of the inductor current can be effectively reduced, thereby reducing conduction losses and improving the efficiency of the converter.

[0061] When the primary-to-secondary voltage ratio M = 0.5, according to calculation, D = 0, and at this time the primary voltage V ab It is a two-level circuit, and V ab The effective value is V in / 2.

[0062] When the primary-to-secondary voltage ratio M = 0.25, according to calculation, D = 0.5, at this time the primary voltage V ab It is a two-level circuit, and V ab The effective value is V in / 4.

[0063] Under the two operating conditions mentioned above, the converter operates in single-phase-shift (SPS) regulation mode, and due to the matching of primary and secondary voltages, soft switching of all switching transistors can be achieved. In this operating mode, the circulating current is also minimized.

[0064] The control methods mentioned above include:

[0065] Step 1: Sample the input and output voltages and calculate the voltage conversion ratio M;

[0066] Step 2: Calculate the duty cycle D of the primary-side active bridge based on the voltage conversion ratio M;

[0067] Step 3: Sample the output voltage and current, compare them with the reference power value, and obtain the phase shift angle of the primary and secondary sides through closed-loop adjustment.

[0068] Step 4: Generate control signals for switching transistors S1-S8 based on the duty cycle D and the phase shift angle of the primary and secondary sides to control the switching transistors to turn on and off.

[0069] The block diagram of the digital control scheme is as follows Figure 5 As shown. The converter can be controlled by two variables D and Closed-loop control is implemented. The asymmetric duty cycle D is calculated through voltage feedback from both sides to achieve voltage matching. The real-time power of the converter is calculated by sampling the voltage and current on the output side, and the phase shift ratio is adjusted through closed-loop control. To adjust the power of the converter.

[0070] Based on the proposed circuit topology and control method, a prototype was fabricated for experimental verification. In the specific embodiment circuit, the primary-side voltage is a fixed 800V, the secondary-side voltage is a wide range of 100V-200V, the operating frequency of the switching transistor is set to 100kHz, and the turns ratio of the transformer's primary and secondary sides is 2:1. Through the proposed circuit structure and control method, a large primary-to-secondary voltage gain can be achieved with only a 2:1 transformer turns ratio. Reducing the turns ratio effectively reduces transformer losses and design complexity.

[0071] Figure 6 The diagram shows the waveforms of the primary and secondary voltages and the inductor current when the power is transmitted in the forward direction at 800V on the primary side and 200V on the secondary side in an embodiment of the present invention. Figure 7 The figure shown is a schematic diagram of the primary and secondary voltages and inductor current when the power is transmitted in the forward direction at 800V on the primary side and 100V on the secondary side in an embodiment of the present invention. Figure 8 The diagram shows the waveforms of the primary and secondary voltages and the inductor current when power is transmitted in the forward direction at 800V on the primary side and 150V on the secondary side in an embodiment of the present invention. Figure 8The diagram shows the waveforms of the primary and secondary voltages and the inductor current when power is transmitted in reverse at 800V on the primary side and 150V on the secondary side in an embodiment of the present invention.

Claims

1. A hybrid modulation method based on a stacked bridge dual active bridge converter, characterized in that, The circuit topology of the dual active bridge converter based on stacked bridges includes a primary-side stacked bridge and a transformer T. r The circuit includes a secondary-side full-bridge circuit. The primary-side stacked bridge consists of switches S1 to S4 connected in series. The drain of switch S1 is connected to the positive terminal of the primary-side DC bus. The source of switch S1 is connected to the drain of switch S2. The source of switch S2 is connected to the drain of switch S3. The source of switch S3 is connected to the drain of switch S4. The source of switch S4 is connected to the negative terminal of the primary-side DC bus. Support capacitors C1 and C2 are connected in series between the positive and negative terminals of the primary-side DC bus. The first terminal of support capacitor C1 is connected to the positive terminal of the primary-side DC bus. The second terminal of support capacitor C1 is connected to the first terminal of support capacitor C2. The second terminal of support capacitor C2 is connected to the negative terminal of the primary-side DC bus. The connection point of support capacitors C1 and C2 is connected to the connection point of switches S2 and S3. Support capacitors C1 and C2 have the same capacitance value. The connection point of switches S1 and S2 is connected to the connection point of support capacitor C1 and C2. p The first end is connected to the supporting capacitor C. p The second terminal is connected to the inductor L s The first terminal is connected, inductor L s The second end is connected to transformer T r The first end of the primary winding of transformer T r The second end of the primary winding is connected to the connection point of switching transistors S3 and S4; the secondary full-bridge circuit includes four switching transistors S5 to S8, with S5 and S6 connected in series between the positive and negative terminals of the converter's output, and S7 and S8 also connected in series between the positive and negative terminals of the converter's output; transformer T r The first end of the secondary winding is connected to the center point of switching transistors S5 and S6. Transformer T r If the second end of the secondary winding is connected to the center point of switching transistors S7 and S8, then the hybrid modulation method includes the following steps: Step 1: Input voltage V p and output voltage V s The voltage conversion ratio M is obtained by sampling and calculation; Step 2: Calculate the duty cycle D of the primary active bridge based on the voltage conversion ratio M, where D = 1 - 2M; Step 3: Adjust the output voltage V s and output current i s Sampling is performed, and the calculated power value P is compared with the reference power value P. ref The phase shift angles of the primary and secondary sides are compared and obtained through closed-loop adjustment. ,in, Phase shift angle The difference between the conduction times of switching transistors S1 and S5 is the power flow direction on the primary and secondary sides, determined by the phase shift angle. adjust; Step 4: Based on the duty cycle D and phase shift angle Control signals are generated for switching transistors S1 to S8 to control their on / off states, wherein: when At that time, the primary and secondary voltages are matched within the time period t1~t2, and the inductor L s The current remains constant; when At that time, the primary and secondary voltages are matched within the time period t1~t2, and the inductor L s The current remains constant; when During the time period t0~t1, primary and secondary voltage matching is achieved, and the inductor L... s The current remains constant; when During the time period t0~t1, primary and secondary voltage matching is achieved, and the inductor L... s The current remains constant, where t0 is the start time of the switching cycle, t1 is the time when the secondary voltage first flips, and t2 is the time when the primary voltage first flips.

2. The hybrid modulation method for a dual active bridge converter based on stacked bridges as described in claim 1, characterized in that, The voltage conversion ratio M is calculated using the following formula: In the formula, For the transformer T r The turns ratio.

3. The hybrid modulation method for a dual active bridge converter based on stacked bridges as described in claim 1, characterized in that, In step four, based on the phase shift angle Carriers are generated for input to the primary-side PWM generator and the secondary-side PWM generator, respectively. The primary-side PWM generator generates control signals for switching transistors S1-S4, and the secondary-side PWM generator generates control signals for switching transistors S5-S8. The carriers input to the primary-side PWM generator and the carriers input to the secondary-side PWM generator are phase-differential. .

4. The hybrid modulation method for a dual active bridge converter based on stacked bridges as described in claim 1, characterized in that, The drive signals of switching transistors S1 and S2 are complementary, and the drive signals of switching transistors S3 and S4 are complementary. Therefore, based on the operating state of the primary side, the primary-side switching transistors S1~S4 have two sets and four states: State 1: Switch S1 and switch S4 are both turned on, while switch S2 and switch S3 are turned off. State 2: Switch S2 and switch S4 are both turned on, while switch S1 and switch S3 are turned off. State 3: Switch S2 and switch S3 are both turned on, while switch S1 and switch S4 are turned off. State 4: Switch S1 and switch S3 are both turned on, while switch S2 and switch S4 are turned off. In state one, the duty cycle is always 50% in one switching cycle. In state two and state four, the duty cycle is the same in one switching cycle and this duty cycle is called the duty cycle D. In state three, the duty cycle is 1-D in one switching cycle. The first group of switches operates in three phases: state one with a 50% duty cycle, state two with a duty cycle of D, and state three with a duty cycle of 1-D. The second set of switches operates in the following cycles: state one with a 50% duty cycle, state four with a duty cycle of D, and state three with a duty cycle of 1-D. The alternation of the first and second sets of switching states constitutes the switching state of the primary side. Based on the switching state of the primary-side switching transistor, the primary side can generate three voltage levels.

5. The hybrid modulation method for a dual active bridge converter based on stacked bridges as described in claim 1, characterized in that, The duty cycle of the secondary side switches S5~S8 is 50%, and switches S5 and S6 are complementary in conduction, and switches S7 and S8 are complementary in conduction; switches S5 and S8 are simultaneously in conduction, and switches S6 and S7 are simultaneously in conduction; the secondary side can generate a two-level square wave.

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