An isolated single-stage quasi-resonant DC-DC converter based on coupled inductors

Abstract

This invention discloses an isolated single-stage quasi-resonant DC-DC converter based on coupled inductors, comprising transformer T1 and transformer T2. The primary windings of transformers T1 and T2 are respectively connected to V via corresponding phase-shifted full-bridge structures with coupled inductors. in The positive and negative terminals of transformers T1 and T2 are connected to V via corresponding current-multiplying rectifier circuits with resonant capacitors. o The positive and negative terminals; the external inductance of the primary side of transformer T1 and the external inductance of the primary side of transformer T2 are integrated in the coupling inductor L. c In this invention, a phase-shifted full-bridge structure is used to achieve zero-voltage turn-on of the primary-side switching transistor, while the secondary-side synchronous rectifier achieves zero-current turn-off and zero-voltage turn-on through resonance. This invention achieves current sharing between modules, zero-voltage turn-on of the primary-side switching transistor, and zero-voltage turn-on and zero-current turn-off of the secondary-side synchronous rectifier, thereby improving energy conversion efficiency.

Description

Technical Field

[0001] This invention belongs to the field of converter technology, specifically relating to an isolated single-stage quasi-resonant DC-DC converter based on a coupled inductor. Background Technology

[0002] With the rapid development of artificial intelligence, big data, and cloud computing, the load demand of data centers is increasing year by year, and high-performance CPUs are developing towards lower voltage and higher current. The voltage regulator module (VRM), as a bridge between the bus voltage and the microprocessor operating voltage, has its structural performance significantly limiting energy conversion efficiency and power density. Currently, the 48V DC bus voltage is under development and has been applied in industrial fields. Many different topologies have been proposed to achieve DC-DC conversion from 48V to 1.xV for the 48V bus voltage.

[0003] Voltage regulation modules are classified into two categories based on their topology:

[0004] (1) Two-stage topology: the first stage topology converts the high input voltage to the intermediate voltage (48V to 12V), and then the second stage converts the intermediate voltage to the low output voltage (12V to 1.xV);

[0005] (2) Single-stage topology, directly converting the bus voltage into the CPU's operating voltage.

[0006] For two-stage converters, numerous isolated and non-isolated topologies have been developed and accepted by both industry and academia. However, they all suffer from significant drawbacks: firstly, the intermediate bus voltage must be carefully selected, otherwise the overall efficiency will be affected; secondly, the overall efficiency of a two-stage converter is usually limited by the product of the efficiencies of the two stages, especially under light and maximum loads, making it impossible to obtain a smooth efficiency curve. Currently, isolated single-stage topologies have attracted the attention of researchers, with the isolated single-stage quasi-resonant topology being one of the most popular and already adopted by STMicroelectronics. Due to the large output current of the voltage regulation module, multiple modules are typically connected in parallel to reduce high current stress. However, as the number of modules increases, the number of magnetic components in the converter also increases. Simultaneously, mismatched device parameters can lead to uneven current distribution between modules, causing instability and uneven heat distribution during converter operation.

[0007] Current current sharing methods are mainly divided into two categories: one is to ensure current sharing through control, and the other is a passive current sharing method. When there are multiple modules connected in parallel in the converter, the first method increases the complexity of the control system and cannot solve the problem of too many magnetic components. Summary of the Invention

[0008] The technical problem to be solved by the present invention is to provide an isolated single-stage quasi-resonant DC-DC converter based on coupled inductors to address the shortcomings of the prior art and solve the problem of uneven current flow when multiple modules are connected in parallel.

[0009] The present invention adopts the following technical solution:

[0010] An isolated single-stage quasi-resonant DC-DC converter based on coupled inductors includes transformers T1 and T2. The primary windings of transformers T1 and T2 are respectively connected to V via corresponding phase-shifted full-bridge structures with coupled inductors. in The positive and negative terminals of transformers T1 and T2 are connected to V via corresponding current multiplier rectifier circuits with resonant capacitors. o The positive and negative terminals; the external inductance of the primary side of transformer T1 and the external inductance of the primary side of transformer T2 are integrated in the coupling inductor L. c In this circuit, the phase-shifted full-bridge structure is used to achieve zero-voltage turn-on of the primary-side switching transistor, while the secondary side achieves zero-current turn-off and zero-voltage turn-on of the synchronous rectifier transistor through resonance.

[0011] Specifically, the phase-shifted full-bridge structure with coupled inductors is as follows:

[0012] One end of the primary winding of transformer T1 is connected to the coupling inductor L. c Then, the source of switching transistor Q1 and the drain of Q2 are connected respectively. The other end of the primary side of transformer T1 is connected to the source of switching transistor Q3 and the drain of Q4 respectively. The drains of switching transistors Q1 and Q3 are connected to V. in The positive terminal of the transistor, and the source terminals of switching transistors Q2 and Q4 are connected to V. in The negative terminal of the transistors is connected in parallel with the drain and source of the switching transistors Q1, Q2, Q3 and Q4, respectively, with body diode D1 and parasitic capacitance C1, body diode D2 and parasitic capacitance C2, body diode D3 and parasitic capacitance C3 and body diode D4 and parasitic capacitance C4.

[0013] One end of the primary winding of transformer T2 is connected to the coupling inductor L. c Then, connect the source of switching transistor Q5 and the drain of Q6 respectively. The other end of the primary winding of transformer T2 is connected to the source of switching transistor Q7 and the drain of Q8 respectively. The drains of switching transistors Q5 and Q7 are connected to V... in The positive terminal of the transistor, and the source terminals of switching transistors Q6 and Q8 are connected to V. in The negative terminal of the transistors is connected in parallel with the drain and source of the switching transistors Q5, Q6, Q7 and Q8, respectively, with body diode D5 and parasitic capacitance C5, body diode D6 and parasitic capacitance C6, body diode D7 and parasitic capacitance C7, and body diode D8 and parasitic capacitance C8.

[0014] Specifically, the current multiplier rectifier circuit with a resonant capacitor is as follows:

[0015] One end of the secondary side of transformer T1 is split into three paths. One path connects to one end of inductor L2 and V via inductor L1. o The positive terminal of inductor L2 is connected to the other end of the secondary winding of transformer T1; the second path is connected to the switching transistor Q. s1 The drain of the switching transistor Q s1 The source terminals are respectively connected to V o The negative terminal and the switching transistor Q s2 The source of the switch Q s2 The drain of the first circuit is connected to the other end of the secondary side of transformer T1; the third circuit is connected via capacitor C. r1 Connect the other end of the secondary side of transformer T1;

[0016] One end of the secondary side of transformer T2 splits into three paths. One path connects to one end of inductor L4 and V via inductor L3. o The positive terminal of inductor L4 is connected to the other end of the secondary winding of transformer T2; the second path is connected to the switching transistor Q. s3 The drain of the switching transistor Q s3 The source terminals are respectively connected to V o The negative terminal and the switching transistor Q s4 The source of the switch Q s4 The drain of the first circuit is connected to the other end of the secondary side of transformer T2; the third circuit is connected via capacitor C. r2 Connect the other end of the secondary side of transformer T2.

[0017] Specifically, the voltage equation for the primary side of the transformer is as follows:

[0018]

[0019] Where M is the coupling inductance L c Mutual intuition, L pk1 =L k1 +L leak1 L pk2 =L k2 +L leak2 L k1 and L k2 They are respectively the coupling inductors L c In the leakage inductance of Module 1 and Module 2, L leak1 and L leak2 The leakage inductance of the transformers in modules 1 and 2 are respectively, di pc1 For the primary current i of module 1 pc1 The differential form of di pc2 For the primary current i of module 2 pc2 The differential form of v Lc1 The voltage across the coupling inductance and transformer leakage inductance of module 1, v Lc2 This refers to the voltage across the coupling inductor and the leakage inductance of the transformer in module 2.

[0020] Specifically, the uneven current i on the primary side of the transformer pc12 for:

[0021] i pc12 =i pc12_1 +i pc12_2

[0022] Among them, i pc12_1 The primary side uneven current i pc12 The first item, i pc12_2 The primary side uneven current i pc12 The second item.

[0023] Furthermore, i pc12_1 for:

[0024]

[0025] Among them, K cs t is the flow equalization coefficient. c Let t0 be the conduction time of Q1 and Q5, and L be the conduction time of Q1 and Q5. pk1 L is the sum of the coupling inductance leakage inductance and the transformer leakage inductance of module 1. pk2 v is the sum of the coupling inductance leakage inductance and the transformer leakage inductance of module 2. Lc1 The voltage across the coupling inductance and transformer leakage inductance of module 1, v Lc2 The voltage across the coupling inductance and transformer leakage inductance of module 2, v Ln1 and v Ln2 These are the voltages across the discrete inductors and transformer leakage inductance of Module 1 and Module 2, respectively. p12_1 This is the first term of the primary-side uneven current when using discrete external inductors.

[0026] Furthermore, i pc12_2 for:

[0027] i pc12_2 =K cs i p12_2

[0028] Among them, K cs i is the flow equalization coefficient. p12_2 This is the second term of the primary-side uneven current when using discrete external inductors.

[0029] Furthermore, the flow equalization coefficient K cs for:

[0030]

[0031] Where k is the coupling coefficient and M is the coupling inductance L c Mutual intuition, L leak1 and Lleak2 These are the leakage inductances of the transformers in Module 1 and Module 2, respectively.

[0032] Furthermore, the coupling coefficient k is:

[0033] k = M / L c .

[0034] Specifically, the coupling inductor L c This is reverse coupling.

[0035] Compared with the prior art, the present invention has at least the following beneficial effects:

[0036] An isolated single-stage quasi-resonant DC-DC converter based on coupled inductors integrates the discrete external inductors of two modules into a single magnetic core. This not only achieves current sharing but also generates sufficient leakage inductance to achieve resonance, while reducing the number of magnetic components. The primary side of the transformer is a phase-shifted full-bridge structure with coupled inductors. By using coupled inductors, current sharing between modules can be achieved, and the number of external inductors is reduced from two to one. Furthermore, the phase-shifted full-bridge structure enables zero-voltage turn-on of the primary-side switching transistors. The secondary side of the transformer is a current-doubling rectifier circuit with resonant capacitors. Through resonance, zero-current turn-off and zero-voltage turn-on of the synchronous rectifier transistors can be achieved, and the current-doubling rectifier circuit can provide a large output current.

[0037] Furthermore, by adopting a phase-shifted full-bridge structure with a coupling inductor, zero-voltage conduction of the primary-side switching transistor can be achieved. The coupling inductor can not only share the current between modules, but also provide sufficient resonant inductance, while reducing the number of magnetic components.

[0038] Furthermore, by employing a current-doubling rectifier circuit with a resonant capacitor, a large output current can be provided, and the synchronous rectifier tube can achieve zero-current turn-off and zero-voltage turn-on.

[0039] Furthermore, by deriving the voltage equation of the primary side of the transformer, the relationship between the primary current and voltage, mutual inductance and leakage inductance can be derived, thereby obtaining the uneven current in the primary side.

[0040] Furthermore, the primary-side uneven current i pc12 The difference between the primary currents of module 1 and module 2 is determined by setting i. pc12 This can visually illustrate the differences in current between modules.

[0041] Furthermore, i pc12_1 The primary side uneven current i pc12 The first item is achieved by setting i pc12_1 This can represent the relationship between the first term of the primary side uneven current when using a coupled inductor and when using a discrete external inductor.

[0042] Furthermore, ipc12_2 The primary side uneven current i pc12 The second item is achieved by setting i pc12_2 This can represent the relationship between the second term of the primary side uneven current when using a coupled inductor and when using a discrete external inductor.

[0043] Furthermore, the flow equalization coefficient K cs The relationship between the coupling coefficient k, mutual inductance M, and primary-side uneven current is shown. It can be seen that when both the mutual inductance and coupling coefficient of the coupled inductors are large, K... cs The current becomes very small, thus reducing the uneven current on the primary side and achieving current sharing among modules.

[0044] Furthermore, by setting the coupling coefficient k, its relationship with the mutual inductance M and the self-inductance L can be expressed. c The relationship between them can simplify the flow equalization coefficient K. cs .

[0045] Furthermore, integrating the discrete external inductors of the two modules into a reverse-coupled inductor can achieve current sharing between the modules.

[0046] In summary, this invention can achieve current sharing and soft switching between modules, while reducing the number of magnetic components and improving energy conversion efficiency.

[0047] The technical solution of the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. Attached Figure Description

[0048] Figure 1 This is a circuit topology diagram of the present invention;

[0049] Figure 2 This is the equivalent topology diagram of the circuit of the present invention;

[0050] Figure 3 This is a schematic diagram of the primary current of module 1 of the present invention;

[0051] Figure 4 This is a schematic diagram of the current sharing error on the primary side. Detailed Implementation

[0052] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0053] In the description of this invention, it should be understood that the terms "center," "longitudinal," "lateral," "upper," "lower," "front," "rear," "left," "right," "vertical," "horizontal," "top," "bottom," "inner," "outer," "one side," "one end," and "one side," etc., indicating the orientation or positional relationship, are based on the orientation or positional relationship shown in the accompanying drawings and are only for the convenience of describing the invention and simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation, and therefore should not be construed as a limitation of the invention. Furthermore, in the description of this invention, unless otherwise stated, "a plurality of" means two or more.

[0054] In the description of this invention, it should be noted that, unless otherwise explicitly specified and limited, the terms "installation," "connection," and "linking" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art can understand the specific meaning of the above terms in this invention based on the specific circumstances.

[0055] It should be understood that, when used in this specification and the appended claims, the terms "comprising" and "including" indicate the presence of the described features, integrals, steps, operations, elements and / or components, but do not exclude the presence or addition of one or more other features, integrals, steps, operations, elements, components and / or collections thereof.

[0056] It should also be understood that the terminology used in this specification is for the purpose of describing particular embodiments only and is not intended to limit the invention. As used in this specification and the appended claims, the singular forms “a,” “an,” and “the” are intended to include the plural forms unless the context clearly indicates otherwise.

[0057] It should also be further understood that the term "and / or" as used in this specification and the appended claims refers to any combination of one or more of the associated listed items and all possible combinations, and includes such combinations.

[0058] The accompanying drawings illustrate various structural schematic diagrams according to embodiments disclosed in this invention. These drawings are not to scale, and some details have been enlarged for clarity, and some details may have been omitted. The shapes of the various regions and layers shown in the drawings, as well as their relative sizes and positional relationships, are merely exemplary and may deviate from reality due to manufacturing tolerances or technical limitations. Furthermore, those skilled in the art can design regions / layers with different shapes, sizes, and relative positions as needed.

[0059] This invention provides an isolated single-stage quasi-resonant DC-DC converter based on a coupled inductor. For a voltage regulation module (VRM) ranging from 48V to 1.xV, a coupled inductor is formed by integrating discrete external inductors. This not only achieves current sharing through coupling but also generates sufficient leakage inductance to achieve resonance. For the control signals of the primary-side switches, the primary-side switches corresponding to modules 1 and 2, such as Q1 and Q5, are controlled by the same drive signal. The external inductors of modules 1 and 2 are integrated into the coupled inductor L. c In the circuit, the primary side is a phase-shifted full-bridge structure with a coupled inductor, and the secondary side is a current-doubling rectifier circuit with a resonant capacitor. The phase-shifted full-bridge structure on the primary side enables zero-voltage turn-on of the primary-side switching transistor, while the secondary side achieves zero-current turn-off and zero-voltage turn-on of the synchronous rectifier transistor through resonance, thereby further reducing the converter's losses.

[0060] Please see Figure 1 The present invention discloses an isolated single-stage quasi-resonant DC-DC converter based on coupled inductors, comprising module 1 and module 2. Module 1 converts a 48V input voltage to a 1.xV output voltage and provides half of the output current. Module 2 converts a 48V input voltage to a 1.xV output voltage and provides half of the output current.

[0061] Module 1 includes a transformer T1, one end of which is connected to a coupling inductor L. c Then, the source of switching transistor Q1 and the drain of Q2 are connected respectively. The other end of the primary side of transformer T1 is connected to the source of switching transistor Q3 and the drain of Q4 respectively. The drains of switching transistors Q1 and Q3 are connected to V. in The positive terminal of the transistor, and the source terminals of switching transistors Q2 and Q4 are connected to V. in The negative terminal of the transistors is connected in parallel with the drain and source of the switching transistors Q1, Q2, Q3 and Q4, respectively, with body diode D1 and parasitic capacitance C1, body diode D2 and parasitic capacitance C2, body diode D3 and parasitic capacitance C3 and body diode D4 and parasitic capacitance C4.

[0062] One end of the secondary side of transformer T1 is split into three paths. One path connects to one end of inductor L2 and V via inductor L1. o The positive terminal of inductor L2 is connected to the other end of the secondary winding of transformer T1; the second path is connected to the switching transistor Q. s1 The drain of the switching transistor Q s1 The source terminals are respectively connected to V o The negative terminal and the switching transistor Q s2 The source of the switch Q s2 The drain of the first circuit is connected to the other end of the secondary side of transformer T1; the third circuit is connected via capacitor C. r1 Connect the other end of the secondary side of transformer T1.

[0063] Module 2 includes transformer T2, one end of the primary winding of transformer T2 is coupled to inductor L. c Then, connect the source of switching transistor Q5 and the drain of Q6 respectively. The other end of the primary winding of transformer T2 is connected to the source of switching transistor Q7 and the drain of Q8 respectively. The drains of switching transistors Q5 and Q7 are connected to V... in The positive terminal of the transistor, and the source terminals of switching transistors Q6 and Q8 are connected to V. in The negative terminal of the transistors is connected in parallel with the drain and source of the switching transistors Q5, Q6, Q7 and Q8, respectively, with body diode D5 and parasitic capacitance C5, body diode D6 and parasitic capacitance C6, body diode D7 and parasitic capacitance C7 and body diode D8 and parasitic capacitance C8.

[0064] One end of the secondary side of transformer T2 splits into three paths. One path connects to one end of inductor L4 and V via inductor L3. o The positive terminal of inductor L4 is connected to the other end of the secondary winding of transformer T2; the second path is connected to the switching transistor Q. s3 The drain of the switching transistor Q s3 The source terminals are respectively connected to V o The negative terminal and the switching transistor Q s4 The source of the switch Q s4 The drain of the first circuit is connected to the other end of the secondary side of transformer T2; the third circuit is connected via capacitor C. r2 Connect the other end of the secondary side of transformer T2.

[0065] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. The components of the embodiments of the present invention described and shown in the accompanying drawings can generally be arranged and designed in various different configurations. Therefore, the following detailed description of the embodiments of the present invention provided in the accompanying drawings is not intended to limit the scope of the claimed invention, but merely to illustrate selected embodiments of the invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without inventive effort are within the scope of protection of the present invention.

[0066] This invention integrates discrete external inductors in a module to form a coupled inductor. This not only achieves current sharing through coupling but also generates sufficient leakage inductance to achieve resonance, acting as a resonant inductor. The coupled inductor is reverse-coupled, meaning it is driven by the primary current i. pc1 and i pc2 The direction of the generated magnetic flux is opposite.

[0067] The flow equalization principle of the present invention will be described in detail below with reference to the accompanying drawings.

[0068] Please see Figure 2 This is the equivalent topology of the isolated single-stage quasi-resonant converter based on coupled inductors according to the present invention. In the figure, v A1B1 and v A2B2 These are the midpoint voltages of the primary-side full-bridge circuits of modules 1 and 2, respectively. cr1_c and v cr2_c They are C r1 and C r2 The voltage across the terminals, where n is the transformer turns ratio, and L leak1 and L leak2 These are the leakage inductances of the transformers in Module 1 and Module 2, respectively.

[0069] Coupled inductor L c The leakage inductance of module 1 and module 2 are respectively L k1 and L k2 I feel like L c1 and L c2 Mutual inductance is defined as M; L is defined as... pk1 =L k1 +L leak1 L pk2 =L k2 +L leak2 .

[0070] Combining Kirchhoff's voltage law, the voltage equation for the primary side of the transformer is obtained as follows:

[0071]

[0072] Solve for (1) and combine with Figure 3 The primary-side uneven current i of the isolated single-stage quasi-resonant converter based on coupled inductor is obtained. pc12 for:

[0073] i pc12 =i pc1 (t c )-i pc2 (t c ) = i pc12_1 +i pc12_2 (2)

[0074]

[0075]

[0076] With i pc12 The process is similar, yielding the primary-side uneven current i of the isolated single-stage quasi-resonant converter based on discrete inductors. p12 i p12 For i p12_1 and i p12_2 The sum of i pc12and i p12 The relationship is represented as:

[0077]

[0078] i pc12_2 =K cs i p12_2 (6)

[0079] Among them, v Ln1 and v Ln2 These are the voltages across the discrete inductors and transformer leakage inductance of Module 1 and Module 2, respectively.

[0080]

[0081] Flow coefficient K cs The difference in primary current should be minimized to achieve current sharing.

[0082] When the self-inductance values ​​of the coupled inductors are the same, i.e., L c1 =L c2 =L c The equation for the coupling coefficient is written as: k = M / L c At this point, the leakage inductance of the coupled inductor becomes L. k1 =L k2 =L k .

[0083] Let L c1 =L c2 K cs Rewrite it in the following form:

[0084]

[0085] According to the above formula, when both the mutual inductance and coupling coefficient of the coupled inductors are large, K cs The current becomes very small, thus reducing the uneven current on the primary side and achieving current sharing among modules.

[0086] To verify the previous analysis, an isolated single-stage quasi-resonant converter based on coupled inductors was built in Psim. Module 1 is the reference design, and the parameter error between Module 2 and Module 1 is 10%. Figure 4 The expression shows the trend of primary-side current sharing error as a function of coupling coefficient k in an isolated single-stage quasi-resonant converter based on coupled inductors. The primary-side current sharing error δi p Defined as:

[0087]

[0088] Please see Figure 4The coupling coefficient k changes from 0 to 0.97, the mutual inductance M changes from 0 to 40μH, and the leakage inductance L of the coupled inductor... k1 and L k2 The values ​​are 1μH and 1.1μH, respectively. When the isolated single-stage quasi-resonant converter uses discrete inductors, i.e., the coupling coefficient is zero, the primary current sharing error is 44.73%. As the mutual inductance M and the coupling coefficient k increase, the primary current sharing error gradually approaches zero.

[0089] In summary, the present invention provides an isolated single-stage quasi-resonant DC-DC converter based on coupled inductors, which can achieve current sharing between modules, provide high output current, reduce the high current stress of the secondary synchronous rectifier, and realize zero-voltage turn-on of the primary-side switch, zero-voltage turn-on and zero-current turn-off of the secondary synchronous rectifier, thereby improving energy conversion efficiency.

[0090] The above content is only for illustrating the technical concept of the present invention and should not be construed as limiting the scope of protection of the present invention. Any modifications made to the technical solution based on the technical concept proposed in this invention shall fall within the scope of protection of the claims of this invention.

Claims

1. An isolated single-stage quasi-resonant DC-DC converter based on a coupled inductor, characterized in that, Including transformers T 1 and transformer T 2. Transformer T 1 and transformer T The primary sides of 2 are respectively connected via corresponding phase-shifted full-bridge structures with coupled inductors. V in The positive and negative terminals of the transformer. T 1 and transformer T The secondary side of 2 is connected to the corresponding current multiplier rectifier circuit with resonant capacitor. V o The positive and negative terminals; transformer T 1. External inductance and transformer on the primary side T 2. The external inductor on the primary side is integrated into the coupled inductor. L c In the middle, the phase-shifted full-bridge structure is used to achieve zero-voltage turn-on of the primary-side switching transistor, and the secondary side achieves zero-current turn-off and zero-voltage turn-on of the synchronous rectifier transistor through resonance; The phase-shifted full-bridge structure with coupled inductors is specifically as follows: transformer T One end of the primary side of 1 is coupled to an inductor. L c Then connect the switching transistors respectively. Q 1 source and Q 2's drain, transformer T The other end of the primary side of 1 is connected to the switching transistor. Q 3 source poles and Q 4 drain, switching transistor Q 1 and Q 3 drain connection V in The positive terminal of the switching transistor Q 2 and Q 4 source connections V in The negative terminal of the switching transistor Q 1. Q 2. Q 3 and Q A body diode is connected in parallel between the drain and source of 4. D 1 and parasitic capacitance C 1. Body diode D 2 and parasitic capacitance C 2. Body diode D 3 and parasitic capacitance C 3 and body diode D 4 and parasitic capacitance C 4; transformer T One end of the primary side of 2 is coupled to an inductor. L c Then connect the switching transistors respectively. Q 5 source poles and Q 6 drain, transformer T The other end of the primary side of 2 is connected to the switching transistor. Q 7 source poles and Q 8 drain, switching transistor Q 5 and Q 7 drain connection V in The positive terminal of the switching transistor Q 6 and Q 8 source connections V in The negative terminal of the switching transistor Q 5. Q 6. Q 7 and Q A body diode is connected in parallel between the drain and source of the 8-pin diode. D 5 and parasitic capacitance C 5. Body diode D 6 and parasitic capacitance C 6. Body diode D 7 and parasitic capacitance C 7 and body diode D 8 and parasitic capacitance C 8; The current multiplier rectifier circuit with resonant capacitor is as follows: transformer T One end of the secondary side splits into three paths, one of which goes through an inductor. L 1 Connect the inductors respectively L one end of 2 and V o The positive terminal of the inductor L The other end of 2 is connected to the transformer. T The other end of the first side; the second circuit connects to the switch transistor. Q s1 The drain of the switching transistor Q s1 The source terminals are respectively connected V o negative terminal and switching transistor Q s2 The source, the switch Q s2 Drain connection transformer T The other end of the first side; the third path via capacitor C r1 Connecting transformer T The other end of the second side; transformer T One end of the two-phase circuit splits into three paths, one of which goes through an inductor. L 3 Connect the inductors respectively L one end of 4 and V o The positive terminal of the inductor L The other end of 4 is connected to the transformer. T The other end of the second side; the second circuit connects to the switch tube. Q s3 The drain of the switching transistor Q s3 The source terminals are respectively connected V o negative terminal and switching transistor Q s4 The source, the switch Q s4 Drain connection transformer T The other end of the second side; the third path via capacitor C r2 Connecting transformer T The other end of the second side.

2. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 1, characterized in that, The voltage equation for the primary side of the transformer is as follows: in, For coupled inductors L c Mutual intuition L pk1 = L k1 + L leak1 , L pk2 = L k2 + L leak2 , L k1 and L k2 They are coupled inductors L c Leakage inductance in modules 1 and 2 L leak1 and L leak2 The leakage inductance of the transformers in Module 1 and Module 2 are respectively. The primary current of module 1 i pc1 The differential form, For the primary current of module 2 i pc2 The differential form, The voltage across the coupling inductor and the leakage inductance of the transformer in module 1. This refers to the voltage across the coupling inductor and the leakage inductance of the transformer in module 2.

3. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 1, characterized in that, Uneven current on the primary side of the transformer i pc12 for: in, Non-uniform current on the primary side i pc12 The first item, Non-uniform current on the primary side i pc12 The second item.

4. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 3, characterized in that, for: in, The flow equalization coefficient is... For any time, for Q 1 and Q 5's conduction time, This is the sum of the coupling inductance leakage inductance of module 1 and the transformer leakage inductance. This is the sum of the coupling inductance leakage inductance of module 2 and the transformer leakage inductance. The voltage across the coupling inductor and the leakage inductance of the transformer in module 1. The voltage across the coupling inductor and the leakage inductance of the transformer in module 2. v Ln1 and v Ln2 These are the voltages across the discrete inductors and transformer leakage inductance of Module 1 and Module 2, respectively. This is the first term of the primary-side uneven current when using discrete external inductors.

5. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 3, characterized in that, for: in, The flow equalization coefficient is... This is the second term of the primary-side uneven current when using discrete external inductors.

6. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 4 or 5, characterized in that, Flow coefficient K cs for: in, The coupling coefficient is... For coupled inductors L c Mutual intuition L leak1 and L leak2 These are the leakage inductances of the transformers in Module 1 and Module 2, respectively.

7. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to claim 6, characterized in that, Coupling coefficient for: k = M / L c 。 8. The isolated single-stage quasi-resonant DC-DC converter based on coupled inductors according to any one of claims 1 to 7, characterized in that, Coupled inductor L c This is reverse coupling.

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