Isolated current control method for salient pole multiphase permanent magnet synchronous motor

Abstract

The application relates to a method for controlling the current of a salient pole multiphase permanent magnet synchronous motor, and belongs to the technical field of motor control. The method is used for solving the problems of poor control performance at medium and high speeds and the unfixed switching frequency of H-bridge driving hysteresis control in the prior art of multiphase permanent magnet synchronous motor. The method is characterized in that: the multiphase permanent magnet synchronous motor is driven by an H-bridge for each phase winding, and the current of each phase winding is separately controlled by a current loop controller; in scheme one, the current loop controller is a PIR controller which is composed of a PI controller, a fundamental wave resonance controller and a third harmonic resonance controller; in scheme two, the current loop controller is a PIR controller+active disturbance rejection controller in parallel connection, and the active disturbance rejection controller is a 2-order extended state observer.

Description

Technical Field

[0001] This invention relates to H-bridge isolation current control technology for salient-pole multiphase permanent magnet synchronous motors, and belongs to the field of motor control technology. Background Technology

[0002] Multiphase permanent magnet synchronous motors (PMSMs) possess numerous advantages, including high reliability, strong fault tolerance, and excellent torque characteristics, making them promising candidates for applications in key fields such as electric vehicles, aerospace, and deep-sea exploration. Currently, multiphase motor systems often employ half-bridge drive with non-isolated current control, H-bridge drive with PI control, or H-bridge drive with hysteresis control to manage the current during normal operation and fault tolerance. (See also...) Figure 2 When using half-bridge drive and non-isolated current control, the motor windings are star-connected, requiring corresponding multi-phase coordinate transformations. Switching a fault-tolerant motor from normal to fault-tolerant operation necessitates changes to both the coordinate transformation and control method, which is cumbersome. Furthermore, this winding connection method weakens the motor's fault tolerance. H-bridge drive can achieve more types of fault tolerance for motors, and switching between normal and fault-tolerant conditions only requires changing the current setpoint, improving the fault tolerance of multi-phase motor systems. However, PI control exhibits poor performance at medium to high speeds, and hysteresis control results in an unstable switching frequency and reliance on wide-bandgap devices. Therefore, it is necessary to research and design an isolated current control method based on H-bridge drive with a fixed switching frequency, suitable for medium to high-speed operation of fault-tolerant motors, to improve their performance. Summary of the Invention

[0003] To address the issues of poor control performance at medium and high speeds when using H-bridge driven PI control for existing multiphase permanent magnet synchronous motors, and the inconsistent switching frequency when using H-bridge driven hysteresis control, this invention provides an isolated current control method suitable for salient-pole multiphase permanent magnet synchronous motors.

[0004] The present invention discloses an isolated current control method applicable to salient-pole multiphase permanent magnet synchronous motors, the method comprising:

[0005] The multiphase permanent magnet synchronous motor uses an H-bridge to drive each phase winding. The current of each phase winding is individually controlled by a current loop controller. The current loop controller includes a PIR controller, which is composed of a PI controller, a fundamental resonant controller, and a third harmonic resonant controller. The transfer function of the PIR controller is:

[0006]

[0007] Where K p K is the proportionality coefficient. i K is the integral coefficient. r K is the resonance coefficient. r1 and K r3Let be the fundamental frequency and the third harmonic resonance coefficient, respectively; s denotes the Laplace operator; ω0 denotes the resonant frequency; ω c ω represents the frequency selectivity bandwidth of the resonant controller. c1 and ω c1 These represent the frequency selection bandwidths of the fundamental and third harmonic resonant controllers, respectively.

[0008] Preferably, the current loop controller further includes an active disturbance rejection controller (PIR controller), and the PIR controller and the PIR controller are connected in parallel to form a joint control law, with the PIR controller outputting a value u_ PIR With the output value u_ of the active disturbance rejection controller ESO The sum, minus the interference term E, is used to generate the current for the control winding.

[0009] Preferably, the active disturbance rejection controller employs a second-order extended state observer, expressed as:

[0010]

[0011] In the formula, b0(t) is the known output gain, b0(t) = 1 / (L m0 +L m2 cos(2θ)), θ is the electric angle of the motor, L m0 L represents the constant component of the inductance matrix. m2 This represents the amplitude of the second harmonic component of the inductance matrix;

[0012] β1 and β2 are the first-order and second-order error coefficients, respectively, where β1 = 2α0 and β2 = α0. 2 α0 is the eigenvalue of the extended state observer;

[0013] e1 Observation error; i s Let be the current of the s-th phase winding of the m-phase motor, where s = 1, 2, ..., m;

[0014] z1 is the observed value of the phase current, and z2 is the observed value of the first derivative of the phase current. These are the observed values ​​of phase current disturbance. The observed value of the first derivative of the phase current disturbance;

[0015] u _ESO This is the output value of the active disturbance rejection controller;

[0016] Connecting the PIR controller and the active disturbance rejection controller in parallel yields the following joint control law:

[0017]

[0018] In the formula, u0 represents the current deviation compensation. k pz The proportional control law of the active disturbance rejection controller. Given the phase current value, For current disturbance compensation. It is the first derivative of the phase current setpoint, used to compensate for disturbances caused by changes in the current setpoint.

[0019] Preferably, the relationship between the voltage and current of each phase winding of the multiphase motor in the complex frequency domain can be expressed as follows:

[0020]

[0021] In the formula u s (s) represents the phase voltage in a salient-pole permanent magnet synchronous motor. E(s) is the back electromotive force of the motor, including the back electromotive force of the permanent magnet and the back electromotive force of the electric excitation.

[0022] M0 represents the constant component of the reciprocal of the inductance, and M2 represents the second harmonic coefficient of the reciprocal of the inductance.

[0023] R s Represents the resistance matrix, ω e This indicates the electrical frequency of the motor.

[0024] Preferably, the motor inductance meets the following conditions:

[0025]

[0026] Preferably, it also includes a speed loop, which employs proportional-integral control.

[0027] The beneficial effects of this invention are as follows: The isolated current control method for a multiphase fault-tolerant motor system disclosed in this invention establishes a mathematical model of the multiphase permanent magnet synchronous motor in a stationary coordinate system. Two schemes are employed for the current loop controller: one uses a proportional-integral-resonant controller (PIR controller) as the current loop controller to control the phase current of the motor under normal operating conditions and phase-open-circuit fault-tolerant operating conditions; the other uses a PIR controller and an active disturbance rejection controller connected in parallel as the current loop controller to achieve tracking of the multiphase motor phase current to the sinusoidal current input and improve the anti-disturbance capability of the motor current loop. This enables the fault-tolerant motor system to drive the motor at medium and high speeds under both normal operating conditions and phase-open-circuit fault-tolerant conditions at a fixed switching frequency, further realizing the control isolation of the fault-tolerant motor and fully utilizing the fault-tolerant performance of the fault-tolerant motor system. Attached Figure Description

[0028] Figure 1 This is a schematic diagram of the H-bridge driving method using the method of the present invention;

[0029] Figure 2 This is a schematic diagram of the half-bridge drive method mentioned in the background technology;

[0030] Figure 3 This is a block diagram of the motor winding transfer function;

[0031] Figure 4 This is a structural block diagram driven by the method of the present invention;

[0032] Figure 5 This is a block diagram of the current loop controller of the present invention using a PIR controller;

[0033] Figure 6 This is a block diagram of the setpoint and disturbance transfer function of the current loop controller using a PIR controller, where... Figure 6 (a) Given the winding current i s ref The transfer function block diagram of the back EMF disturbance E. Figure 6 (b) is a block diagram of the transfer function of the disturbance generated by the modulation characteristics of the salient pole motor;

[0034] Figure 7 This is the Bode plot of a current loop controller using a PIR controller, where... Figure 7 (a) ω c Fixed, K r change, Figure 7 (b) K r Fixed, ω c change;

[0035] Figure 8 This is the Bode plot of the open-loop transfer function of the current under the action of the PIR controller, where... Figure 8 (a) PI parameters and ω c Fixed, K r change, Figure 8 (b) PI parameter and K r Fixed, ω c change, Figure 8 (c) ω c and K r Fixed, PI parameter varies;

[0036] Figure 9 It is the Bode plot of the current closed-loop transfer function under the action of PI controller and PIR controller;

[0037] Figure 10 The diagram shows the Bode plot of the disturbance closed-loop transfer function under the action of the PIR controller. Figure 10 (a) is the Bode plot of the motor's transfer function to back EMF disturbance. Figure 10 (b) is the Bode plot of the disturbance transfer function caused by the modulation characteristics of the salient pole motor;

[0038] Figure 11The block diagram shows the structure of the current loop controller, which uses a PIR controller and an active disturbance rejection controller for joint control.

[0039] Figure 12 The simulation results of phase current and waveforms of current setpoints for different control methods are shown in the example under the conditions of a speed of 3000 rpm and a load of 6 N·m.

[0040] Figure 13 The following are simulation results for an example of accelerating from 0 rpm to 3000 rpm under a given speed ramp under a load of 2 N·m. Figure 13 (a) is a simulation waveform of torque and speed. Figure 13 (b) is a simulation waveform of the phase current. Detailed Implementation

[0041] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0042] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0043] Currently, multiphase motor systems operating in the H-bridge-open winding mode exhibit greater fault tolerance under numerous fault-tolerant conditions, even enabling three-phase motors to possess a degree of fault tolerance. However, this mode primarily employs hysteresis control, which, without wide-bandgap devices, suffers from drawbacks such as low control bandwidth, inconsistent switching frequency, and low bus voltage utilization. Therefore, it is necessary to research high-bandwidth, fixed-frequency current control methods based on H-bridge-open windings to improve motor fault tolerance performance.

[0044] The present invention will be further described below with reference to the accompanying drawings and specific embodiments, but this is not intended to limit the scope of the invention.

[0045] Specific Implementation Method 1: The following is combined with... Figures 1 to 12 This embodiment describes an isolated current control method suitable for salient-pole multiphase permanent magnet synchronous motors, employing... Figure 1 The block diagram of the isolated current control method for the H-bridge driven m-phase motor with individual control of each phase winding is shown below. Figure 4 As shown, the current setpoint and feedback real value of each phase winding are processed by the current loop controller of that phase and then sent to the SPWM to generate a control signal for the inverter to control the motor winding current. This embodiment provides two current loop controller schemes:

[0046] Option 1: Use a PIR controller, see [link / reference] Figure 5 As shown.

[0047] The transfer function of the PIR controller is:

[0048]

[0049] Where K p K is the proportionality coefficient. i K is the integral coefficient. r K is the resonance coefficient. r1 and K r3 Let be the fundamental frequency and the third harmonic resonance coefficient, respectively; s denotes the Laplace operator; ω0 denotes the resonant frequency; ω c ω represents the frequency selectivity bandwidth of the resonant controller. c1 and ω c1 These represent the frequency selection bandwidths of the fundamental and third harmonic resonant controllers, respectively.

[0050] Option 2: Use a PIR controller and an active disturbance rejection controller in parallel. (See [reference]) Figure 11 As shown.

[0051]

[0052] In the formula, u0 represents the current deviation compensation. k pz The proportional control law of the active disturbance rejection controller. Given the phase current value, For current disturbance compensation. It is the first derivative of the phase current setpoint, used to compensate for disturbances caused by changes in the current setpoint.

[0053] The active disturbance rejection controller uses a second-order extended state observer, denoted as:

[0054]

[0055] In the formula, b0(t) is the known output gain, b0(t) = 1 / (L m0 +L m2 cos(2θ)), θ is the electric angle of the motor, L m0 L represents the constant component of the inductance matrix. m2 This represents the amplitude of the second harmonic component of the inductance matrix;

[0056] β1 and β2 are the first-order and second-order error coefficients, respectively, where β1 = 2α0 and β2 = α0. 2 α0 is the eigenvalue of the extended state observer;

[0057] e1 Observation error; is Let be the current of the s-th phase winding of the m-phase motor, where s = 1, 2, ..., m;

[0058] z1 is the observed value of the phase current, and z2 is the observed value of the first derivative of the phase current. These are the observed values ​​of phase current disturbance. The observed value of the first derivative of the phase current disturbance;

[0059] u _ESO This is the output value of the active disturbance rejection controller;

[0060] The above two schemes are applicable to the normal and fault-tolerant operating conditions of salient-pole multiphase permanent magnet synchronous motors that meet the following conditions:

[0061]

[0062] L m0 L represents the constant component of the inductance matrix. m2 This represents the amplitude of the second harmonic component of the inductance matrix.

[0063] Working principle:

[0064] First, the application of the control method proposed in this invention, the motor theory, and the related mathematical model are explained. This invention is applied to the H-bridge drive control system of a multiphase fault-tolerant motor, and its winding connection method is as follows: Figure 1 As shown. Based on Figure 1 The current control method for the winding connection shown allows for independent control of each phase winding, further achieving control isolation on top of physical, magnetic, and thermal isolation. This is similar to a half-bridge drive (such as...). Figure 2 Compared to non-isolated current control methods, the full-bridge drive isolated current control method offers more fault tolerance types and stronger fault tolerance capabilities, while maintaining the same current control method for both normal and fault-tolerant motor operation. However, traditional fault-tolerant motor full-bridge drive systems employ hysteresis control, resulting in inconsistent switching frequencies of the switching transistors and poor control performance at medium and high speeds. To address these issues, this invention proposes a current control method based on a proportional-integral-resonant and active disturbance rejection controller, which fully leverages the fault-tolerant potential of multiphase motor systems while achieving current control for multiphase fault-tolerant motors.

[0065] The current forms of an m-phase permanent magnet synchronous motor under normal and open-circuit fault-tolerant conditions are as follows:

[0066]

[0067] The formula gives two ways to express the motor operating current: ① Amplitude (I i ) + phase angle (θ) i ); ② Cosine component (I id )+sine component(I iq), i = 1 2…m. This way of expressing the current can represent the current given when the motor is working normally, or the current given under certain fault-tolerant conditions.

[0068] Factors such as the neutral point of the motor and the given dq-axis current will impose the following constraints on the motor's operating current:

[0069]

[0070] In the formula, the first h equations represent open-circuit constraints, the middle k equations with a value of 0 represent constraints imposed by the k neutral points, and the last two equations represent dq current constraints. Other constraints may arise if other performance characteristics are considered; this invention primarily involves neutral point constraints and dq current constraints. Each phase current is determined by two variables: the current amplitude and the phase angle, or by the sine and cosine components. Therefore, the current of an m-phase motor is determined by 2m variables. Each equation in the constraints can constrain two variables. Therefore, when the number of equations in the constraints is less than or equal to the number of motor phases, the motor current can satisfy the constraints; otherwise, the motor current cannot satisfy all constraints.

[0071] Taking a 3-phase permanent magnet synchronous motor as an example, a typical 3-phase motor has a neutral point star connection and its current constraints are as follows:

[0072]

[0073] When providing fault tolerance for an open-circuit phase in a three-phase motor, an additional constraint of zero current in one phase is added. This results in four current constraints, more than the number of phases. Therefore, a three-phase motor using a half-bridge driven, non-isolated current control method lacks fault tolerance. However, if an H-bridge driven, isolated current control method is used, the motor current has one less neutral point constraint, thus enabling fault tolerance for an open-circuit phase.

[0074] When using the isolated current control method, the normal operating current is given by equation (1). The relevant parameters in equation (1) need to satisfy the constraints shown in equation (2). Since the winding connection of the drive mode of the isolated current control method has no neutral point, only the dq current constraint needs to be satisfied. The current given under open-circuit fault tolerance is also given by equation (1), and the constraints shown in equation (2) need to be satisfied, including the dq current constraint and the open-circuit constraint.

[0075] The mathematical model of the permanent magnet synchronous motor used in this embodiment is as follows:

[0076]

[0077] In the formula U s R represents the phase voltage vector. s Represents the resistance matrix, Ψ sRepresents the stator flux linkage vector, which consists of the permanent magnet flux linkage Ψ. f and stator current flux L s I s Together, L s I represents the stator inductance matrix. s This represents the stator current vector, where p is the number of pole pairs of the motor, and ω is the stator current vector. m θ represents the mechanical speed of the motor, and θ represents the electrical angle of the motor.

[0078] In a stationary coordinate system, the stator inductance matrix of the salient-pole motor is L. s =L0+L2, where L0 represents the DC part of the stator inductance matrix and L2 represents the second harmonic part of the stator inductance matrix;

[0079]

[0080]

[0081] In the formula L m0 L represents the constant component of the inductance matrix. m2 The amplitude of the second harmonic component of the inductance matrix is ​​represented by θ, which is the electrical angle of the motor and the mechanical angle of the motor. m There exists θ = pθ m The relationship, α i (i=1 2...m) is the spatial position angle of the motor winding. For 3-phase motors (α1=0, α2=-2π / 3, α3=2π / 3), for 5-phase motors (α1=0, α2=-2π / 5, α3=-4π / 5, α4=-6π / 5, α5=-8π / 5), for 6-phase motor (α1=0, α2=-2π / 3, α3=2π / 3, α4=-π, α5=-π / 3, α6=π / 3) or (α1=0, α2=-2π / 3, α3=2π / 3, α4=-π / 6, α5=-5π / 6, α6=π / 2).

[0082] In a salient-pole motor, the inductance matrix contains a second harmonic matrix L. m2 The voltage equation for a single phase winding can be written in the form shown below:

[0083]

[0084] Where E(i1...i i-1 i i+1 ...i m ,t,ψ f E(i1...i) represents the back electromotive force generated by other windings and rotor permanent magnets on the i-th phase winding to be controlled. In the control loop, E(i1...i) represents the back electromotive force generated by other windings and rotor permanent magnets on the i-th phase winding to be controlled. i-1 i i+1 ...i m,t,ψ f This is considered interference.

[0085] When the phase current of the motor is working normally tracks the given value, the current is a sine wave with one electric frequency as shown in equation (1). The combined back EMF generated by the current and the permanent magnet is generally dominated by the fundamental back EMF and the third harmonic back EMF.

[0086] To facilitate subsequent analysis of the salient-pole motor current controller performance, it is necessary to obtain the transfer function of the salient-pole motor winding. However, due to the presence of the second harmonic of the inductor, the transfer function of the winding cannot be directly solved. To facilitate subsequent control and analysis, a certain simplification analysis method is adopted. Equation (7) is rewritten as shown in Equation (8).

[0087]

[0088] The reciprocal of an inductance containing second harmonics is difficult to analyze and process. Therefore, a Fourier series expansion is performed on the reciprocal of the inductance.

[0089]

[0090] In the formula, M0 represents the constant component of the reciprocal of the inductance, M ai M represents the cosine component coefficient of the inductor's reciprocal i-th harmonic. bi This represents the sinusoidal component coefficient of the i-th reciprocal harmonic of the inductor.

[0091] When the salient pole motor satisfies Under certain conditions, the influence of higher harmonics can be ignored, and the Fourier series of the inverse of the inductance can be retained up to the second harmonic, as shown in equation (10).

[0092]

[0093] In equation (10), M2 represents the second harmonic coefficient of the inductance. Substituting the result of the Fourier series expansion into equation (8) yields the result shown in equation (11).

[0094]

[0095] α in equation (11) m Let α represent the spatial position angle (electric angle) of the stator winding. All AC quantities in equation (11) can be expanded into linear combinations of sin(2θ), cos(2θ) and sin(4θ). For ease of drawing and writing after the Laplace transformation, let α be used. m Taking α = 0 as an example for analysis, the rest are analyzed similarly. m Taking θ = 0 as an example, let θ = ω eThe Laplace transform result of equation (11) is shown in equation (12).

[0096]

[0097] The mathematical model of the salient pole motor expressed by equation (12) is drawn as a flowchart as follows: Figure 3 As shown in the figure, the stator winding of the salient-pole motor is a first-order system. However, due to the magnetic field modulation effect of the motor, the alternation of the inductance causes the system output to produce additional frequencies compared to the input, in addition to changes in amplitude and phase. For ease of analysis and explanation, the characteristic of changing amplitude and phase is called the first-order system characteristic, and the performance of changing frequency is called the modulation characteristic.

[0098] Equation (12) and Figure 3 The processing method in this paper separates the first-order system characteristics and modulation characteristics of the salient-pole motor. The first-order system characteristics are reflected in the transfer function M0 / (1+R) s In M0), the modulation characteristics are reflected in the frequency offset term after the Laplace transform. After this separation, the transfer function M0 / (1+R) can be used. s The M0 section is designed as a feedback controller for the forward channel. At the same time, the frequency offset term is used as a disturbance to verify the control system's ability to suppress disturbances. This completes the design and performance analysis of the single winding current control system for the salient pole motor.

[0099] This invention proposes an isolated current control method, such as... Figure 4 As shown, the current loop controller in this method adopts two schemes. Scheme 2 is a joint controller, which is composed of a proportional-integral-resonant controller (PIR controller) and an active disturbance rejection controller.

[0100] The proportional-integral-resonant controller (PIR controller) is used to control the current in the motor windings. The PIR controller consists of a PI controller, a fundamental resonant controller, and a third harmonic resonant controller, and its expression is as follows:

[0101]

[0102] The controller block diagram is as follows Figure 5 As shown. Due to the modulation characteristics of salient-pole motors, the fundamental voltage and current of the motor will also generate a large number of third harmonic disturbances during operation. Therefore, the resonant controller needs to adopt a structure in which the fundamental resonant controller and the third harmonic resonant controller are connected in parallel.

[0103] When analyzing a salient-pole motor load, it is necessary to consider the first-order system characteristics and modulation characteristics of the salient-pole motor load. The first-order system characteristics characterize the relationship between the system output and the input at the same frequency, which is reflected in the control loop. The block diagram with the input at the same frequency is shown below. Figure 6As shown in (a); inputs of other frequencies generated by the modulation characteristics in the system manifest as disturbances in the system. The block diagram using inputs of other frequencies generated by the modulation characteristics as inputs is shown below. Figure 6 As shown in (b).

[0104] Further explanation of the impact of each parameter on the performance of the PIR controller. An example is given: the inductance L of a salient-pole motor. m0 =0.45mH, L m2 =0.13mH, which satisfies The condition, phase resistance R s =0.0169Ω, so M0 = 2321H can be calculated. -1 M2 = -685H -1 The adjusted controller parameters are as follows: K p =2000 / M0, K i =2000R s K r1 =20, ω c1 =20, ω0=250π, K r3 =40, ω c3 =30, this set of parameters is referred to as the given parameters below. Based on the given parameters, the resonance coefficient K is adjusted individually. r With filter bandwidth ω c The Bode diagram of the post-PIR controller is as follows: Figure 7 As shown, K can be seen r K has a significant impact on the amplitude gain and phase change of the signal near the resonant frequency. r The larger the value of ω, the greater the gain at the resonant frequency, and the larger the range of phase change near the resonant frequency; c It has no effect on the gain and phase variation range at the resonant frequency, but it will affect the frequency selectivity bandwidth of the PIR controller, ω. c The larger the value, the larger the frequency selectivity bandwidth near the resonant frequency ω0, and the slower the phase change rate near the resonant frequency.

[0105] Analyzing system stability requires starting with the first-order system characteristics of a salient-pole motor. Based on... Figure 6 (a) and (13) further adjust K individually based on the given parameters. r ω c Using PI parameters, plot the open-loop Bode plot of the current control system under the action of the PIR controller, as shown below. Figure 8 As shown. The system stability can be analyzed from the Bode plot of the open-loop transfer function, and the influence of each parameter on the system stability can be further analyzed. The stability of the control system requires that the phase margin of the Bode plot of the open-loop transfer function at the crossover frequency meets the requirement of 60°. (Summary) Figure 8 (a)(b)(c). It can be seen that the system stability is determined by K. rParameters, ω c The resonance coefficient K is determined jointly by the parameters and the PI parameter. r The larger the value, the greater the open-loop gain at the harmonic frequency; however, the range of phase angle variation near the cutoff frequency also increases, and the phase margin decreases; the frequency selectivity bandwidth ω c Increasing the gain of the PIR controller near the resonant frequency will increase the open-loop gain of the PIR controller, enhance the harmonic suppression bandwidth of the PIR controller, and also slow down the rate of phase change near the resonant frequency, reducing the phase margin at the cutoff frequency, which is detrimental to the stability of the system. Figure 8 (c) In this case, PI parameter 2 is a given parameter, and the K value of PI parameter 1 is... p1 =5K p2 K i1 =5K i2 K of PI parameter 3 p3 =5K p2 K i3 =5K i2 PI parameter K p K i Increasing the frequency can change the crossover frequency, which can adjust the phase margin of the system to some extent.

[0106] by Figure 6 (a) Current setpoint As input, current i s For the output, plot the current closed-loop control function G of the system given by the PIR controller and the PI controller under the given parameters. i_ref Bode plot of (s), as follows Figure 9 As shown, under the same PI parameters, the closed-loop system controlled by the PIR controller has a gain of 0dB (unity gain) and a phase shift of almost 0 at the resonant frequency, while the closed-loop system controlled by the PI controller exhibits both amplitude attenuation and phase shift. This demonstrates that the PIR controller can achieve zero steady-state error tracking of AC signals, thus compensating for the shortcomings of the traditional PI controller in controlling AC signals.

[0107] by Figure 6 (a) In this diagram, the back EMF disturbance term E is the input and the current i is the output. Draw the transfer function G of the motor phase current closed-loop control system for the back EMF E. i_E Bode plot of (s), as follows Figure 10 As shown in (a), it can be seen that the closed-loop control system has a strong ability to suppress DC, fundamental and third harmonic waves in back EMF disturbances.

[0108] Figure 6 (b) shows the Bode plot of the disturbance input and output as follows: Figure 10 As shown in (b). Figure 3According to equation (12), the disturbances caused by the modulation characteristics of the salient pole motor winding can be divided into two categories: one is the frequency conversion disturbance (M2 / 2) caused by the input voltage, and the other is the back electromotive force disturbance (M0 or M2 / 2) generated by the phase current. The input gain k corresponding to the two disturbances is... Figure 6 (b) differs from k) in that the input gain k for voltage-type frequency converter disturbances is M² / 2. Generally, when a motor is operating, the input voltage u is similar to (R... s +jω e Since L)i are of the same order of magnitude, when analyzing the back electromotive force disturbance generated by the phase current, s—>s±2jω e (2ω higher or lower than the input frequency) e The input gain of the part is M0, and its transfer function is denoted as C. i_2ω (s), s—>s±4jω e (2ω higher or lower than the input frequency) e The input gain of the part is M² / 2, and its transfer function is denoted as C. i_4ω (s).

[0109] Equation (12) and Figure 3 The mathematical model of the motor shown indicates that, due to the modulation characteristics of the salient-pole motor, the motor system generates disturbances at other frequencies when operating using the fundamental frequencies of voltage and current. These disturbances are predominantly third harmonics, with a small amount of fifth harmonics in addition. Figure 10 As shown in (b), the PIR controller has a good suppression effect on fundamental and third harmonic disturbances. In contrast, its ability to suppress fifth harmonic disturbances is somewhat worse. However, under the constrained salient pole motor parameters, the fifth harmonic disturbance is relatively small, and this suppression ability is acceptable.

[0110] The active disturbance rejection controller of this invention is a second-order extended state observer, expressed by the following formula:

[0111]

[0112] Connecting the PIR controller and the active disturbance rejection controller in parallel yields the following joint control law:

[0113]

[0114] The active disturbance rejection controller outputs u _ESO The process is as follows:

[0115] The current in phase s is i s The difference between the observed phase current z1 and the observed phase current z1 is multiplied by the first-order error coefficient β1 and the second-order error coefficient β2 to generate i. s β1, i s β2;

[0116] i sThe observed value z2 of the first derivative of the phase current after integration of β1, z2 and i s β2, u _ESO The phase current observation value z1 is generated by adding b0(t) and then integrating it.

[0117] Phase current setpoint The circuit is divided into two paths. One path is subtracted from the observed phase current value z1 and then divided by the proportional control law k. pz The product is used as the value of the first branch;

[0118] Phase current given value i s ref Another path is differentiated and then subtracted from the observed value z2 of the first derivative to obtain the value of the second branch;

[0119] The sum of the first branch value and the second branch value, multiplied by 1 / b0(t), generates the active disturbance rejection controller output u. _ESO .

[0120] The control of the joint control law in equation (16) mainly includes three parts. z2 is used to compensate for disturbances caused by changes in the current setpoint, including amplitude changes during dynamic operation and sinusoidal setpoint disturbances during steady state. z2 is used to compensate for disturbances observed by the extended state observer, including observed external harmonic disturbances and internal parameter disturbances. PIR The output of the PIR control law is the core of the combined control law. It is used to control the fundamental wave of the motor's phase current while suppressing the third harmonic of the current, ensuring the normal output of the motor torque.

[0121] In summary, an isolated current control method based on the joint control of a proportional-integral-resonant (PIR) controller and an active disturbance rejection (ESO) controller is obtained, and its structure is as follows: Figure 11 As shown.

[0122] This method enables individual control of each phase winding of the motor, and its significance lies in: (1) enabling the fault-tolerant motor system to achieve control isolation in addition to phase-to-phase physical isolation, thermal isolation and magnetic isolation; (2) enabling the fault-tolerant motor system to change from fault state to fault-tolerant operating condition without changing the control strategy and modulation strategy, only the setpoint of the phase current needs to be changed; (3) the fault-tolerant function that this method can achieve is more diverse than that of a motor system with a star connection of half-bridge drive windings, and the fault-tolerant capability is stronger; (4) compared with traditional hysteresis control, this method can enable the fault-tolerant motor system to be driven at a fixed switching frequency; (5) compared with traditional PI control, this method has better performance in high-speed operation of the fault-tolerant motor system.

[0123] To verify the effectiveness of the isolated current control method proposed in this invention, a simulation model was built to simulate the control effects of the fault-tolerant motor system under winding isolation using both traditional PI control and the proposed isolated current control method. The results are as follows: Figure 12 As shown in the figure. Simulation results demonstrate that the isolation control method proposed in this invention can achieve tracking of the given AC current.

[0124] Specific Implementation Method Two, as follows: Figure 13 This implementation method adds a speed loop (using a PI converter) and builds a simulation model to simulate the starting speed of a six-phase motor from low speed to a given speed under a certain torque load. The simulation results for speed, torque, and current are as follows: Figure 13 As shown, simulation results indicate that, under the action of the outer speed loop given by the ramp, the isolation control method proposed in this invention can adapt well to the variable speed operation of the fault-tolerant motor system, and further realize the speed regulation and servo application of the fault-tolerant motor system.

[0125] While the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that different dependent claims and features described herein can be combined in ways different from those described in the original claims. It is also understood that features described in conjunction with individual embodiments can be used in other described embodiments.

Claims

1. An isolated current control method applicable to salient-pole multiphase permanent magnet synchronous motors, characterized in that, The method is as follows: The multiphase permanent magnet synchronous motor uses an H-bridge to drive each phase winding. The current of each phase winding is individually controlled by a current loop controller. The current loop controller includes a PIR controller, which is composed of a PI controller, a fundamental resonant controller, and a third harmonic resonant controller. The transfer function of the PIR controller is: in This is the proportionality coefficient. The integral coefficient is... The resonance coefficient, and Let be the fundamental frequency and the third harmonic resonance coefficients, respectively. Represents the Laplace operator. Indicates the resonant frequency. This indicates the frequency selectivity bandwidth of the resonant controller. and These represent the frequency selection bandwidths of the fundamental and third harmonic resonant controllers, respectively. The current loop controller also includes an active disturbance rejection controller (PIR controller). The PIR controller and the PIR controller are connected in parallel to form a joint control law. The PIR controller outputs a value... With the output value of the active disturbance rejection controller The sum, after subtracting the interference term E, is used to generate the current of the control winding; The active disturbance rejection controller uses a second-order extended state observer, denoted as: In the formula, Given the output gain, , For the electric angle of the motor, Represents the constant components of the inductance matrix. This represents the amplitude of the second harmonic component of the inductance matrix; , These are the first-order error coefficients and the second-order error coefficients, respectively. , , These are the eigenvalues ​​of the extended state observer; Observational error; For the m-phase motor Phase winding current, ; These are the phase current observations. The observed value of the first derivative of the phase current. These are the observed values ​​of phase current disturbance. The observed value of the first derivative of the phase current disturbance; This is the output value of the active disturbance rejection controller; Connecting the PIR controller and the active disturbance rejection controller in parallel yields the following joint control law: In the formula For current deviation compensation, , The proportional control law of the active disturbance rejection controller. Given the phase current value, For current disturbance compensation. It is the first derivative of the phase current setpoint, used to compensate for disturbances caused by changes in the current setpoint.

2. The isolated current control method for a salient-pole multiphase permanent magnet synchronous motor according to claim 1, characterized in that, The relationship between the voltage and current of each phase winding of a multiphase motor in the complex frequency domain can be expressed as follows: In the formula u s (s) represents the phase voltage in a salient-pole permanent magnet synchronous motor. , E(s) is the back electromotive force of the motor, including the permanent magnet back electromotive force and the electric excitation back electromotive force; The constant component representing the reciprocal of inductance. The second harmonic coefficient represents the reciprocal of the inductance. Represents the resistance matrix. This indicates the electrical frequency of the motor.

3. The isolated current control method for a salient-pole multiphase permanent magnet synchronous motor according to claim 2, characterized in that, The motor inductance meets the following conditions: 。 4. The isolated current control method for a salient-pole multiphase permanent magnet synchronous motor according to claim 1, characterized in that, It also includes a speed loop, which employs proportional-integral control.

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