A two-degree-of-freedom current controller for permanent magnet synchronous motors

Abstract

The application belongs to the field of aviation electrical and power technology, and particularly relates to a two-degree-of-freedom current controller of a permanent magnet synchronous motor. The application provides a two-degree-of-freedom current controller, which makes up for the deficiency of the existing proportional integral controller technology. The application applies the improved two-degree-of-freedom control technology to current tracking of the permanent magnet synchronous motor, is simple to operate, has strong robustness, and can realize high-precision tracking of speed. The method has certain expansibility and can be popularized to other fields.

Description

Technical Field

[0001] This invention belongs to the field of aviation electrical and power technology, and specifically relates to a two-degree-of-freedom current controller for a permanent magnet synchronous motor. Background Technology

[0002] Europe and the United States have made far greater strides than my country in the electrification of aircraft and aero engines. This electrification has accelerated the development of next-generation "more electric" and "all-electric" aircraft, such as the US Joint Strike Fighter F-35, Boeing B787, and Airbus A380. "More electric / all-electric" aircraft simplify aircraft and engine equipment, reduce engine frontal area, improve aerodynamics, reduce weight, enhance aircraft performance, and reduce fuel consumption. Specifically, tests of GE's FPS-9 energy-efficient engine show that extracting 1.5 pounds of air per second increases fuel consumption by 2.18%, while extracting the equivalent of 150 shaft horsepower of mechanical energy from bleed air only increases fuel consumption by 0.85%.

[0003] Integrated starter / generator technology is a key indicator of aircraft electrification. This technology combines the aircraft's starting and power generation systems. During takeoff, the integrated starter / generator ignites and starts the engines; after normal flight, it provides power to the aircraft's electrical loads. The integrated starter / generator operates in both motoring and generating modes, and the bidirectional power converter operates in both inversion and rectification modes. The bidirectional power converter's operation is controlled by a control system. When the bidirectional power converter operates in inversion mode, the outer loop controller is a speed controller, and the inner loop controller is a current controller. When the bidirectional power converter operates in rectification mode, the outer loop controller is a voltage controller, and the inner loop controller is a current controller. Therefore, for the bidirectional power converter's control system, the current controller is the core. The current controller is responsible for tracking the given current value. Because the integrated starter / generator operates in a complex and variable environment, influenced by the aircraft's flight environment, the current controller is susceptible to various disturbances, such as periodic and aperiodic interference, affecting its control performance. Therefore, it is necessary to design a suitable controller that enables the system to quickly and accurately track a given signal and effectively suppress disturbances. Tracking performance mainly evaluates the controller's ability to track a given signal, while disturbance rejection performance mainly evaluates the controller's ability to suppress disturbances. However, traditional single-degree-of-freedom controllers (such as proportional-integral controllers) can usually only make a trade-off between tracking performance and disturbance rejection performance, which cannot meet the requirements of high-performance current controllers. Two-degree-of-freedom controllers can solve the problem that single-degree-of-freedom controllers cannot balance tracking performance and disturbance rejection performance. Researching advanced two-degree-of-freedom control strategies is of great significance for improving the current loop controller of permanent magnet synchronous motors.

[0004] Therefore, especially when there are unknown interferences in the permanent magnet synchronous motor, accurate tracking of the permanent magnet synchronous motor current is very valuable, as it can improve the reliability of the permanent magnet synchronous motor and thus ensure the stable operation of the aviation starter / generator system.

[0005] In the field of permanent magnet synchronous motor (PMSM) current controllers, published literature both domestically and internationally primarily focuses on designing current controllers based on proportional-integral (PI) or proportional-integral-derivative (PI-DE) methods. In a dual-loop control system for a PMSM, the current loop controller is required to rapidly track the given current while effectively suppressing disturbances within the current loop. Traditional proportional-integral (PI) controllers, being single-degree-of-freedom controllers, cannot simultaneously achieve both tracking and disturbance rejection performance, requiring a trade-off between these two aspects. This clearly fails to meet the increasingly demanding performance requirements of modern PMSM control systems. Therefore, traditional two-degree-of-freedom control techniques have been proposed. While the performance of typical two-degree-of-freedom proportional-extended state observer (P-LESO) current controllers has been improved to some extent, their disturbance rejection remains effective, necessitating active suppression of enhanced disturbances.

[0006] Current design methods for permanent magnet synchronous motor (PMSM) current controllers are increasingly failing to meet practical engineering needs, particularly in suppressing the sixth current harmonic. Therefore, developing an effective and practically applicable design method for high-precision PMSM current controllers has become a pressing issue. This also presents broad research and application prospects for high-precision and robust current tracking in PMSMs. Summary of the Invention

[0007] This invention is a two-degree-of-freedom current tracking control method proposed to address the problems of low accuracy, weak robustness, and low accuracy of current tracking control for permanent magnet synchronous motors in existing technologies.

[0008] A two-degree-of-freedom current controller for a permanent magnet synchronous motor, the specific steps of which are as follows:

[0009] Step 1: To facilitate analysis, we first define a coordinate system, the specific mathematical expression of which is as follows:

[0010] For ease of analysis, we first define a coordinate system. The purpose of defining the coordinate system is to represent the alternating current V in the three-phase stationary rotating coordinate system. abc Transform to two-phase stationary coordinate system υ αβ Finally, it is transformed into a two-phase rotating coordinate system υ qdThus, the decoupling control of the three-phase current of the permanent magnet synchronous motor is realized. That is, only the dq axis current needs to be controlled, so as to achieve the purpose of controlling the three-phase current of abc. The transformation relationship between the various coordinate systems can be described by formula (1).

[0011]

[0012] in,

[0013] V abc ·[V a V b V c ] T ,υ αβ =[υ α υ β ] T ,υ qd =[υ q υ d ] T ,

[0014] This is the electrical angle of the permanent magnet synchronous motor.

[0015] Step 2: Based on the vector control architecture of the permanent magnet synchronous motor, obtain the current loop model of the permanent magnet synchronous motor;

[0016] Since permanent magnet synchronous motors generally adopt a cascade control structure, the inner loop is a dq axis current loop controller, and its current loop mathematical model satisfies formula (2).

[0017]

[0018] In the formula, u d u q These represent the input voltages along the d and q axes, respectively; ω e Indicates the electrical angular velocity of a permanent magnet synchronous motor; i d i q These represent the components of the stator current along the dq axis; L d L q Let L and q represent the components of the inductance along the d and q axes, respectively. For a surface-mounted permanent magnet synchronous motor, L satisfies... d =L q ; ψ f This indicates the magnetic flux linkage of a permanent magnet.

[0019] Step 3: Considering the periodic disturbance, the mathematical expression between the periodic disturbance and the permanent magnet synchronous motor current is obtained as follows:

[0020] In the presence of periodic disturbances, these disturbances are generally caused by the dead time of the flux harmonic components. To avoid the simultaneous conduction of the upper and lower switches on the same bridge arm of a three-phase bridge inverter circuit, which would generate a short-circuit current, a short delay is required after the upper (lower) bridge arm is turned off before the lower (upper) bridge arm is turned on. This short delay is the dead time. The flux harmonic components and the dead time can introduce periodic disturbances, satisfying equations (3) and (4) respectively.

[0021]

[0022]

[0023] In the formula, ψ fd6n ψ fq6n T represents the amplitude of the 6nth flux linkage harmonic component along the d-axis and q-axis, respectively. s For the switching period, T d For dead time, U dc ψ is the DC bus voltage, where n is a positive integer. fd Let ψ be the amplitude of the 6nth harmonic flux linkage component along the d-axis. fq This represents the amplitude of the 6nth harmonic flux linkage component along the q-axis.

[0024] Step 4: Considering the aperiodic disturbance, the mathematical expression between the aperiodic disturbance and the permanent magnet synchronous motor current is obtained as follows:

[0025] When considering aperiodic disturbances, the aperiodic disturbances caused by inductance parameters are mainly considered, as shown in formula (5).

[0026]

[0027] Among them, let b d =1 / L d b q =1 / L q Define L d0 L q0 These are the nominal values ​​of the inductance parameters of the motor's dq axis.

[0028] Δb d =b d -b d0 , Δb q =b q -b q0 f d1 and f q1 Let f represent the periodic perturbations along the d-axis and q-axis, respectively. d2 ,f q2 Δb represents the non-periodic disturbances along the d-axis and q-axis, respectively. d =b d -bd0 , Δb q =b q -b q0 f d1 and f q1 Let f represent the periodic perturbations along the d-axis and q-axis, respectively. d2 ,f q2 These represent the non-periodic disturbances along the d-axis and q-axis, respectively.

[0029] Step 5: Design a two-degree-of-freedom current controller based on steps 3 and 4;

[0030] Step 6: In the MATLAB / Simulink environment, using modular modeling technology, a permanent magnet synchronous motor model and a permanent magnet synchronous motor current controller are built to verify the performance of the two-degree-of-freedom controller. The tracking effect of the two-degree-of-freedom speed controller is compared with that of the proportional-integral speed controller. Under the conditions of periodic and aperiodic disturbances, the two-degree-of-freedom current controller has higher accuracy, indicating that the method proposed in this invention can accurately track and control the current.

[0031] The beneficial effects of this invention are:

[0032] Existing current controllers for permanent magnet synchronous motors (PMSMs) face challenges in achieving high-precision current tracking. In engineering practice, speed controllers are currently commonly designed using proportional-integral (PI) methods. This invention provides a two-degree-of-freedom (DOF) current controller, overcoming the shortcomings of existing PIF technology. This invention applies improved two-degree-of-freedom control technology to current tracking in PMSMs, offering simple operation, strong robustness, and high-precision speed tracking. This method also has scalability and can be extended to other fields. Attached Figure Description

[0033] Figure 1 A schematic diagram of an improved two-degree-of-freedom q-axis current controller.

[0034] Figure 2 A schematic diagram of an improved two-degree-of-freedom d-axis current controller.

[0035] Figure 3(a) is a schematic diagram of the current response of the q-axis current controller under nominal conditions.

[0036] Figure 3(b) is a schematic diagram of the current response of the q-axis current controller under nominal conditions.

[0037] Figure 4(a) shows the current response of the q-axis current controller when the stator inductance becomes 50% of the nominal value.

[0038] Figure 4(b) is a schematic diagram of the current response of the d-axis current controller when the stator inductance becomes 50% of the nominal value.

[0039] Figure 5(a) shows the current response of the q-axis current controller when the stator inductance becomes 150% of the nominal value.

[0040] Figure 5(b) is a schematic diagram of the current response of the d-axis current controller when the stator inductance becomes 150% of the nominal value.

[0041] Figure 6(a) is a schematic diagram of the current response of the q-axis current controller when the inverter dead time is applied.

[0042] Figure 6(b) is a schematic diagram of the current response of the d-axis current controller when the inverter dead time is applied.

[0043] Figure 7(a) shows the FFT analysis of the A-phase stator current harmonics using a PI controller current loop.

[0044] Figure 7(b) shows the A-phase stator current harmonic FFT analysis using the P-LESO current loop.

[0045] Figure 7(c) shows the A-phase stator current harmonic FFT analysis using the QPRC–LESO current loop. Detailed Implementation

[0046] To make the objectives, techniques, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and examples.

[0047] A two-degree-of-freedom current controller for a permanent magnet synchronous motor (PMSM) is disclosed for tracking and controlling the current of the PMSM. Disturbances in the current loop can be mainly categorized into aperiodic and periodic disturbances. Aperiodic disturbances include changes in motor parameters, back electromotive force, and some unknown disturbances; periodic disturbances mainly include dead-zone effects, voltage harmonics caused by inverter nonlinearity, and permanent magnet flux linkage harmonics. These voltage and flux linkage harmonics generate current harmonics of corresponding frequencies in the current loop. The core work of this patent is to design a two-degree-of-freedom current controller to track the current of the PMSM under the conditions of periodic and aperiodic disturbances, including the following steps:

[0048] Step 1: For ease of analysis, first define a coordinate system. The purpose of defining the coordinate system is to represent the alternating current V in the three-phase stationary rotating coordinate system. abc Transform to two-phase stationary coordinate system υ αβ Finally, it is transformed into a two-phase rotating coordinate system υ qd Thus, the decoupling control of the three-phase current of the permanent magnet synchronous motor is realized. That is, only the dq axis current needs to be controlled, so as to achieve the purpose of controlling the three-phase current of abc. The transformation relationship between the various coordinate systems can be described by formula (1).

[0049]

[0050] in,

[0051] V abc =[V a V b V c ] T ,υ αβ [υ α υ β ] T ,υ qd =[υ q υ d ] T ,

[0052] This is the electrical angle of the permanent magnet synchronous motor.

[0053] Step 2: Since permanent magnet synchronous motors generally adopt a cascade control structure, the inner loop is a dq axis current loop controller, and its current loop mathematical model satisfies formula (2).

[0054]

[0055] In the formula, u d u q These represent the input voltages along the d and q axes, respectively; ω e Indicates the electrical angular velocity of a permanent magnet synchronous motor; i d i q These represent the components of the stator current along the dq axis; L d L q Let L and q represent the components of the inductance along the d and q axes, respectively. For a surface-mounted permanent magnet synchronous motor, L satisfies... d =L q ; ψ f This indicates the magnetic flux linkage of a permanent magnet.

[0056] Step 3: Due to the presence of periodic disturbances, these disturbances are generally caused by the dead time of the flux harmonic components. To avoid the simultaneous conduction of the upper and lower switches on the same bridge arm of a three-phase bridge inverter circuit, which would generate a short-circuit current, a short delay is needed after the upper (lower) bridge arm is turned off before the lower (upper) bridge arm is turned on. This short delay is the dead time. The flux harmonic components and the dead time can introduce periodic disturbances, satisfying formulas (3) and (4) respectively.

[0057]

[0058]

[0059] In the formula, ψfd6n ψ fq6n T represents the amplitude of the 6nth flux linkage harmonic component along the d-axis and q-axis, respectively. s For the switching period, T d For dead time, U dc ψ is the DC bus voltage, where n is a positive integer. fd Let ψ be the amplitude of the 6nth harmonic flux linkage component along the d-axis. fq This represents the amplitude of the 6nth harmonic flux linkage component along the q-axis.

[0060] Step 4: Considering aperiodic disturbances, we mainly consider aperiodic disturbances caused by inductance parameters, as shown in formula (5);

[0061]

[0062] Among them, let b d =1 / L d b q =1 / L q Define L d0 L q0 Δb represents the nominal value of the motor's dq-axis inductance parameter. d =b d -b d0 , Δb q =b q -b q0 f d1 and f q1 Let f represent the periodic perturbations along the d-axis and q-axis, respectively. d2 ,f q2 Δb represents the non-periodic disturbances along the d-axis and q-axis, respectively. d =b d -b d0 , Δb q =b q -b q0 f d1 and f q1 Let f represent the periodic perturbations along the d-axis and q-axis, respectively. d2 ,f q2 These represent the non-periodic disturbances along the d-axis and q-axis, respectively.

[0063] Step 5: Considering periodic and aperiodic disturbances, the effects of periodic and aperiodic disturbances on the d / q axis current can be derived mathematically, i.e., compared with formula (2), formula (6) is obtained.

[0064]

[0065] in,

[0066]

[0067] Step 6: Based on the introductions in Steps 4 and 5, it is shown that periodic and aperiodic disturbances will cause static errors in the current tracking of permanent magnet synchronous motors. In order to eliminate these static errors, it is necessary to actively observe and compensate for periodic and aperiodic disturbances. Traditional proportional-integral speed controllers / proportional-integral-derivative speed controllers cannot actively suppress periodic and aperiodic disturbances. At the same time, their anti-interference ability is not strong and they are passive disturbance suppression. Therefore, a two-degree-of-freedom current controller needs to be designed here to suppress periodic and aperiodic disturbances.

[0068] (1) Traditional two-degree-of-freedom controller (abbreviated as P-LESO)

[0069] Define f q =f q1 +f q2 For the total disturbance of the q-axis current loop, f q Expanding to a new state variable, the state-space equation of the q-axis current loop can be obtained as shown in equation (7);

[0070]

[0071] Where h represents the total disturbance f q The derivative of the derivative. For the system shown in equation (7), the linear extended state observer is designed as shown in equation (8).

[0072]

[0073] in, and Representing i respectively q and f q The estimated values ​​are β1 and β2, which represent the gain coefficients of the linearly extended state observer.

[0074] Define f d =f d1 +f d2 The total disturbance of the d-axis current loop will be expanded into a new state variable, and the state space equation of the d-axis current loop can be obtained as shown in formula (9).

[0075]

[0076] Among them, h d The total disturbance f represents d The derivative of the derivative. For the system shown in equation (9), design a linear extended state observer as shown in equation (10).

[0077]

[0078] in, and Representing i respectively d and f d The estimated value, β 1d β 2d This represents the gain coefficient of the linearly extended state observer.

[0079] (2) Improved two-degree-of-freedom current controller (abbreviated as QPR-LESO)

[0080] The linear extended state observer acts as a low-pass filter for the total disturbance, accurately tracking the aperiodic disturbances in the low-frequency part of the total disturbance and effectively suppressing high-frequency noise. However, its tracking effect on periodic disturbances near the mid-frequency range is relatively poor. Therefore, the linear extended state observer can be modified so that it is only used to estimate aperiodic disturbances, while the quasi-resonant controller tracks periodic disturbances. The two work in parallel to track the total disturbance in the system and effectively compensate the output of the main controller, thereby significantly improving the system's disturbance rejection performance. The redesigned linear extended state observer is shown in Equation (11).

[0081]

[0082] in, This is an estimate of the non-periodic disturbance.

[0083] The redesigned linear extended state observer outputs estimates of aperiodic disturbances instead of total disturbances, which reduces the burden on the linear extended state observer to some extent, thereby improving its estimation performance for aperiodic disturbances.

[0084] The following quasi-resonant controller is designed to address the 6th current harmonic in the current loop:

[0085]

[0086] The input to the quasi-resonant controller is The output is an estimate of the periodic disturbance. The periodic disturbance can be suppressed by directly compensating the output of the quasi-resonant controller to the output of the main controller. Together, they form a quasi-proportional resonant controller. The controlled object after compensating for the total disturbance can be considered a pure integral element; therefore, the main controller can be designed as a proportional element with the following transfer function:

[0087] G c (s)=k p (13)

[0088] Combining the above steps, we can obtain the q-axis current loop two-degree-of-freedom control, such as... Figure 1As shown. From Figure 1 It can be seen that the q-axis current loop two-degree-of-freedom control structure includes a main controller loop and a disturbance compensation loop, wherein the disturbance compensation loop includes a quasi-resonant controller and a linear extended state observer.

[0089] The linearly extended state observer is redesigned as follows:

[0090]

[0091] in, This is an estimate of the non-periodic disturbance.

[0092] As can be seen from equation (14), the redesigned linear extended state observer outputs an estimate of the aperiodic disturbance rather than the total disturbance. This can reduce the burden on the linear extended state observer to some extent, thereby improving its estimation effect on aperiodic disturbances.

[0093] The following quasi-resonant controller is designed to address the 6th current harmonic in the current loop:

[0094]

[0095] The input to the quasi-resonant controller is The output is an estimate of the periodic disturbance. The periodic disturbance can be suppressed by directly compensating the output of the quasi-resonant controller to the output of the main controller. Together, they form a quasi-proportional resonant controller. The controlled object after compensating for the total disturbance can be considered a pure integral element; therefore, the main controller can be designed as a proportional element with the following transfer function:

[0096] G c (s)=k p (16)

[0097] Combining the above steps, we can obtain the d-axis current loop two-degree-of-freedom control, such as... Figure 2 As shown. From Figure 2 It can be seen that the d-axis current loop two-degree-of-freedom control structure includes a main controller loop and a disturbance compensation loop, wherein the disturbance compensation loop includes a quasi-resonant controller and a linear extended state observer.

[0098] Step 7: In the MATLAB / Simulink environment, using modular modeling technology, build a permanent magnet synchronous motor model and the two-degree-of-freedom current controller proposed in this patent. Verify the performance of the two-degree-of-freedom current controller proposed in this patent. Compare the tracking performance of the two-degree-of-freedom speed controller proposed in this patent with current controllers designed by traditional methods (traditional single-degree-of-freedom proportional-integral (PI) and proportional controllers based on linear extended state observers (P-LESO)). Under the conditions of periodic and aperiodic disturbances, the two-degree-of-freedom current controller proposed in this patent has higher tracking accuracy, thus demonstrating that the method proposed in this invention can accurately track the speed of the permanent magnet synchronous motor.

[0099] The specific comparison process is as follows:

[0100] (1) Tracking performance simulation verification

[0101] First, the tracking performance of the d-q axis current was verified using three different controllers for the current loop. The simulation results are shown in Figure 3. Figure 3 shows that when the current loop uses a PI controller, the overshoot is largest during the no-load start-up acceleration phase and the load adjustment phase. Compared to the PI controller, the overshoot is reduced when the current loop uses a P-LESO controller during these phases. Finally, compared to the former two, the overshoot is smallest when the current loop uses a two-degree-of-freedom control structure based on QPRC-LESO during these phases. Therefore, the two-degree-of-freedom control structure based on QPRC-LESO can significantly improve the system's tracking performance for a given signal.

[0102] (2) Simulation verification of anti-periodic disturbance performance

[0103] Next, we verified the anti-periodic disturbance performance of the current loop using three different controllers. The aperiodic disturbances in the current loop are mainly changes in the stator winding inductance and resistance parameters. We designed two simulations by changing the motor inductance parameters to 50% and 150% of their nominal values, respectively, while keeping other conditions constant. The simulation results are shown in Figures 4 and 5. The simulation results show that when the stator inductance parameters change, the d-q axis current of the current loop using the PI controller exhibits significant overshoot during both the no-load start-up acceleration phase and the load adjustment phase, and also shows oscillations during the start-up acceleration phase. The current loops using the P-LESO and QPRC-LESO controllers show very small overshoots and virtually no oscillations during the start-up acceleration phase. Therefore, the linear extended state observer can effectively suppress aperiodic disturbances in the current loop, and the P-LESO and QPRC-LESO controllers with linear extended state observers are more robust to parameter changes.

[0104] (3) Simulation verification of anti-periodic disturbance performance

[0105] To verify the anti-periodic disturbance performance of the current loop using three different controllers, a 2s inverter dead time was added to the simulation model. As shown in the above disturbance analysis of the current loop, the addition of the dead time generates a 6nth current harmonic on the d-q axis. This verifies the current harmonic suppression capability of the current loop using three different controllers. The d-q axis current response waveform with the inverter dead time added is shown in Figure 6. Figure 6 shows that due to the presence of current harmonics, the current in the steady-state phase fluctuates significantly when using a PI controller. Compared to the PI controller, the current fluctuation amplitude in the steady-state phase of the P-LESO current loop is reduced, but it is still quite noticeable. Compared to the former two, the current fluctuation in the steady-state phase of the QPRC-LESO current loop is also present, but the amplitude is much smaller. To more intuitively demonstrate the harmonic suppression effect of the current loop using three different controllers, the FFT analysis tool in the Simulink module was used to perform FFT analysis on the A-phase current. The results are shown in Figure 7. As shown in Figure 7(a), the A-phase stator current of the current loop using the PI controller has a significant amount of 5th and 7th harmonics, with the 5th harmonic content at 5.54% and the 7th harmonic content at 4.14%, resulting in a total harmonic distortion (THD) of 8.74%. As shown in Figure 7(b), the A-phase stator current of the current loop using the P-LESO controller has a small amount of 5th and 7th harmonics, with the 5th harmonic content at 2.88% and the 7th harmonic content at 1.98%, resulting in a THD of 6.39%. As shown in Figure 7(c), the 5th and 7th harmonics in the A-phase stator current of the current loop using the QPRC-LESO controller are almost completely eliminated, with the 5th harmonic content at 0.11% and the 7th harmonic content at 0.10%, resulting in a THD of 5.52%. Based on the above analysis, we can conclude that the PI controller is very poor at suppressing aperiodic disturbances in the current loop; P-LESO can suppress aperiodic disturbances to some extent, but this depends on a large observer gain; while the proposed QPRC-LESO can effectively suppress periodic disturbances in the current loop and significantly improve the current loop's resistance to periodic disturbances.

[0106] In summary, the two-degree-of-freedom current controller proposed in this patent has higher tracking accuracy, thus demonstrating that the method proposed in this invention can accurately track the speed of a permanent magnet synchronous motor.

Claims

1. A two-degree-of-freedom current controller for a permanent magnet synchronous motor, characterized in that, The steps are as follows: Step 1: First, define a coordinate system. The purpose of defining a coordinate system is to represent the alternating current in a three-phase stationary rotating coordinate system. Transform to two-phase stationary coordinate system Finally, it is transformed into a two-phase rotating coordinate system. Thus, the decoupling control of the three-phase current of the permanent magnet synchronous motor is realized. That is, only the dq axis current needs to be controlled, so as to achieve the purpose of controlling the three-phase current of abc. The transformation relationship between the various coordinate systems can be described by formula (1). （1） in, The electrical angle of the permanent magnet synchronous motor; Step 2: Based on the vector control architecture of the permanent magnet synchronous motor, obtain the current loop model of the permanent magnet synchronous motor; Since permanent magnet synchronous motors generally adopt a cascade control structure, the inner loop is a dq axis current loop controller, and its current loop mathematical model satisfies formula (2). （2） In the formula, , These represent the input voltages along the d and q axes, respectively. This indicates the electrical angular velocity of a permanent magnet synchronous motor; , These represent the components of the stator current along the dq axis, respectively. , Let represent the components of the inductance along the d and q axes, respectively. For a surface-mounted permanent magnet synchronous motor, the following conditions must be met: ; Indicates permanent magnet flux linkage; Step 3: Considering the periodic disturbance, the mathematical expression between the periodic disturbance and the permanent magnet synchronous motor current is obtained as follows: The magnetic flux harmonic components and dead time can introduce periodic disturbances, satisfying formulas (3) and (4) respectively. （3） （4） In the formula, , These represent the amplitudes of the 6nth flux harmonic components along the d-axis and q-axis, respectively. For the switching cycle, Dead time, This is the DC bus voltage. It is a positive integer; The amplitude of the 6nth flux linkage harmonic component along the d-axis. Given the amplitude of the 6nth flux linkage harmonic component under the q-axis, the expression for the periodic disturbance can be obtained, as shown in (5); （5） and These represent periodic disturbances along the d-axis and q-axis, respectively. Step 4: Considering the aperiodic disturbance, the mathematical expression between the aperiodic disturbance and the permanent magnet synchronous motor current is obtained as follows: When considering aperiodic disturbances, the aperiodic disturbances caused by inductance parameters are mainly considered, as shown in formula (6); （6） Among them, let , ,definition , These are the nominal values ​​of the inductance parameters of the motor's dq axis. , , and These represent periodic perturbations along the d-axis and q-axis, respectively; [Definition] , These represent the non-periodic disturbances along the d-axis and q-axis, respectively. , , and These represent periodic perturbations along the d-axis and q-axis, respectively; [Definition] , These represent the non-periodic disturbances along the d-axis and q-axis, respectively. Step 5: Design a two-degree-of-freedom current controller based on steps 3 and 4; In step 5, the specific operation of the q-axis current controller of the two-degree-of-freedom current controller is as follows: The linearly extended state observer is designed as follows: （12） in, This is an estimate of the non-periodic disturbance; The following quasi-resonant controller is designed to address the 6th current harmonic in the current loop: （13） The input of the quasi-resonant controller is The output is an estimate of the periodic disturbance. The periodic disturbance can be suppressed by directly compensating the output of the quasi-resonant controller to the output of the main controller. The two together constitute a quasi-proportional resonant controller. The main controller is designed as a proportional element, and its transfer function is as follows: （14） q The two-degree-of-freedom control structure of the shaft current loop includes a main controller loop and a disturbance compensation loop, wherein the disturbance compensation loop includes a quasi-resonant controller and a linear extended state observer; Step 6: In the MATLAB / Simulink environment, using modular modeling technology, build a permanent magnet synchronous motor model and a permanent magnet synchronous motor current controller to verify the performance of the two-degree-of-freedom controller.

2. The two-degree-of-freedom current controller for a permanent magnet synchronous motor as described in claim 1, characterized in that, In step 5, the specific operation of the d-axis current controller of the two-degree-of-freedom current controller is as follows: The linearly extended state observer is designed as follows: （15） in, This is an estimate of the non-periodic disturbance; The following quasi-resonant controller is designed to address the 6th current harmonic in the current loop: （16） The input of the quasi-resonant controller is The output is an estimate of the periodic disturbance. The periodic disturbance can be suppressed by directly compensating the output of the quasi-resonant controller to the output of the main controller. The two together constitute a quasi-proportional resonant controller. The main controller is designed as a proportional element, and its transfer function is as follows: （17） The d-axis current loop two-degree-of-freedom control structure includes a main controller loop and a disturbance compensation loop, wherein the disturbance compensation loop includes a quasi-resonant controller and a linear extended state observer.

3. A two-degree-of-freedom current controller for a permanent magnet synchronous motor as described in claim 1, characterized in that, In step 6, the specific operation is as follows: the tracking effect of the two-degree-of-freedom speed controller is compared with that of the proportional-integral speed controller. Under the condition of periodic disturbance and non-periodic disturbance, the two-degree-of-freedom current controller has higher accuracy, indicating that the method proposed in this invention can accurately track and control the current.

4. A two-degree-of-freedom current controller for a permanent magnet synchronous motor as described in claim 2, characterized in that, In step 6, the specific operation is as follows: the tracking effect of the two-degree-of-freedom speed controller is compared with that of the proportional-integral speed controller. Under the condition of periodic disturbance and non-periodic disturbance, the two-degree-of-freedom current controller has higher accuracy, indicating that the method proposed in this invention can accurately track and control the current.

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