Control device for an electric motor

Abstract

A control device of an electric motor includes: a higher harmonic voltage operation section that operates a higher harmonic voltage added to a fundamental voltage command value in a two-phase rotating coordinate system synchronized with a rotor phase; and a higher harmonic phase delay compensation section that compensates for a phase delay at the time of operation of the higher harmonic voltage operation section according to a higher harmonic order of the higher harmonic voltage and an electrical angular frequency of the fundamental voltage.

Description

Technical Field

[0001] This invention relates to a control device for an electric motor. Background Technology

[0002] Permanent magnet synchronous motors are designed so that the magnetic flux generated by the stator and the rotor are sinusoidal. However, in many motors, due to factors such as the stator tooth construction and the rotor magnet configuration, higher harmonics synchronized with the rotor position (the rotor's phase as observed from the stator) overlap the linked magnetic flux. This causes distortion in the linked magnetic flux, thus contributing to motor noise or torque pulsation. When this type of motor is used in electric vehicles, the vehicle's drivability and marketability deteriorate.

[0003] Distortion in the linked magnetic flux manifests as distortion in the induced voltage. Furthermore, the higher harmonic voltages contained in the induced voltage are generated synchronously with the rotor phase. Therefore, by superimposing the out-of-phase higher harmonic voltages according to the rotor phase using an inverter (controller), distortion can be suppressed, thereby suppressing noise and torque ripple generated from the motor. Such prior art is described, for example, in Patent Document 1. Existing technical documents Patent documents

[0004] Patent Document 1: Japanese Patent Application Publication No. 2004-032944 Summary of the Invention The problem the invention aims to solve

[0005] When generating voltage command values ​​based on rotor phase, the delay time from detecting the rotor phase to reflecting the inverter voltage needs to be considered. This is because the rotor phase advances during the delay time. In current vector control, the delay of the detected rotor phase is compensated when transforming the voltage command value coordinates on the rotating coordinates generated by the current controller into the voltage command value on the fixed coordinates. This is represented below as PWM phase delay compensation. PWM phase delay compensation compensates for the fundamental voltage and does not affect higher harmonic voltages superimposed in the controller.

[0006] The invention described in Patent Document 1 involves calculating an inverted high-order harmonic voltage based on the rotor's phase and superimposing it onto a voltage command value on a fixed coordinate of two phases. However, there is no PWM phase delay compensation for the superimposed high-order harmonic voltage in the controller. Therefore, it is difficult to correctly superimpose the high-order harmonic voltage at high motor speeds.

[0007] The problem to be solved by the present invention is to superimpose the anti-phase high-order harmonic voltage in the controller without phase delay in order to suppress noise and torque pulsation generated from the motor. Technical means to solve the problem

[0008] Here, the control device for the electric motor of the present invention includes: a high-order harmonic voltage calculation unit that calculates a high-order harmonic voltage that is added to the fundamental voltage command value in a two-phase rotating coordinate system synchronized with the rotor phase; and a high-order harmonic phase delay compensation unit that compensates for the phase delay during the calculation of the high-order harmonic voltage calculation unit based on the high-order harmonic number of the high-order harmonic voltage and the electrical angular frequency of the fundamental voltage. The effects of the invention

[0009] It can superimpose the inverse high-order harmonic voltages in the controller without phase delay in order to suppress noise and torque ripple generated from the motor. Attached Figure Description

[0010] Figure 1 It is a diagram showing the schematic circuit structure of a power conversion device. Figure 2 This is a processing block diagram illustrating the existing processing of controller 104. Figure 3 This is a diagram illustrating the principle of PWM phase delay compensation. Figure 4 This is a block diagram of the high-order harmonic overlay circuit 211. Figure 5 It is an analog waveform in the existing processing of controller 104. Figure 6 This is a processing block diagram illustrating the processing of the controller 104 in this invention. Figure 7 It is a simulated waveform in the processing of the controller 104 in this invention. Detailed Implementation

[0011] Hereinafter, embodiments of the power conversion device of the present invention will be described with reference to the accompanying drawings. Furthermore, in the drawings, the same symbols are used to denote the same elements, and repeated descriptions are omitted.

[0012] Figure 1 This is a diagram showing the schematic circuit configuration of a power conversion device. Figure 1 In this circuit, the power conversion device for the motor 102 mainly consists of an inverter 101, a battery 103, and a controller 104. The inverter 101 is a three-phase voltage-type two-level inverter.

[0013] The electric motor 102 is connected to the drive mechanism of the electric vehicle system, and the vehicle is propelled by the rotation of the electric motor 102. In this embodiment, the electric motor 102 is an AC motor, specifically a three-phase embedded magnet synchronous motor (IPMSM). The motor 102 operates through the interaction of magnetic flux generated from a permanent magnet disposed on a rotor (not shown) and magnetic fields generated by currents iu, iv, and iw flowing through the three-phase windings 105, 106, and 107 fixed to the armature. iu, iv, and iw are the U-phase current, V-phase current, and W-phase current, respectively. Furthermore, while a permanent magnet synchronous motor is used in this embodiment, the present invention applies to other synchronous motors, such as wound-excited synchronous motors.

[0014] The electric motor 102 includes a magnetic pole position sensor 108. The magnetic pole position sensor 108 functions to detect the magnetic pole position of the rotor of the electric motor 102 as its phase relative to the stator. The magnetic pole position sensor 108 outputs a rotor phase 109 (θ), which is input to the controller 104. Figure 2 The processing in the controller 104 regarding rotor phase 109(θ) will be described later. The magnetic pole position sensor 108 can be composed of a rotary transformer, a rotary encoder, an absolute encoder, etc.

[0015] Current sensor 110 detects the current flowing through windings 105, 106, and 107, and based on this, outputs U-phase current sensor signal 111 (representing current iu), V-phase current sensor signal 112 (representing current iv), and W-phase current sensor signal 113 (representing current iw), which are then input to controller 104. The processing of current sensor signals 111, 112, and 113 in controller 104 also utilizes... Figure 2 To be described later.

[0016] Inverter 101 includes switching elements 114, 115, 116, 117, 118, and 119, and freewheeling diodes 120, 121, 122, 123, 124, and 125. In this embodiment, switching elements 114 to 119 are Si-IGBTs, each including a gate terminal, a collector terminal, and an emitter terminal. Freewheeling diodes 120 to 125 are connected between the collector terminal and the emitter terminal of each switching element 114 to 119. When the collector terminal of the switching elements 114 to 119 becomes at a higher potential than the emitter terminal, current flows through the freewheeling diodes 120 to 125, preventing high reverse voltages from being applied to the switching elements 114 to 119. However, the inverter circuit is not limited to the combination of Si-IGBTs and freewheeling diodes; other semiconductor components can also be used to construct the inverter circuit.

[0017] The switching of switching elements 114-119 between on and off is achieved through gate drive signals 126, 127, 128, 129, 130, and 131 connected to the gate terminals of each switching element 114-119. Gate signal 132, which forms the basis of each gate drive signal 126-131, is generated by controller 104 and output to gate drive circuit 135. Gate drive circuit 135 converts gate signal 132 into the potential required for switching elements 114, 115, 116, 117, 118, and 119 between on and off, and outputs gate drive signals 126, 127, 128, 129, 130, and 131. The generation of gate signal 132 in controller 104 uses... Figure 2 To be described later.

[0018] The emitter terminal of switching element 114 and the collector terminal of switching element 115 are interconnected, with their connection point connected to winding 105, through which current iu flows. The emitter terminal of switching element 116 and the collector terminal of switching element 117 are interconnected, with their connection point connected to winding 106, through which current iv flows. The emitter terminal of switching element 118 and the collector terminal of switching element 119 are interconnected, with their connection point connected to winding 107, through which current iw flows. The collector terminals of switching elements 114, 116, and 118 are interconnected and connected to high-potential DC wiring 133. Additionally, the emitter terminals of switching elements 115, 117, and 119 are interconnected and connected to low-potential DC wiring 134.

[0019] Therefore, based on the generated gate signal 132, the controller 104 switches the switching elements 114, 115, 116, 117, 118, and 119 on and off at appropriate times, controlling the currents iu, iv, and iw flowing through the windings 105, 106, and 107 to achieve rotation control of the motor 102. The gate signal 132 is formed as a PWM (Pulse Width Modulation) signal, in which the currents iu, iv, and iw are sinusoidal signals with a phase difference of 120 degrees.

[0020] Voltage sensor 136 is connected to high-potential DC wiring 133 and low-potential DC wiring 134 to detect their potential difference. Since the potential difference between high-potential DC wiring 133 and low-potential DC wiring 134 is typically a high voltage, for example, above 100V, voltage sensor 136 generates a DC voltage sensor signal 137 (Vdc) that is converted into a low voltage that controller 104 can detect, and inputs it to controller 104.

[0021] The inverter 101 includes a smoothing capacitor 138 connected between the high-potential DC wiring 133 and the low-potential DC wiring 134. The smoothing capacitor 138 has the function of suppressing the DC voltage ripples generated by the switching operation of the switching elements 114-119.

[0022] In battery 103, the high-potential side terminal is connected to high-potential DC wiring 133, and the low-potential side terminal is connected to low-potential DC wiring 134. Thus, it functions as a DC power supply to power inverter 101 or motor 102.

[0023] Torque command 139(T*) from a higher-level controller of the vehicle, such as an ECU (Electronic Control Unit), is input to controller 104. Based on this torque command 139(T*), controller 104 performs torque control of motor 102. Furthermore, although in Figure 1 Not shown in the figure, but the rotor temperature of the motor 102 is input to the controller 104. The rotor temperature is preferably detected by a temperature sensor, but it can also be a composition of the temperature estimated by known methods.

[0024] Figure 2 This is a processing block diagram illustrating the existing processing of controller 104. In Figure 2 In the controller 104, the processing block includes: a current command calculation unit 201, a three-phase to two-phase conversion unit 202, a current control unit 203, a two-phase to three-phase conversion unit 204, a speed calculation unit 205, a PWM gate control signal generation unit (control signal generation unit) 206, a deviation calculator 207, a deviation calculator 208, a PWM phase delay compensator 209, and a high-order harmonic overlapper 211. The current command calculation unit 201, the three-phase to two-phase conversion unit 202, the current control unit 203, the two-phase to three-phase conversion unit 204, the deviation calculator 207, and the deviation calculator 208 form a three-phase voltage command value generation unit.

[0025] The controller 104 causes the three-phase currents iu, iv, and iw to flow to the motor 102 for rotation control, but internally the controller 104 uses a so-called current vector control method that processes the rotation in a coordinate system that transforms the three-phase fixed coordinates into the two-phase rotating coordinates that are synchronized with the rotor phase represented by the d-axis and q-axis.

[0026] The current command calculation unit 201 calculates the d-axis current command value id* and the q-axis current command value iq*. The torque command 139 (T*), rotational angular velocity ω, and DC voltage sensor signal 137 (Vdc) are input to the current command calculation unit 201, and the d-axis current command value id* and the q-axis current command value iq* are calculated. The rotational angular velocity ω is calculated by the speed calculation unit 205 based on the rotor phase 109 (θ).

[0027] The three-phase to two-phase conversion unit 202 performs coordinate transformation to the d-axis and q-axis for the current sensor signals 111(iu), 112(iv), and 113(iw) based on the rotor phase 109(θ), and outputs the d-axis detection current id and the q-axis detection current iq.

[0028] Deviation calculator 207 calculates the deviation between the d-axis current command value id* output from the current command calculation unit 201 and the d-axis detected current id output from the three-phase two-phase converter 202, and outputs the d-axis current deviation Δid to the current control unit 203. Deviation calculator 208 calculates the deviation between the q-axis current command value iq* output from the current command calculation unit 201 and the q-axis detected current iq output from the three-phase two-phase converter 202, and outputs the q-axis current deviation Δiq to the current control unit 203.

[0029] The current control unit 203 performs feedback control such that the d-axis current deviation Δid and q-axis current deviation Δiq, which represent the deviation between the command value (as a target value) and the measured value (as an output value), are zero. To update the output value, it calculates and outputs the d-axis voltage command Vd* and the q-axis voltage command Vq*, which are voltage commands. The feedback control in the current control unit 203 is, for example, performed using PI control.

[0030] Adder 212 adds Vd* and Vdhrm* output from the higher harmonic overlapper 211 (described later) to output a new d-axis voltage command value Vd**. Adder 213 adds Vq* and Vqhrm* output from the higher harmonic overlapper 211 (described later) to output a new q-axis voltage command value Vq**.

[0031] Vd** and Vq** are input to the two-phase to three-phase conversion unit 204, which calculates and outputs the three-phase voltage command values ​​Vu*, Vv*, and Vw* based on the corrected rotor phase 210 (θdelay).

[0032] The corrected rotor phase 210 (θdelay) input to the two-phase three-phase converter 204 is calculated by the PWM phase delay compensator 209 as in equations (1) and (2). θdelay=θ+ωxTdelay…(1) Tdelay=Ts[n]+0.5Ts[n+1]…(2) In equations (1) and (2), Tdelay is the control delay time, Ts[n] is the current control period (carrier period), and Ts[n+1] is the control period at the next interruption.

[0033] Here, PWM phase delay compensation, a technology related to this invention, will be explained. PWM phase delay compensation is a function that, when transforming a voltage command value on a rotating coordinate synchronized with the rotor phase to a value on a fixed UVW coordinate, estimates the rotor phase advancing during the control delay time to perform coordinate transformation. The reasons for generating the control delay time and the method of compensation will be explained below.

[0034] Figure 3 This example illustrates the relationship between the controller 4's action time (phase current and rotor phase detection and current control, PWM duty cycle update) and rotor phase, with the horizontal axis representing time, as well as the control delay time. However, the detection time of the DC voltage sensor signal 137 (Vdc) used for PWM duty cycle calculation is omitted. Figure 3 As shown, the update time of the PWM duty cycle is delayed by the current control period Ts[n] relative to the detection of phase current and rotor phase. In addition, the PWM duty cycle is constant during the control period Ts[n+1] at the next interruption, and the average PWM duty cycle during this period needs to be set. Therefore, the control delay time Tdelay from the detection of phase current and rotor phase to the update of PWM duty cycle is represented by the above equation (2).

[0035] If the rotational angular velocity ω changes slowly and does not change during the control delay time Tdelay, the rotor phase difference θdiff during the Tdelay period can be calculated as in equation (3).

[0036] θdiff=ω×Tdelay…(3) By adding θdiff to the detected rotor phase θ, the rotor phase θdelay during PWM duty cycle update can be estimated as shown in equation (1). The above is the principle of PWM phase delay compensation.

[0037] The PWM gate control signal generation unit (control signal generation unit) 206 calculates the PWM duty cycle based on the three-phase voltage command values ​​Vu*, Vv*, Vw*, and Vdc. Then, by comparing the PWM duty cycle with a carrier wave (not shown), it generates six gate signals 132 as PWM signals and outputs them to the gate drive circuit 135.

[0038] Figure 4This is a processing block diagram of the high-order harmonic overlapper 211. Through the calculations of equations (4) and (5), the d-axis high-order harmonic voltage command value Vdhrm* and the q-axis high-order harmonic voltage command value Vqhrm* are output. Vdhrm*=∑(AdixSIN(ni(e+φi))…(4) Vqhrm*=∑(AqixSIN(ni(e+φi))…(5) In equations (4) and (5), Adi is the amplitude of the higher harmonic voltage along the d-axis, Aqi is the amplitude of the higher harmonic voltage along the q-axis, ni is the order of the higher harmonic, and Φi is the phase of the higher harmonic voltage. Additionally, i is an integer from 1 to m. Vdhrm* and Vqhrm* become signals synthesized with multiple higher harmonic voltages. The overlapping higher harmonic voltages are selected based on the higher harmonic voltages contained in the induced voltage of the motor. Furthermore, the higher harmonic voltages contained in the induced voltage are mainly of order 6n±1 on the fixed coordinate system (n is a natural number other than 0, in the case of three-phase balance) and of order 6n on the rotating coordinate system synchronized with the rotor phase. Therefore, the overlapping higher harmonic is selected as order 6n.

[0039] Adi, Aqi, and Φi are set to cancel the high-order harmonic voltage superimposed on the induced voltage. These values ​​may not be fixed, but rather vary depending on torque, current, speed, etc., and can be modified through lookup tables, etc.

[0040] The above describes the existing processing of controller 4. Here, the issues with the existing processing will be explained.

[0041] Figure 5 Indicates in Figure 2 In the existing processing of the controller 104 shown, the waveforms (simulated waveforms) of the U-phase current iu and the q-axis higher harmonic voltage Vqhrm are superimposed only on the q-axis. The waveforms under ideal conditions are represented by dashed lines, and the waveforms under existing control are represented by solid lines. Furthermore, to compare low and high rotation speeds, the results at 300 rpm and 3000 rpm were compared. The ideal state represents the analysis results in a continuous system. For ease of comparison, the existing control does not simulate the inverter's PWM control but instead uses a model that considers the delay caused by discrete control. The control period is 10 kHz.

[0042] At a low rotation speed of 300 rpm, the q-axis higher harmonic voltage of the existing control is delayed relative to the ideal state, but not significantly. Therefore, it can be concluded that the U-phase current does not differ significantly between the ideal state and the existing control. On the other hand, at a high rotation speed of 3000 rpm, the q-axis higher harmonic voltage of the existing control is significantly delayed relative to the ideal state, thus indicating a large difference in the U-phase current.

[0043] Thus, without PWM phase delay compensation for the overlapping high-order harmonic voltages in the controller, as the motor speed increases, the high-order harmonic voltages that should be overlapping cannot be correctly overlapped. Consequently, not only are the noise and torque ripples generated from the motor not suppressed, but depending on the magnitude of the phase delay, the noise and torque ripples may also be amplified.

[0044] Figure 6 This is a processing block diagram illustrating the processing of the controller 104 in this invention. Figure 6 Yes Figure 2 The configuration includes an additional high-order harmonic phase delay compensation unit 601.

[0045] As in Figure 3 As explained in the text, the delay is Tdelay from the detection of the rotor phase until it is reflected in the voltage command. Since the phase of the nth higher harmonic is advanced n times relative to the fundamental wave, the phase θdiffn advanced by the nth higher harmonic during Tdelay is given by equation (6). θdiffn=n×ω×Tdelay…(6) The higher harmonic phase delay compensation unit 601 compensates for the phase delay relative to the higher harmonics by advancing the phase of the overlapping higher harmonics by θdiffn. Furthermore, if we consider the phase delay compensation for higher harmonics and express equations (4) and (5), it is as follows. Vdhrm*=∑(Adi×SIN(ni(θ+ω×Tdelay+φi))…(7) Vqhrm*=∑(Adi×SIN(ni(θ+ω×Tdelay+φi))…(8) The above formula can be transformed using formula (1) as follows. Vdhrm*=∑(Adi×SIN(ni(θdelay+φi))…(9) Vqhrm*=∑(Aqi×SIN(ni(θdelay+φi))…(10) According to equations (9) and (10), in actual installation, by... Figure 4 The input θ is replaced with the output θdelay of the PWM phase delay compensator 209, which can achieve phase delay compensation for higher harmonics.

[0046] The Vdhrm* and Vqhrm* calculated as described above are added to the dq-axis voltage command on the rotating coordinate system, and then converted into three-phase voltage command values ​​by the two-phase to three-phase conversion unit 204.

[0047] In this configuration, for overlapping high-harmonic voltage commands, phase delay compensation is performed twice: in the high-harmonic phase delay compensation unit 601 and in the two-phase / three-phase conversion unit 204 and the PWM phase delay compensator 209. The principle is explained below.

[0048] The overlapping higher harmonic voltages Vdhrm and Vqhrm are represented by the following equations (11) and (12). Vdhrm=Ad×SIN(n(θ+Φ))…(11) Vqhrm=Aq×SIN(n(θ+φ))…(12) The phase voltages Vu, Vv, and Vw after the overlap of higher harmonics are expressed as follows. Vu=Ad×SIN(n(θ+φ))×COSθ-Aq×SIN(n(θ+φ))×SINθ…(13) Vv=Ad×SIN(n(θ+φ))×COS(θ-2π / 3)-Aq×SIN(n(θ+φ))×SIN(θ-2π / 3)…(14) Vw=Ad×SIN(n(θ+Φ))×COS(θ+2π / 3)-Aq×SIN(n(θ+φ))×SIN(θ+2π / 3)…(15) In equation (13), Ad×SIN(n(θ+Φ)) or Aq×SIN(n(θ+Φ)) represents the overlapping higher harmonics, and the terms following them represent terms used for coordinate transformation. The same applies to equations (14) and (15). θ used in the calculation of higher harmonics and coordinate transformation needs to be calculated using θ reflecting the voltage of the inverter. That is, phase delay compensation is required for each calculation.

[0049] Therefore, phase delay compensation is required in the high-order harmonic phase delay compensation unit 601 and in the two-phase / three-phase conversion unit 204 and the PWM phase delay compensator 209.

[0050] Figure 7 Indicates in Figure 6 In the processing of the controller 104 shown in this invention, the waveforms (simulated waveforms) of the U-phase current iu and the q-axis higher harmonic voltage Vqhrm are superimposed only on the q-axis when the higher harmonic voltage is present. Simulation conditions and Figure 5 The same as shown.

[0051] At either 300 rpm or 3000 rpm, the q-axis higher harmonic voltage controlled by this invention showed no delay relative to the ideal state. Therefore, it can be seen that although the U-phase current has a difference caused by discretization error, it does not produce a large difference between the ideal state and the control of this invention.

[0052] As described above, the motor control device of the present invention includes: a high-order harmonic voltage calculation unit that calculates a high-order harmonic voltage added to a fundamental voltage command value in a two-phase rotating coordinate system synchronized with the rotor phase; and a high-order harmonic phase delay compensation unit that compensates for the phase delay during the calculation of the high-order harmonic voltage calculation unit based on the high-order harmonic number of the high-order harmonic voltage and the electrical angular frequency of the fundamental voltage. Therefore, it is possible to superimpose the out-of-phase high-order harmonic voltage superimposed in the controller to suppress noise and torque pulsation generated from the motor without phase delay.

[0053] Furthermore, the present invention is not limited to the above embodiments, but also includes various modifications. For example, the above embodiments are detailed descriptions provided for ease of understanding of the present invention, and are not necessarily limited to having all the described configurations. Symbol Explanation

[0054] 101: Inverter, 102: Motor, 103: Battery, 104: Controller, 105-107: Three-phase winding, 108: Magnetic pole position sensor, 109: Rotor phase, 110: Current sensor, 111: U-phase current sensor signal, 112: V-phase current sensor signal, 113: W-phase current sensor signal, 114-119: Switching element, 120-125: IGBT, 126-131: Gate drive signal, 132: Gate signal, 133: High-potential DC wiring, 134: Low-potential DC wiring, 135: Gate drive circuit, 136: voltage sensor, 137: DC voltage sensor signal, 138: smoothing capacitor, 139: torque command, 201: current command calculation unit, 202: three-phase to two-phase conversion unit, 203: current control unit, 204: two-phase to three-phase conversion unit, 205: speed calculation unit, 206: PWM gate control signal generation unit (control signal generation unit), 207: deviation calculator, 208: deviation calculator, 209: PWM phase delay compensator, 211: high-order harmonic overlapper, 601: high-order harmonic phase delay compensation unit.

Claims

1. A control device for an electric motor, characterized in that, have: The higher harmonic voltage calculation unit calculates the higher harmonic voltage, which is added to the fundamental voltage command value, in a two-phase rotating coordinate system synchronized with the rotor phase; and The higher harmonic phase delay compensation unit compensates for the phase delay during the calculation of the higher harmonic voltage calculation unit based on the higher harmonic order of the higher harmonic voltage and the electrical angular frequency of the fundamental voltage in the two-phase rotating coordinate system. The higher harmonic voltage is added to the fundamental voltage command value in the two-phase rotating coordinate system and then output to the two-phase to three-phase conversion unit.

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