Dual active bridge series resonant converter zero-current-flow power prediction control method and system
By adjusting the phase difference between the primary and secondary bridge ports of the dual active bridge series resonant converter using model predictive control, the problems of high return current power and slow dynamic response are solved, achieving zero return current power control and improving converter efficiency and dynamic performance.
Patent Information
- Authority / Receiving Office
- CN · China
- Patent Type
- Patents(China)
- Current Assignee / Owner
- SHANDONG UNIV
- Filing Date
- 2023-07-07
- Publication Date
- 2026-07-14
Smart Images
Figure CN116827136B_ABST
Abstract
Description
Technical Field
[0001] This invention belongs to the field of converter optimization control technology, specifically relating to a zero-return power prediction control method and system for a dual active bridge series resonant converter. Background Technology
[0002] The statements in this section are merely background information related to the present invention and do not necessarily constitute prior art.
[0003] Dual Bridge Series Resonant Converters (DBSRCs) offer advantages such as higher inductor current sinusoidality and lower switching transistor turn-off current, making them widely used in various energy storage device interfaces. However, under traditional single-phase shift (SPS) modulation, DBSRCs inevitably generate backflow power on both the primary and secondary sides. This requires a larger input voltage or current to transmit the same power, causing the resonant cavity to withstand greater voltage and current stress, increasing system switching losses, and reducing converter efficiency.
[0004] Traditional PI control, when applied to DBSRC, suffers from long system settling times due to the presence of the integral term. Furthermore, for multi-input multi-output systems, the coupling between control loops makes PI parameters difficult to tune. Model predictive control, with its advantages of multi-objective control, fast dynamic response, and strong robustness, is widely used in converter control and has already been applied to DBSRC.
[0005] According to the inventors, in terms of return power suppression, an Extended Phase Shift (EPS) strategy combined with frequency conversion modulation can be used. By adjusting the relationship between the zero-crossing point of the resonant current and the phase of the voltage at the primary and secondary H-bridge ports, zero return power control under arbitrary transmission power is achieved. This method only involves two control degrees of freedom, and the transmission power control uses frequency conversion modulation. Regarding model predictive control, the predictive model of DBSRC under single-phase-shift control can be derived, but it does not involve solving the predictive model of multi-phase-shift control under efficiency optimization. Summary of the Invention
[0006] To address the aforementioned issues, this invention proposes a zero-return power predictive control method and system for a dual active bridge series resonant converter (DBSRC). By introducing Model Predictive Control (MPC), the dynamic response speed of the DBSRC is accelerated, solving the problem of high return power in the DBSRC under non-unity voltage gain and achieving a fast dynamic response of the system.
[0007] According to some embodiments, the first aspect of the present invention provides a zero-return-current power prediction control method for a dual active bridge series resonant converter, employing the following technical solution:
[0008] A zero-return-current power prediction control method for a dual active bridge series resonant converter includes:
[0009] Obtain the state parameters of the dual active bridge series resonant converter;
[0010] Based on the obtained state parameters and the multi-phase shift model predictive control model constructed based on the fundamental wave analysis method, the phase difference of the fundamental wave components of the primary and secondary bridge port voltages of the dual active bridge series resonant converter is obtained.
[0011] The phase difference of the fundamental voltage component is subjected to zero-return power control under multiple phase shifts to obtain the optimized phase shift angle;
[0012] Phase-shift modulation of the dual active bridge series resonant converter is performed based on the obtained optimized phase-shift angle to achieve zero-return power prediction control of the dual active bridge series resonant converter.
[0013] As a further technical limitation, the acquired state parameters include at least the input voltage, output voltage, and output current of the dual active bridge series resonant converter.
[0014] As a further technical limitation, in the process of constructing the multi-phase model predictive control model, the obtained state variables are discretized by the forward Euler method to obtain the predicted value of the state variables at the next moment. Based on the obtained predicted value, a cost function is constructed to calculate the phase difference of the fundamental components of the primary and secondary bridge voltages.
[0015] Furthermore, in the process of calculating the phase difference of the fundamental components of the primary and secondary bridge voltages, the constructed cost function is differentiated. When the derivative of the cost function is zero, the outward phase shift angle is obtained. The obtained outward phase shift angle is then transformed to obtain the phase difference of the fundamental components of the primary and secondary bridge voltages.
[0016] Furthermore, in the zero-return power control process under multiple phase shifts, when the voltage gain is less than the preset value, the outer phase shift angle controls the output voltage to achieve the given value of total transmission power; the inner phase shift angle of the primary side is adjusted so that the voltage at the primary and secondary bridge ports rises in the same phase.
[0017] Furthermore, the inner phase shift angle of the secondary side is adjusted to achieve the zero-crossing point of the resonant current being in phase with the edge of the bridge voltage. Based on the resonant current expression and the relationship between the inner and outer phase shift angles, the phase difference of the fundamental wave is replaced by the outer phase shift angle to obtain the optimized phase shift angle.
[0018] Furthermore, in the zero-return power control process under multiple phase shifts, when the voltage gain is greater than the preset value, the phase shift angle on the secondary side is negative. Based on the relationship between the fundamental phase difference, the optimized phase shift angle is obtained.
[0019] According to some embodiments, the second aspect of the present invention provides a zero-return-current power prediction control system for a dual active bridge series resonant converter, employing the following technical solution:
[0020] A zero-return-current power predictive control system for a dual active bridge series resonant converter includes:
[0021] The acquisition module is configured to acquire the state parameters of the dual active bridge series resonant converter;
[0022] The control module is configured to obtain the phase difference of the fundamental components of the primary and secondary bridge voltages of the dual active bridge series resonant converter by constructing a multi-phase-shift predictive control model based on the acquired state parameters and the fundamental analysis method; to perform zero-return power control under multiple phase shifts on the obtained fundamental voltage component phase difference to obtain the optimized phase shift angle; and to perform phase-shift modulation of the dual active bridge series resonant converter based on the obtained optimized phase shift angle to realize zero-return power predictive control of the dual active bridge series resonant converter.
[0023] According to some embodiments, a third aspect of the present invention provides a computer-readable storage medium, employing the following technical solution:
[0024] A computer-readable storage medium having a program stored thereon, which, when executed by a processor, implements the steps in the zero-return power prediction control method for a dual active bridge series resonant converter as described in the first aspect of the present invention.
[0025] According to some embodiments, the fourth aspect of the present invention provides an electronic device, which adopts the following technical solution:
[0026] An electronic device includes a memory, a processor, and a program stored in the memory and executable on the processor. When the processor executes the program, it implements the steps in the zero-return power prediction control method for a dual active bridge series resonant converter as described in the first aspect of the present invention.
[0027] Compared with the prior art, the beneficial effects of the present invention are as follows:
[0028] This invention addresses the challenges of magnetic component design and slow dynamic response caused by frequency conversion modulation in existing control strategies. It proposes a zero-return-current power control strategy based on model predictive control. By adjusting the primary-side phase shift angle to ensure the rising edges of the primary and secondary bridge voltages are in phase, and by adjusting the secondary-side phase shift angle to ensure the zero-crossing point of the resonant current is in phase with the bridge voltage edge, both the primary and secondary return currents are zero. This reduces the voltage and current stress on the resonant cavity when transmitting the same power, lowers system switching losses, and improves converter efficiency. Furthermore, this invention introduces model predictive control to establish an output voltage cost function, enabling a fast dynamic response for the system. Attached Figure Description
[0029] The accompanying drawings, which form part of this embodiment, are used to provide a further understanding of this embodiment. The illustrative embodiments and their descriptions are used to explain this embodiment and do not constitute an improper limitation of this embodiment.
[0030] Figure 1 This is a schematic diagram of the topology of the dual active bridge series resonant converter in Embodiment 1 of the present invention;
[0031] Figure 2 This is a waveform diagram of the dual active bridge series resonant converter under triple phase-shift control in Embodiment 1 of the present invention.
[0032] Figure 3 This is a waveform diagram of the dual active bridge series resonant converter under zero return power control in Embodiment 1 of the present invention.
[0033] Figure 4 This is a control flowchart of the zero-return power prediction control method for a dual active bridge series resonant converter in Embodiment 1 of the present invention. Detailed Implementation
[0034] The present invention will be further described below with reference to the accompanying drawings and embodiments.
[0035] It should be noted that the following detailed descriptions are exemplary and intended to provide further explanation of this application. Unless otherwise specified, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application pertains.
[0036] It should be noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit the scope of exemplary embodiments according to the invention. As used herein, the singular form is intended to include the plural form as well, unless the context clearly indicates otherwise. Furthermore, it should be understood that when the terms "comprising" and / or "including" are used in this specification, they indicate the presence of features, steps, operations, devices, components, and / or combinations thereof.
[0037] Where there is no conflict, the embodiments and features in the embodiments of the present invention can be combined with each other.
[0038] Example 1
[0039] Embodiment 1 of this invention introduces a zero-return power prediction control method for a dual active bridge series resonant converter.
[0040] Regarding return power suppression, the existing technology has the following drawbacks:
[0041] (1) Zero backflow power control across the entire load range is required through frequency conversion modulation, which makes the design of magnetic components such as resonant inductors and high-frequency transformers in the converter difficult.
[0042] (2) Output voltage is stabilized by a PI regulator, but the dynamic response speed is slow.
[0043] To address this, this embodiment proposes a zero-return power control strategy based on predictive control, specifically including: modeling the DBSRC triple phase shift control (TPS) using Fundamental Harmonic Analysis (FHA) to ignore higher harmonic components in the resonant current to reduce computational complexity; replacing the original "extended phase shift + frequency modulation" scheme with triple phase shift control to reduce the design difficulty of magnetic components in the circuit; and improving the dynamic performance of the system by introducing model predictive control, thus solving the problem of long system settling time under traditional PI control.
[0044] A zero-return-current power prediction control method for a dual active bridge series resonant converter includes:
[0045] Obtain the state parameters of the dual active bridge series resonant converter;
[0046] Based on the obtained state parameters and the multi-phase shift model predictive control model constructed based on the fundamental wave analysis method, the phase difference of the fundamental wave components of the primary and secondary bridge port voltages of the dual active bridge series resonant converter is obtained.
[0047] The phase difference of the fundamental voltage component is subjected to zero-return power control under multiple phase shifts to obtain the optimized phase shift angle;
[0048] Phase-shift modulation of the dual active bridge series resonant converter is performed based on the obtained optimized phase-shift angle to achieve zero-return power prediction control of the dual active bridge series resonant converter.
[0049] like Figure 1 The DBSRC topology shown includes an input-side support capacitor C1, an output-side support capacitor C2, and a resonant capacitor L. r Resonant inductor C rThe high-frequency transformer T, switching transistors S1 to S8, and their anti-parallel diodes are also present. The dual active bridge series resonant converter contains four bridge arms. In each bridge arm, the upper and lower two switching transistors are complementary and conduct. The driving signal for the complementary switching transistors is a square wave signal with a 50% duty cycle, that is, each switching transistor conducts for half a switching cycle.
[0050] The DBSRC working waveform under TPS is as follows: Figure 2 As shown, let the phase shift angle of the primary side be α1, the phase shift angle of the secondary side be α2, and the phase shift angle of the outer side be... Taking the rising edge of the voltage at the midpoint of the primary arm as the zero point of timing, the fundamental component phase of the voltage at the midpoint of the primary arm is zero, and the fundamental component phase of the potential difference at the midpoint of the secondary arm is zero.
[0051]
[0052] Perform Fourier decomposition on the midpoint voltage of the primary and secondary bridge arms and extract its fundamental component:
[0053]
[0054] In formula (1), ω s v is the switching angular frequency. AB V' represents the fundamental component of the voltage at the midpoint of the primary side bridge arm. CD.1 Let the fundamental component of the voltage at the midpoint of the secondary bridge arm be represented. Then the resonant current can be expressed as:
[0055]
[0056] Where M represents the voltage gain, with a value of nV2 / V1. Ignoring losses during power transmission caused by conductor impedance, switching transistor on-resistance, and parasitic resistance of supporting capacitors, taking the secondary bridge port voltage as an example, the converter's transmitted power is:
[0057]
[0058] When M < 1, the DBSRC operating waveform diagram is as follows: Figure 3 As shown in (a), the output voltage is controlled by an external phase shift angle to achieve a given total transmission power. The rising edges of the primary and secondary bridge port voltages are made in phase by adjusting the internal phase shift angle of the primary side. The zero-crossing point of the resonant current is made in phase with the edge of the bridge port voltage by adjusting the internal phase shift angle of the secondary side. At this time, the primary and secondary bridge port voltages and the resonant current are always in the same direction, and no frequency conversion control is used, resulting in a simple circuit design.
[0059] The expression for the resonant current has been given in equation (2). To achieve the above-mentioned relationship between the inner and outer phase shift angles, equation (2) should satisfy:
[0060]
[0061] in
[0062] In solving equation (4), to facilitate controller operation, the fundamental phase difference β will be replaced by an outward phase shift angle. Solving equation (4) yields the optimized phase shift angle expression:
[0063]
[0064] When M>1, the secondary side inward phase shift angle is negative, and the working waveform is as follows: Figure 3 As shown in (b), the fundamental phase difference equation in equation (4) becomes: The optimized phase shift angle expression is:
[0065]
[0066] like Figure 1 As shown, according to Kirchhoff's current law, the instantaneous value i of the output supporting capacitor current... f (t) can be expressed as:
[0067] i f (t)=i2(t)-i o (t) (7)
[0068] Capacitor current i f i(t) can be represented by the differential term of the capacitor voltage, and i2(t) can be represented by the system output power, that is:
[0069]
[0070] Where C o For the output support capacitor of DBSRC; u o (t) represents the output voltage; u in (t) represents the output voltage; n represents the turns ratio of the high-frequency transformer; X LC α1 / 2 and α2 / 2 are the phase shift angles of the primary and secondary sides of the DBSRC, respectively; β is the phase difference between the fundamental components of the primary and secondary bridge voltages; i o (t) represents the load current.
[0071] To obtain the prediction model for DBSRC, the output voltage differential term is discretized using the forward Euler method:
[0072]
[0073] Where k represents the working time, T s For work cycles.
[0074] Substituting equation (9) into equation (8), we obtain the expression for the predicted output voltage at time k+1:
[0075]
[0076] To ensure that the output voltage reaches a given value as quickly as possible and remains stable, a cost function can be established.
[0077]
[0078] The smaller the cost function, the smaller the deviation between the output supporting capacitor voltage and the given voltage at the next sampling time. Therefore, the desired control quantity should minimize the cost function, i.e., J(t) k The derivative is zero. Substituting equation (10) into equation (11) and differentiating it, setting the derivative to zero, we obtain the expression for the outward phase angle:
[0079]
[0080] like Figure 4 As shown, the specific implementation method of DBSRC zero-return power control is as follows:
[0081] Step S01: The sensor inputs the DBSRC voltage u at time k. in (t k Output voltage u o (t k and output current i o (t k Sampling is performed to set the output voltage setpoint.
[0082] Step S02: Combine the inner phase shift angles α1 and α2 obtained in step S04 of the previous working cycle at time k-1, and the input voltage u obtained in step S01 of this working cycle. in (t k Output voltage u o (t k and output current i o (t k Substituting this into the predictive controller of the DBSRC model, i.e., equation (12), the outward phase shift angle is obtained.
[0083] Step S03: Shift the phase angle outwards This is converted to the phase difference β of the fundamental component of the primary and secondary bridge port voltages;
[0084] Step S04: Calculate the voltage gain M based on the transformer turns ratio, i.e. the input-output voltage ratio. Substitute the voltage gain M and the fundamental phase difference β into equation (5) or equation (6) to calculate α1 and α2, and temporarily store them for use in the next working cycle step S02.
[0085] Step S05: Calculate the obtained phase shifts α1, α2, and α3. The input is fed to the phase-shift modulator for phase-shift modulation, which generates control signals for each switch transistor, thereby controlling the converter to achieve zero-return power control.
[0086] By repeating the above steps, zero-return-current power predictive control of the dual active full-bridge series resonant converter can be achieved. This control strategy effectively improves the dynamic performance of the system, achieving zero return-current power within one duty cycle, thus increasing the converter efficiency.
[0087] This embodiment addresses the challenges of magnetic component design and slow dynamic response caused by frequency conversion modulation in existing control strategies. It proposes a zero-return-current power control strategy based on model predictive control. By adjusting the primary-side phase shift angle to ensure the rising edges of the primary and secondary bridge port voltages are in phase, and by adjusting the secondary-side phase shift angle to ensure the zero-crossing point of the resonant current is in phase with the bridge port voltage edge, both the primary and secondary return currents are zero. This reduces the voltage and current stress on the resonant cavity when transmitting the same power, lowers the system's switching losses, and improves converter efficiency. Furthermore, by introducing model predictive control and establishing an output voltage cost function, a fast dynamic response of the system is achieved.
[0088] Example 2
[0089] Embodiment 2 of the present invention introduces a zero-return-current power prediction and control system for a dual active bridge series resonant converter.
[0090] A zero-return-current power predictive control system for a dual active bridge series resonant converter includes:
[0091] The acquisition module is configured to acquire the state parameters of the dual active bridge series resonant converter;
[0092] The control module is configured to obtain the phase difference of the fundamental components of the primary and secondary bridge voltages of the dual active bridge series resonant converter by constructing a multi-phase-shift predictive control model based on the acquired state parameters and the fundamental analysis method; to perform zero-return power control under multiple phase shifts on the obtained fundamental voltage component phase difference to obtain the optimized phase shift angle; and to perform phase-shift modulation of the dual active bridge series resonant converter based on the obtained optimized phase shift angle to realize zero-return power predictive control of the dual active bridge series resonant converter.
[0093] The detailed steps are the same as those of the zero-return power prediction control method for the dual active bridge series resonant converter provided in Example 1, and will not be repeated here.
[0094] Example 3
[0095] Embodiment 3 of the present invention provides a computer-readable storage medium.
[0096] A computer-readable storage medium having a program stored thereon, which, when executed by a processor, implements the steps in the zero-return power prediction control method for a dual active bridge series resonant converter as described in Embodiment 1 of the present invention.
[0097] The detailed steps are the same as those of the zero-return power prediction control method for the dual active bridge series resonant converter provided in Example 1, and will not be repeated here.
[0098] Example 4
[0099] Embodiment 4 of the present invention provides an electronic device.
[0100] An electronic device includes a memory, a processor, and a program stored in the memory and executable on the processor. When the processor executes the program, it implements the steps in the zero-return power prediction control method for a dual active bridge series resonant converter as described in Embodiment 1 of the present invention.
[0101] The detailed steps are the same as those of the zero-return power prediction control method for the dual active bridge series resonant converter provided in Example 1, and will not be repeated here.
[0102] The above description is merely a preferred embodiment of this practice and is not intended to limit the scope of this practice. Various modifications and variations can be made to this practice by those skilled in the art. Any modifications, equivalent substitutions, or improvements made within the spirit and principles of this practice should be included within the protection scope of this practice.
Claims
1. A zero-return-current power prediction and control method for a dual active bridge series resonant converter, characterized in that, include: Obtain the state parameters of the dual active bridge series resonant converter; Based on the obtained state parameters and the multi-phase shift model predictive control model constructed based on the fundamental wave analysis method, the phase difference of the fundamental wave components of the primary and secondary bridge port voltages of the dual active bridge series resonant converter is obtained. The phase difference of the fundamental voltage component is subjected to zero-return power control under multiple phase shifts to obtain the optimized phase shift angle; Phase-shift modulation of the dual active bridge series resonant converter is performed based on the obtained optimized phase-shift angle to achieve zero-return power prediction control of the dual active bridge series resonant converter; In the process of constructing the multi-phase model predictive control model, the obtained state variables are discretized by the forward Euler method to obtain the predicted value of the state variables at the next moment. Based on the obtained predicted value, a cost function is constructed to calculate the phase difference of the fundamental components of the primary and secondary bridge voltages. In the process of calculating the phase difference of the fundamental components of the primary and secondary bridge voltages, the constructed cost function is differentiated. When the derivative of the cost function is zero, the outward phase shift angle is obtained. The obtained outward phase angle is converted to obtain the phase difference of the fundamental component of the primary and secondary bridge voltages; In the zero-return power control process under multiple phase shifts, when the voltage gain is less than the preset value, the output voltage is controlled by the external phase shift angle to achieve the given value of total transmission power. Adjust the phase shift angle of the primary side so that the voltage at the bridge ports of the primary and secondary sides rises in the same phase. Adjust the inner phase shift angle of the secondary side to achieve the zero-crossing point of the resonant current in phase with the edge of the bridge voltage; based on the resonant current expression and the relationship between the inner and outer phase shift angles, replace the fundamental phase difference with the outer phase shift angle to obtain the optimized phase shift angle.
2. The zero-return-current power prediction control method for a dual active bridge series resonant converter as described in claim 1, characterized in that, The acquired state parameters include at least the input voltage, output voltage, and output current of the dual active bridge series resonant converter.
3. The zero-return-current power prediction control method for a dual active bridge series resonant converter as described in claim 1, characterized in that, In the zero-return power control process under multiple phase shifts, when the voltage gain is greater than the preset value, the phase shift angle on the secondary side is negative. Based on the relationship between the fundamental phase difference, the optimized phase shift angle is obtained.
4. A zero-return-current power prediction and control system for a dual active bridge series resonant converter, employing the zero-return-current power prediction and control method for a dual active bridge series resonant converter as described in any one of claims 1-3, characterized in that, include: The acquisition module is configured to acquire the state parameters of the dual active bridge series resonant converter; The control module is configured to obtain the phase difference of the fundamental components of the primary and secondary bridge port voltages of the dual active bridge series resonant converter by constructing a multi-phase shift model predictive control model based on the acquired state parameters and the fundamental analysis method. The obtained voltage fundamental component phase difference is subjected to zero-return power control under multiple phase shifts to obtain the optimized phase shift angle; the phase shift modulation of the dual active bridge series resonant converter is performed based on the obtained optimized phase shift angle to realize zero-return power prediction control of the dual active bridge series resonant converter.
5. A computer-readable storage medium having a computer program stored thereon, characterized in that, When the program is executed by the processor, it implements the steps of the zero-return power prediction control method for a dual active bridge series resonant converter as described in any one of claims 1-3.
6. An electronic device comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, characterized in that, When the processor executes the program, it implements the steps of the zero-return power prediction control method for a dual active bridge series resonant converter as described in any one of claims 1-3.