Power conversion device

Abstract

The power conversion device of the present application includes: a power converter that outputs a signal that makes the output frequency, output voltage, and output current of a motor variable to the motor; and a control section that controls the power converter, the control section calculating a first electric power from the output voltage and the output current, and calculating a second electric power from the output current, a circuit parameter, and a frequency estimation value, calculating a first phase error estimation value in a first frequency range in such a manner that the first electric power follows the second electric power, calculating a second phase error estimation value in a second frequency range that is different from the first frequency range, and controlling the frequency estimation value in such a manner that the first phase error estimation value or the second phase error estimation value follows a command value of the phase error estimation value.

Description

Technical Field

[0001] This invention relates to power conversion devices. Background Technology

[0002] In the low-speed range without position sensor control, as a stable and high-precision control method, there is a technique, as described in Patent Document 1, that calculates the frequency of the magnet motor by calculating the reactive power based on the voltage command value, current detection value, circuit parameters and frequency estimation value of the power converter.

[0003] Existing technical documents

[0004] Patent documents

[0005] Patent Document 1: Japanese Patent Application Publication No. 2006-197712 Summary of the Invention

[0006] The problem that the invention aims to solve

[0007] The technology described in Patent Document 1 calculates the inverter's frequency estimate by calculating two reactive power values ​​(Q and Qhat) to make their deviation ΔQ zero. The frequency estimate reduces sensitivity to temperature changes in the winding resistance of the magnet motor, thus enabling high-precision control characteristics.

[0008] However, there are situations where different speed control techniques are used for higher frequencies (medium-high speed) and lower frequencies (low speed). In such cases, for example, in the low-speed range, the technique of Patent Document 1 is used to calculate the two reactive power values ​​so that their deviation ΔQ is zero to estimate the frequency of the first motor. On the other hand, for the medium-high speed range, for example, another technique is used to estimate the phase error (the phase of the control and the phase of the magnet motor) based on the extended back electromotive force, and the frequency of the second motor is estimated using PI control to achieve zero deviation. When switching between the frequencies of the first and second motors in the low-speed and medium-high speed ranges, it is assumed that a torque surge (torque fluctuation) will occur due to the difference between the two frequencies.

[0009] Furthermore, by reducing sensitivity to temperature changes in the winding resistance of the magnet motor, a stable and high-precision control characteristic is required that does not require adjustment of circuit parameters such as resistance values.

[0010] The purpose of this invention is to provide a power conversion device with stable and high-precision control characteristics that can prevent torque shock when the frequency of the motor changes and does not require adjustment of circuit parameters.

[0011] Technical solutions for solving the problem

[0012] This invention is a power conversion device, comprising:

[0013] A power converter that outputs a signal to the motor that allows for variable output frequency, output voltage, and output current; and

[0014] The control unit that controls the power converter

[0015] The control unit,

[0016] A first electrical power is calculated from the output voltage and the output current, and a second electrical power is calculated from the output current, circuit parameters, and frequency-estimated values. The first phase error is calculated within a first frequency range in such a manner that the first electrical power follows the second electrical power.

[0017] The second phase error estimate is calculated in a second frequency range that is different from the first frequency range.

[0018] The frequency calculation value is controlled in such a way that the first phase error calculation value or the second phase error calculation value follows the command value of the phase error calculation value.

[0019] Invention Effects

[0020] According to the present invention, stable and high-precision control characteristics can be achieved that prevent torque shocks when the frequency of the motor changes and do not require adjustment of circuit parameters. Attached Figure Description

[0021] Figure 1 This is a system structure diagram of the power conversion device, etc., in Example 1.

[0022] Figure 2 This is a structural diagram of the calculation unit for calculating phase error using extended back electromotive force in the high-speed range.

[0023] Figure 3 This is a structural diagram of the phase error calculation unit in the low-speed range of Example 1.

[0024] Figure 4 This is a structural diagram of the frequency and phase calculation unit of Embodiment 1.

[0025] Figure 5 This is a diagram showing the control characteristics when the extended back EMF method in the medium-to-high speed range is applied to the low-speed range.

[0026] Figure 6 This is a diagram illustrating the control characteristics of Example 1.

[0027] Figure 7 This is a graph showing the control characteristics when switching between low-speed and medium-high-speed ranges in Example 1.

[0028] Figure 8 This is a structural diagram used to confirm the manifestation of the present invention.

[0029] Figure 9 This is a system structure diagram of the power conversion device, etc., in Example 2.

[0030] Figure 10 This is a structural diagram of the phase error calculation unit in the low-speed range of Example 2.

[0031] Figure 11 This is a system structure diagram of the power conversion device, etc., in Example 3.

[0032] Figure 12 This is a structural diagram of the phase error calculation unit in the medium-to-high speed range of Example 3.

[0033] Figure 13 This is a system structure diagram of the power conversion device, etc., in Example 4.

[0034] Figure 14 This is a structural diagram of the phase error calculation unit in the medium-to-high speed range of Example 4.

[0035] Figure 15 This is a system structure diagram of the power conversion device, etc., in Example 5.

[0036] Figure 16 This is a system structure diagram of the power conversion device, etc., in Example 6. Detailed Implementation

[0037] The present embodiment will now be described in detail with reference to the accompanying drawings. Common structures in all the drawings are given the same reference numerals. Furthermore, the embodiments described below are not limited to the examples shown in the drawings.

[0038] Example 1

[0039] Figure 1 This is a system structure diagram of Embodiment 1, which includes a power conversion device and a magnet motor.

[0040] In this embodiment, the power conversion device achieves stable and high-precision control characteristics in a low-speed range of approximately 10% of the base frequency, in a sensorless control system that omits encoders or similar devices for detecting the phase of the magnet motor.

[0041] The output of the magnet motor 1 is the motor torque obtained by combining the torque component caused by the magnetic flux of the permanent magnet and the torque component caused by the inductance of the armature winding.

[0042] Power converter 2 has a semiconductor element as a switching element. Power converter 2 receives a three-phase AC voltage command value v.u * v v * v w * Output voltage command value v for three-phase AC. u * v v * v w * The voltage value is proportional to the output of the power converter 2. Based on the output of the power converter 2, the magnet motor 1 is driven, and the output voltage, frequency, and current of the magnet motor 1 can be variably controlled. An IGBT (Insulated Gate Bipolar Transistor) can be used as the switching element.

[0043] DC power supply 3 provides DC voltage and DC current to power converter 2.

[0044] Current detector 4 outputs the three-phase AC current i of magnet motor 1 u i v i w The detection value is i uc i vc i wc Additionally, the current detector 4 can also detect the alternating current of two phases of the three-phase magnet motor 1, such as phase u and phase w, according to the alternating current conditions (i... u +i v +i w =0), press i v =-(i u +i w Find the alternating current of phase v.

[0045] In this embodiment, an example is shown where the current detector 4 is installed inside the power conversion device, but it can also be installed outside the power conversion device.

[0046] The control unit includes, as described below, a coordinate transformation unit 5, a speed control calculation unit 6, a vector control calculation unit 7, a phase error calculation calculation unit 8 for the medium-to-high speed range, a phase error calculation calculation unit 9 for the low-speed range, a frequency and phase calculation calculation unit 10, and a coordinate transformation unit 11. Furthermore, the control unit controls the output of the power converter 2 to variably control the output voltage, output frequency, and output current of the magnet motor 1.

[0047] The control unit is composed of semiconductor integrated circuits (arithmetic control units) such as microcomputers and DSPs (Digital Signal Processors). The control unit can be partially or entirely constructed using hardware such as ASICs (Application Specific Integrated Circuits) and FPGAs (Field Programmable Gate Arrays). The CPU (Central Processing Unit) of the control unit reads programs stored in recording devices such as memory and executes the processing of the coordinate transformation unit 5 and other units mentioned above.

[0048] Next, the components of the control unit will be explained.

[0049] Coordinate transformation unit 5 is based on the three-phase alternating current i u i v i w The detection value i uc i vc i wc and phase calculation value θ dc Output the current detection values ​​i of the d-axis and q-axis dc i qc .

[0050] Speed ​​control arithmetic unit 6 is based on frequency command value ω r * and frequency-calculated value ω dc Calculate the torque command value τ * The current command value i of the q-axis is output by dividing by the torque coefficient. q * Frequency command value ω r * Used for determining the medium-to-high speed range and the low-speed range, frequency estimation value ω dc The calculated speed corresponding to the motor (the calculated rotational speed of the motor).

[0051] The vector control arithmetic unit 7 outputs current command values ​​i based on the d-axis and q-axis. d * i q * Current detection value i dc i qc Frequency calculation value ω dc The voltage command values ​​v of the d-axis and q-axis calculated from the circuit parameters of magnet motor 1 dc ** v qc ** .

[0052] The phase error calculation unit 8 in the medium-to-high speed range uses the control axis, i.e., d. c axis and q c The voltage command value v of the shaft dc ** v qc ** Frequency calculation value ω dc Current detection value i dc i qc The circuit parameters of magnet motor 1, and the control phase θ in the medium-to-high speed range of the output. dc Phase θ of the magnet of magnet motor 1 d The deviation, i.e., the calculated value of phase error Δθ. c_H .

[0053] The phase error calculation unit 9 in the low-speed range uses the control axis, i.e., d. c axis and q c The voltage command value v of the shaft dc ** v qc ** Frequency calculation value ω dc Current detection value i dc i qc The circuit parameters of magnet motor 1 are used to calculate the control phase θ in the low-speed range. dc Phase θ of the magnet of magnet motor 1 d The deviation, i.e., the calculated value of phase error Δθ. c_L .

[0054] The frequency and phase calculation unit 10 calculates the phase error Δθ in the low-speed range. c_L Or the calculated value of phase error Δθ in the medium to high speed range c_H Calculated output frequency ω dc and phase calculation value θ dc .

[0055] Coordinate transformation unit 11 according to d c axis and q c The voltage command value v of the shaft dc ** v qc ** and the phase calculation value θ dc To output the three-phase AC voltage command value v u * v v * v w * .

[0056] First, the basic operation of the sensorless vector control method when using the phase error calculation unit 9 in the low-speed range will be explained.

[0057] Speed ​​control calculation unit 6 calculates the frequency value ω. dc Follow frequency command value ω r * In this manner, the torque command τ is calculated according to (Equation 1) using proportional control and integral control. * and the current command value i on the q-axis q * .

[0058] [Mathematical Expression 1]

[0059]

[0060] Here, K sp : Proportional gain for speed control, K si Integral gain for speed control, P m Extreme logarithm, K e Inductive voltage coefficient, L d d-axis inductance, L q : q-axis inductance, *: set value, s is the Laplace operator.

[0061] The vector control calculation unit 7 first uses the set value R of the winding resistance, which is a circuit parameter of the permanent magnet motor 1. * The setting value L of the d-axis inductance d * The setting value L of the q-axis inductance q * The value of the induced voltage coefficient K e * d c axis and q c Current command value i of the shaft d * i q * and frequency-calculated value ω dc Output d according to (Equation 2) c axis and q c The voltage reference value v of the shaft dc * v qc * .

[0062] [Mathematical Expression 2]

[0063]

[0064] Here, T acr : The response time constant of current control.

[0065] The vector control arithmetic unit 7 then uses the current detection values ​​i of each component to... dc i qc Follow d c axis and q c Current command value i of the shaft d * i q * In this manner, through proportional control and integral control, d is calculated according to (Equation 3). c axis and q c Voltage correction value Δv of the shaft dc Δv qc .

[0066] [Mathematical Expression 3]

[0067]

[0068] Here, K pd :d c The proportional gain of the axis controlled by current, K id :d c The integral gain of the shaft current control, K pq :q c The proportional gain of the axis controlled by current, K iq :q c Integral gain of shaft current control

[0069] Furthermore, the vector control arithmetic unit 7 performs the operation d according to (equation 4). c axis and q c The voltage command value v of the shaft dc ** v qc ** .

[0070] [Mathematical Expression 4]

[0071]

[0072] exist Figure 2 The diagram shows a block diagram of the phase error estimation calculation unit 8 in the medium-to-high speed range. In the phase error estimation calculation unit 8 in the medium-to-high speed range, based on d... c axis and q c Shaft voltage command v dc ** v qc ** Current detection value i dc i qc Circuit parameters of magnet motor 1 (R) * L q *The calculation unit 81, which calculates the phase error in the medium-to-high speed range using the extended electromotive force method, calculates the calculated value Δθ of the phase error in the medium-to-high speed range according to (Equation 5). c_H .

[0073] [Mathematical Expression 5]

[0074]

[0075] Here, the phase error calculation unit 9 in the low-speed range will be explained. Figure 3 The diagram shows a block diagram of the phase error calculation unit 9 in the low-speed range.

[0076] In the low-speed range phase error calculation unit 9, the first reactive power calculation unit 91 uses d, which is the output voltage of the magnet motor 1. c The voltage command value v of the shaft dc ** and q c The voltage command value v of the shaft qc ** , and d as the output current of the magnet motor 1 c Shaft current detection value i dc q c Shaft current detection value i qc Calculate the first reactive power Q according to (Equation 6). c .

[0077] [Mathematical Expression 6]

[0078]

[0079] The second reactive power calculation unit 92 uses d c Shaft current detection value i dc , and q c Shaft current detection value i qc Frequency calculation value ω dc Circuit parameters of magnet motor 1 (L) d * L q * K e * ), calculate the second reactive power Q according to (Equation 7). c ^ .

[0080] [Mathematical Expression 7]

[0081]

[0082] Input the first reactive power Q to the subtraction unit 93c Second reactive power Q c ^ Calculate its deviation ΔQ c Reactive power deviation ΔQ c The instruction value 94, "0", is input to the PI control calculation unit 95 (hereinafter referred to as the PI control unit) to make it follow the deviation of reactive power. The PI control unit 95 performs P (proportional) + I (integral) control calculation and calculates the estimated value Δθ of the phase error Δθ in the low-speed range according to (Equation 8). c_L .

[0083] [Mathematical Expression 8]

[0084]

[0085] Here, K pθ : Proportional gain for phase error calculation, K iθ : Integral gain of phase error calculation.

[0086] The calculation unit 10 for frequency and phase is explained. Figure 4 The diagram shows a block diagram of the frequency and phase calculation unit 10.

[0087] The calculated value Δθ of the phase error in the low-speed range is input to the switching unit 101. c_L The calculated value of phase error Δθ in the medium and high speed range. c_H and frequency command value ω r * The switching unit 101 operates according to the frequency command ω. r * The magnitude of Δθ depends on the speed range; if it's in the low-speed range, then... c =Δθ c_L If it is in the medium to high speed range, then Δθ c =Δθ c_H The calculated value of phase error Δθ c Output.

[0088] Subtraction unit 103 aims to calculate the aforementioned phase error value Δθ. c Follow the command value Δθ of the phase error c * 102, and the calculated value Δθ of the phase error input to the PI control unit 104. c With Δθ c * The deviation between them. The PI control unit 104 calculates the value ω according to the calculation frequency of (Equation 9) through P (proportional) + I (integral) control calculation. dc Furthermore, in the I control arithmetic unit (I control unit) 105, the phase calculation value θ is calculated based on the output of the PI control unit 104 according to (equation 10).dc .

[0089] [Mathematical Expression 9]

[0090]

[0091] Here, Kp pll : Proportional gain controlled by PLL, Ki pll : Integral gain controlled by PLL.

[0092] [Mathematical Expression 10]

[0093]

[0094] Next, the principle behind the stable and high-precision control characteristics of this invention will be explained.

[0095] Figure 5 This indicates that the phase error calculation unit 9 in the low-speed range of this invention (using Δθ) is not used. c_H The graph shows the control characteristics under the condition of ( ). The frequency command value ω r * Set it to 2% of the base frequency. The d shown in (Equation 2) c axis and q c The voltage command value v of the shaft dc ** v qc ** The set value R of the resistance value R included in the calculation formula for the phase error in the medium-to-high speed range shown in (Equation 5) is... * When there is no error, it is set to (a)R * / R = 1 (baseline).

[0096] Set the resistance value R to R * When an error exists, it is set as (b)R * / R=0.5, in Figure 5 The simulation results for (a) and (b) are shown in the figure.

[0097] Figure 5 In the middle, the upper part represents the load torque T. L The middle part represents the frequency command ω. r * and motor frequency ω r The lower part represents the phase error Δθ. Starting from point A in the figure, a ramp-shaped load torque is applied, which changes to 100% of the value at point B, and the applied load torque is maintained to the right of point B.

[0098] Let it be (a)R * With R = 1 (reference) set, the phase error Δθ is always zero, and the motor frequency ω rWith frequency command ω r * Consistent. Set as (b)R * When / R=0.5, the phase error Δθ increases in the negative direction, and the motor frequency ω r Stalled near zero, magnet motor 1 lost synchronization.

[0099] In this embodiment, d is used. c axis and q c The voltage command value v of the shaft dc ** v qc ** and current detection value i dc i qc Calculate the first reactive power Q according to (Equation 6). c Additionally, using d c axis and q c Shaft current detection value i dc i qc Frequency calculation value ω dc The circuit parameter settings (L) of magnet motor 1 d * L q * K e * ), calculate the second reactive power Q according to (Equation 7). c ^ .

[0100] So that the result of the operation Q c ^ With Q c The deviation follows zero, and the calculated value Δθ of the phase error in the low-speed range is automatically adjusted according to (Equation 8). c_L The calculated value Δθ c_L The calculation unit 10 for frequency and phase can improve control characteristics by reducing sensitivity to resistance values.

[0101] exist Figure 6 The control characteristics in the low-speed range of this embodiment are shown. R is set to... * When / R=0.5, the phase error calculation unit 9 in the low-speed range and the frequency and phase calculation unit 10 are activated to apply the... Figure 5 The same load torque. Because the calculated phase error Δθ is based on reactive power calculations with low sensitivity to resistance R. c_L Therefore, it is set to R * Even with / R=0.5, the actual phase error Δθ can be suppressed to zero.

[0102] Therefore, in this embodiment, the frequency command value ω is set... r * Accelerate from 2% to 20% of the base frequency, then decelerate from 20% to 2%. At this point, the frequency command value ω... r * It is a 10% value, which is the estimated value of the phase error when switching between the low-speed range and the medium-high-speed range.

[0103] For ω r * The low-speed range, less than 10% of the fundamental frequency, is calculated using (Equation 8). For ω r * In the medium-to-high speed range above 10% of the base frequency, calculations are performed according to (Equation 5). Figure 7 The diagram illustrates the switching characteristics from the low-speed range to the medium-high speed range, or the switching characteristics from the medium-high speed range to the low-speed range in this embodiment.

[0104] like Figure 7 As shown in the middle section, the speed range switches from low to medium-high speed in section C, and from medium-high speed to low speed in section D. Observing the phase error Δθ at the bottom, although its magnitude changes slightly during switching, the motor frequency ω... r There was no impact or torque shock, which shows that the effect of this embodiment is obvious.

[0105] In this embodiment, at ω r * It is 10% of the base frequency, a calculated value for the phase error when switching between the low-speed and medium-to-high-speed ranges, but at ω r * There is no problem switching between values ​​above and below 10% of the base frequency.

[0106] Alternatively, the phase error Δθ in the low-speed range can also be calculated separately. c_L Multiplied by a cone gain G that varies between "1" and "0". _L The calculated value of phase error Δθ in the medium-to-high speed range. c_H Multiplied by a cone gain G that varies between "0" and "1". _H The average value of the calculated phase error is taken as Δθ. c .

[0107] Here, use Figure 8 The verification method in this embodiment will be described. A voltage detector 21 and a current detector 22 are installed in the power conversion device 20 that drives the magnet motor 1, and an encoder 23 is installed on the shaft of the magnet motor 1.

[0108] The calculation unit 24 for vector voltage / current components inputs the output of the voltage detector 21, which is the three-phase AC voltage detection value (V). uc v vc v wc ), three-phase AC current detection value (i uc i vc i wc The encoder output, i.e., position θ, is used to calculate the vector voltage component v. dc v qc i of vector current components dc i qc and the detection value ω obtained by differentiating the position θ. dc .

[0109] In the observation unit 25 of each waveform, the phase error Δθ is calculated using (Equation 11). _cal .

[0110]

[0111] Change the parameters set in the controller of power converter 2 (R) * L d * L q * K e * Calculate the value of Δθ in equation (11). _cal If the result is consistent with the actual phase error Δθ, then it can be concluded that the present invention has been used.

[0112] According to this embodiment, a power conversion device can be realized that, in sensorless control that omits encoders or the like for detecting the phase of the magnet motor, it can achieve stable and high-precision control characteristics even in a low-speed range of about 10% of the base frequency, without needing to adjust the circuit parameters and control gain of the magnet motor set in the control unit (controller).

[0113] Furthermore, in the low-speed range, instead of calculating the motor frequency as described in Patent Document 1, this embodiment calculates the phase error based on the deviation of the two types of reactive power, just as it does in the medium- and high-speed range. By adopting such a structure, the motor frequency (motor rotation speed) can be calculated in the low-speed range in the same way as in the medium- and high-speed range, so that the calculated value of the phase error follows its command value, thus preventing torque surges.

[0114] Example 2

[0115] Figure 9 This is a system structure diagram of Embodiment 2, which includes a power conversion device and a magnet motor.

[0116] In Example 1, the phase error calculation unit 9 in the low-speed range calculates the phase error based on d. c axis and q c The voltage command value v of the shaft dc ** v qc ** Current detection value i dc i qc To calculate the first reactive power Q c In this embodiment, the first reactive power calculation unit 9a1 uses the amplitude value V1 of the voltage command of one phase of the three-phase AC motor 1, which is the output voltage of the magnet motor 1. * The amplitude value i1 of the current detection and the phase difference θ between the voltage command value and the current detection value. vi The sinusoidal signal is used to calculate the first reactive power Q. c .

[0117] Figure 9 The phase error calculation unit 8, frequency and phase calculation unit 10, and coordinate transformation unit 11 of the magnet motor 1 in the medium-to-high speed range are... Figure 1 Same as in Example 1. Descriptions of content identical to that in Example 1 are omitted.

[0118] Figure 10 This is the phase error calculation unit 9a in the low-speed range of this embodiment. Figure 10 Functional modules are equivalent to Figure 2 Phase error calculation unit 9 in the low-speed range.

[0119] in addition, Figure 10 The second reactive power calculation unit 9a2, subtraction unit 9a3, reactive power deviation command value 9a4, and PI control unit 9a5 are in the middle. Figure 3 The second reactive power calculation unit 92, subtraction unit 93, reactive power deviation instruction value 94, and PI control unit 95 are the same.

[0120] In this diagram, the amplitude value V1 of the three-phase AC voltage command is calculated in the first reactive power calculation unit 9a1 according to (Equation 12). * Calculate the amplitude value i1 of the current detection value according to (Equation 13), and calculate the phase θ according to (Equation 14). vi Then, using (Equation 15), the voltage amplitude value V1 of one phase of the three-phase AC is used as the output voltage of the magnet motor 1. * The current amplitude value i1 and the phase difference θ between the voltage command value and the current detection value. vi The sinusoidal signal is used to calculate the reactive power Q. c .

[0121] [Mathematical Expression 12]

[0122]

[0123] [Mathematical Expression 13]

[0124]

[0125] [Mathematical Expression 14]

[0126]

[0127] [Mathematical Expression 15]

[0128]

[0129] According to this embodiment, the first reactive power Q is calculated compared with Embodiment 1. c With fewer parameters, the computational load can be reduced.

[0130] This embodiment, which uses AC input, can also achieve high-precision control characteristics in the same way as Embodiment 1.

[0131] Example 3

[0132] Figure 11 This is a system structure diagram of the power conversion device and the magnet motor in Embodiment 3. In Embodiment 1, the phase error calculation unit 8 calculates the calculated value Δθ of the phase error in the medium-to-high speed range according to (Equation 5). c_H .

[0133] In Example 3, the phase error calculation unit 8a in the medium-to-high speed range calculates the phase error from d. c axis and q c The voltage command value v of the shaft dc ** v qc ** and current detection value i dc i qc Calculate the first active power P c .

[0134] Figure 12 This is a diagram showing the structure of the phase error calculation unit 8a in the medium-high speed range of Embodiment 3. Figure 12 Equivalent to Figure 1 Phase error calculation unit 8 in the medium-to-high speed range.

[0135] Figure 11 The magnet motor 1, vector control calculation unit 7, low-speed range phase error calculation unit 9, and frequency and phase calculation unit 10 are connected to the magnet motor 1. Figure 1 Same as in Example 1 or Example 2. Descriptions are omitted for content identical to that in Example 1 or Example 2.

[0136] Figure 12 In the phase error calculation unit 8a in the medium-to-high speed range, the first active power calculation unit 8a1 uses d c axis and q c The voltage command value v of the shaft dc ** v qc ** d c axis and q c Shaft current detection value i dc i qc Calculate the first active power P according to (Mathematical Formula 16). c .

[0137] [Mathematical Expression 16]

[0138]

[0139] The second active power calculation unit 8a2 uses d c axis and q c Shaft current detection value i dc i qc Frequency calculation value ω dc Circuit parameters of magnet motor 1 (R) * L d * L q * K e * ), calculate the second active power P according to (Mathematical Formula 17) c ^ .

[0140] [Mathematical Expression 17]

[0141]

[0142] The first active power P is input to the subtraction unit 8a3. c Second active power P c ^ Calculate its deviation ΔP c To reduce the active power deviation ΔP c The command value 8a4 for tracking the deviation of active power is "0", while the active power deviation ΔP is input to the PI control unit 8a5. c The difference between the commanded value and the deviation from the active power.

[0143] The PI control unit 8a5 calculates the phase error Δθ in the high-speed range using P (proportional) + I (integral) control according to equation (18). c_H .

[0144] [Mathematical Expression 18]

[0145]

[0146] Here, K pθ : Proportional gain for phase error calculation, K iθ : Integral gain of phase error calculation.

[0147] According to this embodiment, compared with the phase error calculation in the medium-to-high speed range using the extended back electromotive force method in Embodiment 1, since the inductor setting value is not used, the phase error calculation in the medium-to-high speed range can be performed without being affected by the inductor calculation error. Furthermore, it has the same effects as Embodiment 1.

[0148] Example 4

[0149] Figure 13 This is a system structure diagram of Embodiment 4, which includes a power conversion device and a magnet motor.

[0150] In Example 3, from d c axis and q c The voltage command value v of the shaft dc ** v qc ** and current detection value i dc i qc Calculate the first active power P c .

[0151] In this embodiment, the amplitude value V1 of the three-phase AC voltage command is used. * The amplitude value i1 and phase θ of the current detection vi The cosine signal is used by the first active power calculation unit 9b1 to calculate the first active power P. c Descriptions of content identical to those in Examples 1 through 3 are omitted.

[0152] Figure 14 This is a diagram showing the calculation unit 8b for calculating the phase error in the medium-high speed range of Embodiment 3. Figure 14 Equivalent to Figure 2 Phase error calculation unit 8 in the medium-to-high speed range. Figure 13 The magnet motor 1 to the vector control calculation unit 7, the phase error calculation unit 9 in the low speed range to the coordinate transformation unit 11 and Figure 1 same.

[0153] Figure 14 The second active power calculation unit 8b2, subtraction unit 8b3, active power deviation command value 8b4, and PI control unit 8b5 are included. Figure 12The second active power calculation unit 8a2, the subtraction unit 8a3, the active power deviation instruction value 8a4, and the PI control unit 8a5 are the same.

[0154] Figure 14 In the first active power calculation unit 9b1, the amplitude value V1 of the three-phase AC voltage command is calculated according to the above (Equation 12). * Calculate the amplitude value i1 of the current detection value according to the above (Equation 13), and then calculate the phase difference θ between the voltage command value and the current detection value according to the above (Equation 14). vi Then, the first active power calculation unit 9b1 uses (Equation 19) the voltage amplitude value V1 of one phase of the three-phase AC power. * The current amplitude value i1, and the phase difference θ between the voltage command value and the current detection value. vi The cosine signal is used to calculate the first active power P. c .

[0155] [Mathematical Expression 19]

[0156]

[0157] According to this embodiment, similar to Embodiment 3, compared to Embodiment 1, it is possible to perform phase error calculation in the medium-to-high speed range without being affected by the inductance calculation error.

[0158] Using this embodiment, the same high-precision control characteristics can be achieved as in Embodiment 1.

[0159] Example 5

[0160] Figure 15 This is a system architecture diagram of Embodiment 5, which includes a power conversion device, a magnet motor, and an IoT (Internet of Things) controller.

[0161] Examples 1 to 4 describe the structure for setting the circuit parameters of the magnet motor 1 in the controller (microcomputer, etc.) of the power converter. However, this example describes the method of resetting the circuit parameters obtained by machine learning to the controller of the power converter by feeding back the state variables of the upper-level IoT controller.

[0162] Figure 15 The magnet motor 1 in the coordinate transformation unit 11 and Figure 1 All functional modules are identical. IoT controller 12 is an IoT controller that performs machine learning. Descriptions of contents identical to those in Examples 1 to 4 are omitted.

[0163] In this embodiment, the voltage command value v dc ** v qc** and current detection value i dc i qc Calculated value of phase error Δθ c Feedback is sent to the upstream IoT controller 12, where machine learning is performed based on current detection waveforms, etc. The circuit parameters (R) obtained through machine learning are then processed. * L d * L q * K e * Reset to the controller of the control unit, i.e., the power converter 2.

[0164] According to this embodiment, the circuit parameters can be reset by the upper-level IoT controller 12 through machine learning of the operating status.

[0165] In this embodiment, the same stable and high-precision control characteristics as in Embodiment 1 can also be achieved.

[0166] Example 6

[0167] Figure 16 This is a system structure diagram of Embodiment 6, which includes a power conversion device, a magnet motor, and devices such as personal computers, tablets, and smartphones.

[0168] This embodiment applies to a magnet motor drive system. In the figure, the constituent elements—magnet motor 1, coordinate transformation unit 5 to coordinate transformation unit 11—are shown. Figure 1 Same as in Examples 1 to 5. Descriptions are omitted for content that is the same as in Examples 1 to 5.

[0169] As Figure 16 The magnetic motor 1, a constituent element of the device, is driven by the power conversion device 20. In the power conversion device 20, Figure 1 The coordinate transformation unit 5, speed control calculation unit 6, vector control calculation unit 7, phase error calculation unit 8 in the medium-to-high speed range, phase error calculation unit 9 in the low-speed range, frequency and phase calculation unit 10, and coordinate transformation unit 11 are implemented as software 20a. Furthermore, Figure 1 The power converter 2, DC power supply 3, and current detector 4 are implemented as hardware in the power conversion device 20.

[0170] In addition, the switching frequency 26 (ω) of the software 20a in the low-speed range / medium-high-speed range can be set and changed using a higher-level device such as a digital operator 20b, a personal computer 28, a tablet 29, or a smartphone 30. chg "Control response of phase error in low speed range 27, i.e., ω" c ".

[0171] right Figure 4 The frequency and phase calculation unit 10, as shown, receives the switching frequency 26 (ω) of the low-speed / medium-high-speed range from an upstream device such as a personal computer 28, tablet 29, or smartphone 30. chg Then, the frequency and phase calculation unit 10 uses the frequency command value ω, which is input to the frequency and phase calculation unit 10. r * The switching frequency between the low-speed range and the medium-to-high-speed range is 26, i.e., ω. chg The structure of the comparison section for comparison.

[0172] If the frequency command value ω r * The switching frequency ω in the low-speed / medium-high-speed range chg If the output is below 26, the switching unit 101 treats its output as a low-speed range and switches it to Δθ. c =Δθ c_L Additionally, if the frequency command value ω r * The switching frequency ω between low-speed and medium-high-speed ranges chg If the speed is 26, then the switching unit 101 is considered to be in the medium-high speed range and switches to switch Δθ. c =Δθ c_H The calculated value of phase error Δθ c Output.

[0173] From upstream devices such as personal computers 28, tablets 29, and smartphones 30 Figure 3 The phase error calculation unit 9 in the low-speed range inputs the control response 27, i.e., ω, which represents the phase error in the low-speed range. c Using this structure, a control response ω based on the phase error in the low-speed range can be achieved. c The gain of the PI control unit 95 is Kp (Equation 9). pll (PLL-controlled proportional gain), Ki pll (PLL-controlled integral gain).

[0174] Additionally, the switching frequency of "low-speed range / medium-high-speed range is 26, i.e., ω" chg "Control response of phase error in low speed range 27, i.e., ω" c "It can also be set on the fieldbus of programmable logic controllers, local area networks connected to computers, IoT controllers, etc."

[0175] Furthermore, the structure of the power conversion device is not limited to Embodiment 1, and the structures of Embodiments 2 to 5 can also be used.

[0176] If this embodiment is applied to a magnet motor drive system, high-precision control characteristics can be achieved in sensorless vector control.

[0177] According to Embodiment 6, the setting values ​​such as the switching frequency 26 in the low-speed / medium-high-speed range can be changed externally from the power conversion device. Furthermore, in this embodiment, the same as in Embodiment 1, more stable and higher-precision control characteristics can also be achieved.

[0178] In Examples 1 to 5, in the first reactive power Q c That is, (Equation 6) and the second reactive power Q c ^ That is, the current detection value i is used in (Equation 7). dc i qc However, the current command value i can also be used. d * i q * In addition, the first active power P c That is, (Equation 16) and the second active power P c ^ That is, the current detection value i is used in (Equation 17). dc i qc However, the current command value i can also be used. d * i q * .

[0179] Furthermore, in Examples 1 to 5, the current command value i d * i q * and current detection value i dc i qc Generate voltage correction value Δv dc Δv qc The calculation shown in Equation 4 involves adding the voltage correction value to the vector control voltage reference value. However, this is not limited to the calculation; the current command value i can also be used. d * i q * and current detection value i dc i qc The intermediate current command value i, shown in Equation 20, is used in the generation of vector control operations. d ** i q ** Using frequency-derived value ω dc The vector control operation shown in (Equation 21) is performed on the circuit parameters of the magnet motor 1.

[0180] [Mathematical Expression 20]

[0181]

[0182] [Mathematical Expression 21]

[0183]

[0184] Here, K pd1 :d c The proportional gain of the axis controlled by current, K id1 :d c The integral gain of the shaft current control, K pq1 :q c The proportional gain of the axis controlled by current, K iq1 :q c The integral gain of the current-controlled shaft, T d : Electrical time constant of the d-axis (L) d / R), T q : Electrical time constant of the q-axis (L q / R).

[0185] Alternatively, it can be derived from the current command value i d * i q * and current detection value i dc i qc The d used in the vector control operation is generated using (Equation 22). c Voltage correction value Δv of the proportional operational component of the axis d_p * d c Voltage correction value Δv of the integral component of the axis d_i * q c Voltage correction value Δv of the proportional operational component of the axis q_p * q c Voltage correction value Δv of the integral component of the axis q_i * Calculate the usage frequency value ω dc The vector control operation is shown in Equation 23 for the circuit parameters of the magnet motor 1.

[0186] [Mathematical Expression 22]

[0187]

[0188] [Mathematical Expression 23]

[0189]

[0190] Here, K pd2 :d c The proportional gain of the axis controlled by current, K id2 :d c The integral gain of the shaft current control, K pq2 :q c The proportional gain of the axis controlled by current, K iq2 :q c Integral gain controlled by shaft current.

[0191] Alternatively, d can be used. c Current command value i of the shaft d * and q c Shaft current detection value i qc First-order hysteresis signal i qctd Frequency calculation value ω dc The circuit parameters of the magnet motor 1 are subjected to vector control calculation as shown in Equation 24.

[0192] [Mathematical Expression 24]

[0193]

[0194] In addition, in Examples 1 to 6, the switching element constituting the power converter 2 can be a Si (silicon) semiconductor element, or a wide-bandgap semiconductor element such as SiC (silicon carbide) and GaN (gallium nitride).

[0195] Explanation of reference numerals in the attached figures

[0196] 1…Magnetic motor, 2…Power converter, 3…DC power supply, 4…Current detector, 5…Coordinate transformation unit, 6…Speed ​​control calculation unit, 7…Vector control calculation unit, 8…Phase error calculation unit for medium-high speed range, 9…Phase error calculation unit for low speed range, 10…Frequency / phase calculation unit, 11…Coordinate transformation unit, 20…Power conversion device, 20a…Software unit of power conversion device, 20b…Digital operator of power conversion device, 21…Voltage detector, 22…Current detector, 23…Encoder, 24…Calculation unit for vector current components, 25…Observation unit for waveforms of each part, 26…Specified switching frequency for low speed range / medium-high speed range, 27…Control response for specified low speed range, 28…Personal computer, 29…Tablet, 30…Smartphone, i d * …the current command value for the d-axis, i q * …the commanded value of the q-axis current, ω dc …frequency estimate, ω r …the frequency of the magnet motor, vdc * v dc ** v dc ** v dc *** v dc **** v dc ***** …the voltage command value for the d-axis, v qc * v qc ** v qc *** v qc **** v qc ***** …the voltage command value for the q-axis, Q c …First reactive power, Q c ^ …Second reactive power, P c …first active power, P c ^ …the second active power, Δθ c_L …the estimated value of the phase error in the low-speed range, Δθ c_H …the estimated value of the phase error in the medium-to-high speed range, Δθ c …the estimated value of the phase error.

Claims

1. A power conversion device, characterized in that, include: A power converter that outputs a signal to the motor that makes the motor's output frequency, output voltage, and output current variable; and The control unit that controls the power converter The control unit, The first reactive power and the first active power are calculated from the output voltage and the output current, and the second reactive power and the second active power are calculated from the output current, circuit parameters, and frequency. The first reactive power and the second reactive power are calculated when the rotational speed of the motor is in the low-speed range. To calculate the first phase error estimate, proportional control and integral control are used in a manner that makes the deviation between the first reactive power and the second reactive power zero. The first active power and the second active power are calculated when the rotational speed of the motor is in the medium-high speed range. To ensure that the deviation between the first active power and the second active power is zero, proportional control and integral control are performed, or the second phase error is calculated using an extended back electromotive force method. The frequency estimation value is controlled in such a way that either the first or second phase error estimation value follows the commanded value of the phase error estimation value. Based on the magnitude of the frequency command value, the first phase error estimate is selected in the low-speed range to control the frequency estimate value, and the second phase error estimate is selected in the medium-to-high-speed range to control the frequency estimate value.

2. The power conversion device as described in claim 1, characterized in that: The control unit, The first reactive power is calculated based on the voltage command values ​​on the d-axis and q-axis, as well as the current detection values ​​on the d-axis and q-axis. The second reactive power is calculated based on the current detection value or current command value of the d-axis, the current detection value or current command value of the q-axis, the circuit parameters of the motor, and the frequency estimation value.

3. The power conversion device as described in claim 1, characterized in that: The control unit, The first reactive power is calculated based on the sinusoidal signal of the voltage amplitude, current amplitude, and phase difference between the voltage command value and the current detection value of one phase of a three-phase AC system. The second reactive power is calculated based on the circuit parameters of the motor, the current detection values ​​or current command values ​​of the d-axis and q-axis, and the frequency estimation value.

4. The power conversion device as described in claim 1, characterized in that: The control unit, The first active power is calculated from the output voltage and the output current of the motor. The second active power is calculated from the circuit parameters, output current, and frequency of the motor. The second phase error estimate is calculated in such a way that the first active power follows the second active power.

5. The power conversion device as described in claim 4, characterized in that: The control unit, The first active power is calculated based on the voltage command values ​​on the d-axis and q-axis, as well as the current detection values ​​on the d-axis and q-axis. The second active power is calculated from the circuit parameters of the motor, the current detection value or current command value of the d-axis, the current detection value or current command value of the q-axis, and the frequency estimation value.

6. The power conversion device as described in claim 4, characterized in that: The control unit, The first active power is calculated based on the cosine signal of the phase difference between the voltage amplitude, current amplitude, voltage command value, and current detection value of one phase of a three-phase AC system. The second active power is calculated based on the circuit parameters of the motor, the current detection values ​​or current command values ​​of the d-axis and q-axis, and the frequency estimation value.

7. The power conversion device as described in claim 1, characterized in that: The control unit, The output voltage, the output current, and the first or second phase error estimate are fed back to the IoT controller, which is the upper-level device, for analysis. Correct the circuit parameters of the motor.

8. The power conversion device as described in claim 1, characterized in that: The control unit, The control response can be set or changed from the upper-level device to switch the frequency values ​​between the low-speed range and the medium-high speed range, or to the proportional control or integral control settings used to calculate the phase error.

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