Switching power supply device and power supply system

Abstract

A switching power supply device of one embodiment of the present application includes an input terminal pair, an output terminal pair, a transformer, an inverter circuit configured in a manner including a first switching element and a second switching element, a resonance inductor, and a resonance capacitor, a rectification smoothing circuit, and a control portion that controls switching operations of the first switching element and the second switching element. The first switching element and the second switching element are connected in series with each other between a pair of connection lines. The resonance inductor, the resonance capacitor, and a primary winding are connected in series with each other in an unspecified order between a connection point of the first switching element and the second switching element and an input terminal of one of the input terminal pair. The control portion controls the switching operations in accordance with a voltage across the resonance capacitor and a voltage across the primary winding.

Description

Technical Field

[0001] The present invention relates to a switching power supply device that uses switching elements for voltage conversion and a power supply system having such a switching power supply device. Background Technology

[0002] As an example of a switching power supply device, various DC-DC converters have been proposed and put into practical use (see, for example, Patent Document 1). Such DC-DC converters generally include: an inverter circuit with switching elements, a power transformer (transformer), and a rectification and smoothing circuit.

[0003] Existing technical documents

[0004] Patent documents

[0005] Patent Document 1: Japanese Patent No. 6323258 Summary of the Invention

[0006] However, such switching power supply devices, including DC-DC converters, generally require miniaturization. There is a need to provide a miniaturized switching power supply device and a power supply system incorporating such a device.

[0007] A switching power supply device according to one embodiment of the present invention includes: an input terminal pair for inputting an input voltage; an output terminal pair for outputting an output voltage; a transformer having a primary winding and a secondary winding; an inverter circuit disposed between the input terminal pair and the primary winding, and configured to include a first switching element, a second switching element, a resonant inductor, and a resonant capacitor; a rectifier smoothing circuit disposed between the output terminal pair and the secondary winding, and configured to include a rectifier circuit having multiple rectifier elements and a smoothing circuit having a smoothing capacitor; and a control unit for controlling the switching operations of the first switching element and the second switching element of the inverter circuit. The first switching element and the second switching element are connected in series between a pair of connecting lines that are individually connected to the input terminal pair. The resonant inductor, the resonant capacitor, and the primary winding are connected in series between the connection point of the first switching element and the second switching element and the input terminal of one of the input terminal pairs in an unspecified order. The control unit controls the switching operation based on the voltage between the two ends of the resonant capacitor and the voltage between the two ends of the primary winding.

[0008] A power supply system according to one embodiment of the present invention includes: a switching power supply device according to one embodiment of the present invention, and a power supply for supplying the input voltage to the input terminals.

[0009] According to one embodiment of the present invention, the switching power supply device and power supply system can be miniaturized. Attached Figure Description

[0010] Figure 1 This is a circuit diagram illustrating a schematic structural example of a switching power supply device according to the first embodiment of the present invention.

[0011] Figure 2 yes Figure 1 The diagram shows the equivalent circuit of the switching power supply device.

[0012] Figure 3 This is a schematic diagram used to illustrate the current phase difference in the first embodiment.

[0013] Figure 4 This is a flowchart illustrating an example of the load current estimation method in the first embodiment.

[0014] Figure 5 This is a circuit diagram illustrating a schematic structural example of the switching power supply device according to the second embodiment.

[0015] Figure 6 This is a timing diagram showing various waveform examples used in method A of the second embodiment.

[0016] Figure 7 This is a flowchart illustrating an example (method A) of the load current estimation method in the second embodiment.

[0017] Figure 8 This is a timing diagram showing various waveform examples used in method B of the second embodiment.

[0018] Figure 9 It is used for explanation Figure 8 The waveform of the transformer excitation current shown is a schematic diagram illustrating the positive and negative asymmetry.

[0019] Figure 10 This is a flowchart illustrating another example (method B) of the method for estimating the load current in the second embodiment.

[0020] Figure 11 This is a circuit diagram illustrating the general structure of a switching power supply device as shown in Modified Example 1.

[0021] Figure 12 This is a flowchart illustrating an example of the pre-correction process for variation 1.

[0022] Figure 13 It is used for explanation Figure 12 The diagram shows an example of the correction process.

[0023] Figure 14 This is a circuit diagram illustrating the general structure of the switching power supply device in Modified Example 2.

[0024] Figure 15This is a circuit diagram illustrating the schematic structure of the switching power supply device in Modified Example 3.

[0025] Figure 16 This is a circuit diagram illustrating the schematic structure of the switching power supply device in variation 4. Detailed Implementation

[0026] The embodiments of the present invention will now be described in detail with reference to the accompanying drawings. The description will proceed in the following order.

[0027] 1. First Implementation Method (Example of estimating load current using current phase difference)

[0028] 2. Second Implementation Method (Example of estimating load current using the current-time product difference)

[0029] 3. Variations

[0030] Variation 1 (An example of performing pre-correction processing for the defined parameters)

[0031] Variation Example 2 (Example using a bridge rectifier circuit)

[0032] Variations 3 and 4 (examples of synchronous rectifier circuits)

[0033] 4. Other variations

[0034] <1. First Implementation Method>

[0035] [structure]

[0036] Figure 1 This is a circuit diagram illustrating a schematic structural example of a switching power supply device (switching power supply device 1) according to the first embodiment of the present invention. This switching power supply device 1 transforms the DC input voltage Vin supplied by the DC input power supply 10 (e.g., a battery) into a DC output voltage Vout, supplying power to the load 9, thereby functioning as a DC-DC converter. Furthermore, examples of the load 9 include electronic devices and batteries. Additionally, as described below, this switching power supply device 1 is a so-called "(isolated half-bridge) LLC resonant type" DC-DC converter. Furthermore, the transformation method of the switching power supply device 1 can be either up-conversion (boost) or down-conversion (buck).

[0037] Here, the DC input voltage Vin corresponds to a specific example of the "input voltage" of the present invention, and the DC output voltage Vout corresponds to a specific example of the "output voltage" of the present invention. Furthermore, the DC input power supply 10 corresponds to a specific example of the "power supply" of the present invention, and the system comprising the DC input power supply 10 and the switching power supply device 1 corresponds to a specific example of the "power supply system" of the present invention.

[0038] The switching power supply device 1 includes two input terminals T1 and T2, two output terminals T3 and T4, an inverter circuit 2, a transformer 3, a rectifier and smoothing circuit 4, a drive circuit 5, and a load current estimation unit 6. A DC input voltage Vin is input between the input terminals T1 and T2, and a DC output voltage Vout is output between the output terminals T3 and T4.

[0039] Here, input terminals T1 and T2 correspond to a specific example of the "input terminal pair" of the present invention, and output terminals T3 and T4 correspond to a specific example of the "output terminal pair" of the present invention.

[0040] Furthermore, an input capacitor may be configured, for example, between the primary high-voltage line L1H connected to input terminal T1 and the primary low-voltage line L1L connected to input terminal T2. Specifically, at the location between the inverter circuit 2 (described later) and input terminals T1 and T2, one end (one end) of the input capacitor may be connected to the primary high-voltage line L1H, and the other end (the other end) of the input capacitor may be connected to the primary low-voltage line L1L. Such an input capacitor is used to stabilize the DC input voltage Vin input from input terminals T1 and T2.

[0041] (Inverter Circuit 2)

[0042] Inverter circuit 2 is configured between input terminals T1, T2 and the primary winding 31 of transformer 3 (described later). This inverter circuit 2 has two switching elements S1, S2, a resonant inductor Lr, and a resonant capacitor Cr, and is a so-called "half-bridge" inverter circuit. Furthermore, the resonant inductor Lr can also be formed from the leakage inductance of transformer 3 (described later), or it can be provided separately from such leakage inductor.

[0043] Here, the aforementioned primary high-voltage line L1H and primary low-voltage line L1L each correspond to a specific example of the "pair of connecting lines" of the present invention. Furthermore, the switching element S1 corresponds to a specific example of the "first switching element" of the present invention, and the switching element S2 corresponds to a specific example of the "second switching element" of the present invention.

[0044] Furthermore, as switching elements S1 and S2, for example, field-effect transistors (MOS-FETs) and IGBTs (Insulated Gate Bipolar Transistors) are used. Figure 1In the example shown, switching elements S1 and S2 are each constructed from a MOS-FET. Thus, when using MOS-FETs as switching elements S1 and S2, the capacitor and diode connected in parallel with each switching element S1 and S2 (in...) Figure 1 (Not shown in the figure) can be formed by the parasitic capacitance or parasitic diode of the MOS-FET, respectively.

[0045] In this inverter circuit 2, two switching elements S1 and S2 are connected in series between the input terminals T1 and T2 (between the primary high-voltage line L1H and the primary low-voltage line L1L). Specifically, switching element S1 is arranged between the primary high-voltage line L1H and the connection point P1, and switching element S2 is arranged between the connection point P1 and the primary low-voltage line L1L.

[0046] Furthermore, the resonant inductor Lr and resonant capacitor Cr of the inverter circuit 2 are connected in series with the primary winding 31 of the transformer 3 (described later) between the aforementioned connection point P1 and the primary side low-voltage line L1L. Specifically, in Figure 1 In the example, the first end (one end) of the resonant inductor Lr is connected to connection point P1, and the second end (the other end) of the resonant inductor Lr is connected to one end (connection point P3) of the primary winding 31. Furthermore, the other end of the primary winding 31 is connected to connection point P2, which serves as the first end (one end) of the resonant capacitor Cr, and the second end (the other end) of the resonant capacitor Cr is connected to the primary side low-voltage line L1L. Furthermore, as... Figure 1 As shown, the second end of the resonant capacitor Cr is connected to ground GND through the primary side low voltage line L1L.

[0047] Furthermore, the aforementioned connection point P1 corresponds to a specific example of the "connection point (the connection point between the first switching element and the second switching element)" of the present invention.

[0048] In the inverter circuit 2 with such a structure, each switching element S1, S2 performs a switching operation (on / off operation) according to the drive signals SG1, SG2 supplied by the drive circuit 5 described later, as follows. That is, the DC input voltage Vin applied between the input terminals T1, T2 is converted into an AC voltage and output to the transformer 3 (primary winding 31).

[0049] (Transformer 3)

[0050] Transformer 3 has one primary winding 31 and two secondary windings 321 and 322.

[0051] In the primary winding 31, one end of the primary winding 31 (connection point P3) is connected to the other end of the aforementioned resonant inductor Lr, and the other end of the primary winding 31 is connected to one end of the aforementioned resonant capacitor Cr (connection point P2).

[0052] In secondary winding 321, the first end of secondary winding 321 is connected to the cathode of rectifier diode 41 via connecting line L21 (described later), and the second end of secondary winding 321 is connected to the center tap P6 within the rectifier smoothing circuit 4 (described later). In secondary winding 322, the first end of secondary winding 322 is connected to the cathode of rectifier diode 42 via connecting line L22 (described later), and the second end of secondary winding 322 is connected to the aforementioned center tap P6. In summary, the second ends of secondary windings 321 and 322 are mutually connected to each other via the center tap P6.

[0053] The transformer 3 performs voltage transformation on the voltage generated by the inverter circuit 2 (the rectangular pulse wave voltage input to the primary winding 31 of the transformer 3), and outputs AC voltage from each end of the secondary windings 321 and 322. Furthermore, the degree of voltage transformation of the DC output voltage Vout to the DC input voltage Vin in this case is determined by the turns ratio of the primary winding 31 to the secondary windings 321 and 322 and the switching frequency fsw, described later.

[0054] (Rectifier smoothing circuit 4)

[0055] The rectifier-smoothing circuit 4 has two rectifier diodes 41 and 42 and one output smoothing capacitor Cout. Specifically, the rectifier-smoothing circuit 4 includes a rectifier circuit with rectifier diodes 41 and 42, and a smoothing circuit with output smoothing capacitor Cout.

[0056] Furthermore, these two rectifier diodes 41 and 42 correspond to a specific example of the "multiple rectifier elements" of the present invention. Additionally, the output smoothing capacitor Cout corresponds to a specific example of the "smoothing capacitor" of the present invention.

[0057] The aforementioned rectifier circuit is a so-called "center-tapped" rectifier circuit. That is, the anodes of rectifier diodes 41 and 42 are each connected to the ground line LG. The cathode of rectifier diode 41 is connected to the aforementioned first terminal of secondary winding 321 via connecting line L21, and the cathode of rectifier diode 42 is connected to the aforementioned first terminal of secondary winding 322 via connecting line L22. Furthermore, as mentioned earlier, the second terminals of secondary windings 321 and 322 are mutually connected to the center tap P6, which is connected to the aforementioned output terminal T3 via the output line LO. Additionally, the aforementioned ground line LG is connected to the aforementioned output terminal T4.

[0058] In the smoothing circuit described above, an output smoothing capacitor Cout is connected between the output line LO and the ground line LG (between output terminals T3 and T4). That is, the first end of the output smoothing capacitor Cout is connected to the output line LO, and the second end of the output smoothing capacitor Cout is connected to the ground line LG.

[0059] Regarding the rectifier-smoothing circuit 4 with this structure, the AC voltage output from the transformer 3 is rectified and output in a rectifier circuit configured with rectifier diodes 41 and 42. Furthermore, in a smoothing circuit configured with an output smoothing capacitor Cout, the voltage rectified by the aforementioned rectifier circuit is smoothed, thereby generating a DC output voltage Vout. Moreover, through the DC output voltage Vout generated in this way, the load current Iout (DC output current) flows to the aforementioned load 9, supplying power to the load 9 from the output terminals T3 and T4 (see reference). Figure 1 ).

[0060] (Driver Circuit 5)

[0061] The drive circuit 5 is a circuit that controls the switching operations of the switching elements S1 and S2 of the inverter circuit 2 separately. Specifically, the drive circuit 5 controls the switching operations (on / off operations) of each switching element S1 and S2 by supplying drive signals SG1 and SG2 to each switching element S1 and S2 individually.

[0062] Here, the drive circuit 5 performs frequency control when controlling the switching action (switching drive) of each switching element S1, S2. That is, PFM (Pulse Frequency Modulation) control is performed in the drive signals SG1, SG2.

[0063] Furthermore, the drive circuit 5 performs the aforementioned switching drive so that the switching frequency fsw of each of the switching elements S1 and S2 is variable, and each of the switching elements S1 and S2 operates with a fixed duty cycle. Additionally, when the on-time of the switching elements S1 and S2 is represented by Ton1 and Ton2 respectively, the duty cycles of each of the switching elements S1 and S2 are as follows. That is, when the switching period is Tsw = (1 / fsw), the duty cycles of each of the switching elements S1 and S2 are expressed using the switching period Tsw as (Ton1 / Tsw) and (Ton2 / Tsw).

[0064] Furthermore, the drive circuit 5 performs switching drive using the estimated value of the load current Iout obtained in the load current estimation unit 6 (hereinafter referred to as the estimated load current value Iout(est)), which will be described in detail later. Specifically, by performing such switching drive using the estimated load current value Iout(est), the drive circuit 5 controls the value of the load current Iout within an appropriate range.

[0065] (Load Current Estimation Unit 6)

[0066] The load current estimation unit 6 performs the aforementioned estimation of the load current Iout (obtaining the estimated load current value Iout(est) using the method described later). Specifically, the load current estimation unit 6 obtains the primary-side converted value IoutAC of the load current Iout based on the resonant inductor current ILr flowing through the resonant inductor Lr and the transformer excitation current Im flowing through the primary winding 31, which will be described in detail later. Furthermore, the load current estimation unit 6 obtains the estimated load current value Iout(est) based on this primary-side converted value IoutAC.

[0067] Furthermore, particularly in this embodiment, the load current estimation unit 6 uses the current phase difference φ1 between the resonant inductor current ILr and the transformer excitation current Im to obtain the estimated load current value Iout(est), which will be described in detail later. Specifically, the load current estimation unit 6 obtains this current phase difference φ1 based on the voltage Vcr between the two ends of the resonant capacitor Cr and the voltage Vnp between the two ends of the primary winding 31.

[0068] Such a load current estimation unit 6 Figure 1 As shown, it has two detection units 61 and 62 and a control unit 60.

[0069] The detection unit 61 is used to detect the distance between the two ends of the resonant capacitor Cr (e.g., Figure 1 As shown, with the second terminal of the resonant capacitor Cr connected to the common ground GND with the detection unit 61, the circuit for the voltage Vcr between the connection point P2 and the primary side low-voltage line L1L is described. Furthermore, the detection unit 61 performs predetermined calculations based on the detected voltage Vcr, which will be described in detail later.

[0070] The detection unit 62 is used to detect the distance between the two ends of the primary winding 31. Figure 1 The circuit shown is for the voltage Vnp between connection points P2 and P3. Furthermore, the detection unit 62 performs predetermined calculations based on the detected voltage Vnp, which will be described in detail later.

[0071] The control unit 60 is a circuit that calculates the estimated load current value Iout(est) based on the calculation results of each detection unit 61 and 62, through the predetermined calculation process described later. Furthermore, the control unit 60 uses the calculated estimated load current value Iout(est) to control the switching drive of the drive circuit 5 (the switching operation of each switching element S1 and S2). Specifically, the control unit 60 controls this switching operation based on the voltage Vcr between the two ends of the resonant capacitor Cr and the voltage Vnp between the two ends of the primary winding 31, which will be described in detail later. Thus, as mentioned above, the value of the load current Iout is controlled within an appropriate range.

[0072] [Actions, Functions, and Effects]

[0073] (A. Basic movements)

[0074] In this switching power supply device 1, in the inverter circuit 2, the DC input voltage Vin supplied from the DC input power supply 10 through the input terminals T1 and T2 is converted by switching elements S1 and S2, thereby generating a rectangular pulse wave voltage. This rectangular pulse wave voltage is provided to the primary winding 31 of the transformer 3, and is transformed in the transformer 3, thereby outputting the transformed AC voltage from the secondary windings 321 and 322.

[0075] In the rectifier-smoothing circuit 4, the AC voltage output from the transformer 3 (the aforementioned transformed AC voltage) is rectified by the rectifier diodes 41 and 42 within the rectifier circuit, and then smoothed by the output smoothing capacitor Cout within the smoothing circuit. As a result, a DC output voltage Vout is output from the output terminals T3 and T4. Then, through this DC output voltage Vout, the load current Iout flows to the load 9, supplying power to the load 9.

[0076] (B. Estimation of load current Iout)

[0077] Next, refer to Figure 1 , Figures 2-4 The estimation operation of the load current Iout of the load current estimation unit 6 (the operation of deriving the estimated load current value Iout(est)) will be described in detail.

[0078] (B-1. Summary of estimated actions)

[0079] Figure 2 yes Figure 1 A schematic circuit diagram of the equivalent circuit of the switching power supply device 1 shown (the equivalent circuit of the aforementioned LLC resonant converter). Furthermore, in this... Figure 2In this diagram, VinAC represents the AC input voltage, Lm represents the magnetizing inductance of transformer 3, RL(AC) represents the primary load, and n represents the turns ratio of the primary winding 31 to the secondary windings 321 and 322 (= number of turns of primary winding 31 np / number of turns of secondary windings 321 and 322 ns). Furthermore, the resistance value of the primary load RL(AC) is expressed using the aforementioned turns ratio n and the resistance value RL of load 9 as (8·n 2 ·RL) / π 2 Furthermore, this method is called the fundamental frequency approximation (FHA), which is a method that focuses only on the fundamental frequency of the rectangular wave voltage for approximate analysis.

[0080] Such as Figure 2 As shown, the aforementioned resonant inductor current ILr is shunted into the primary-side equivalent value IoutAC of the aforementioned transformer magnetizing current Im and the aforementioned load current Iout. In summary, by obtaining the primary-side equivalent value IoutAC of the load current Iout from the difference between the resonant inductor current ILr and the transformer magnetizing current Im (=ILr-Im), it is possible to estimate the load current Iout flowing through the secondary side from the primary side (refer to…). Figure 1 ).

[0081] in addition, Figure 3 This is a schematic diagram used to illustrate the aforementioned current phase difference φ1 in this embodiment. Furthermore, in this... Figure 3 In the example, the three currents i1, i2, i (=i1-i2) composed of sine waves can be replaced with, respectively, as shown below. Figure 3 The values ​​within parentheses above represent the primary-side converted values ​​IoutAC of the resonant inductor current ILr, transformer magnetizing current Im, and load current Iout. Additionally, in this... Figure 3 In this context, it also represents the current phase difference φ1 between current i1 (resonant inductor current ILr) and current i2 (transformer excitation current Im) and the current phase difference φ3 between current i2 and current i (the primary side conversion value IoutAC of load current Iout).

[0082] In this case, if Figure 3 The currents i1 and i2 shown are defined as the following equations (1) and (2), respectively. Then the current i (=i1-i2) can be expressed as the following equations (3)~(8).

[0083] i1 = Im1·sin(ωt+φ1)

[0084] =Im1·sinωt·cоsφ1+Im1·cоsωt·sinφ1……(1)

[0085] i2=Im2·sinωt……(2)

[0086] (Im1, Im2: amplitude values, ω: angular frequency, t: time)

[0087] i(=i1-i2)=A·sinωt+B·cоsωt……(3)

[0088] =(A 2 +B 2 ) 1 / 2·sin(ωt+φ3)……(4)

[0089] A=Im1·cоsφ1-Im2……(5)

[0090] B=Im1·sinφ1……(6)

[0091] (A 2 +B 2 ) 1 / 2 = (Im1) 2 +Im2 2 -2·Im1·Im2·cosφ1) 1 / 2……(7)

[0092] φ3=tan －1 (B / A)……(8)

[0093] Then, if we use these equations (3) to (8) to replace the currents i1, i2, i with the primary-side converted values ​​IoutAC of the resonant inductor current ILr, the transformer excitation current Im, and the load current Iout, respectively, as described above, then it is as follows. That is, ILr(peak), Im(peak), and IoutAC(peak), which are these peak values, are represented by the following equations (9) to (12) (see also...) Figure 2 (The various parameters in the text).

[0094] ILr(peak)=ω·Cr·Vcr(peak)……(9)

[0095] (Cr: capacitance of the resonant capacitor Cr, Vcr(peak): peak value of the voltage Vcr)

[0096] Im(peak) = VoutAC / (ω·Lm)

[0097] = (4·n·Vout) / (π ·ω·Lm)… (10)

[0098] VoutAC=(4 / π)·n·Vout·sinωt……(11)

[0099] IoutAC(peak) =

[0100] (ILr(peak) 2 +Im(peak) 2 -2·ILr(peak)·Im(peak)·cоsφ1) 1 / 2……(12)

[0101] Therefore, in the method for estimating the load current Iout in the following embodiment, such a calculation method is used to estimate the load current Iout.

[0102] (B-2. Detailed information on estimating the motion)

[0103] Figure 4 This is a flowchart illustrating an example of the load current Iout estimation method performed by the load current estimation unit 6 in this embodiment.

[0104] In this example of the estimation method, firstly, the detection unit 61 measures the voltage Vcr between the two ends of the resonant capacitor Cr as detected above (refer to...). Figure 1 ), respectively obtain the zero-crossing points of the above ILr(peak) and the voltage Vcr ( Figure 4 Step S11). Specifically, the detection unit 61 calculates ILr(peak) based on the voltage Vcr using the above formula (9).

[0105] Next, the detection unit 62 detects the voltage Vnp between the two ends of the primary winding 31 as described above (refer to...). Figure 1 Equivalent to Figure 2 In step S12, the detection unit 62 calculates Im(peak) and the zero-crossing point of the voltage Vnp based on the voltage Vnp using the above equations (10) and (11).

[0106] Next, the control unit 60 calculates the phase difference φ1 between the resonant inductor current ILr and the transformer excitation current Im based on the zero-crossing points of the voltages Vcr and Vnp obtained in the detection units 61 and 62 (refer to...). Figure 3 (Step S13). Specifically, the control unit 60 obtains the current phase difference φ1 from the difference between these zero crossings.

[0107] Next, the control unit 60 uses the above formula (12) to obtain the peak value (IoutAC(peak)) of the primary side conversion value of the load current Iout based on ILr(peak), Im(peak), and φ1 obtained in this way (step S14).

[0108] Then, the control unit 60 obtains an estimated value of the load current Iout (load current estimate Iout(est)) based on the IoutAC(peak) obtained in this way (step S15). Specifically, the control unit 60 obtains the load current estimate Iout(est) by using the average value of IoutAC(peak) obtained by the following formula (13).

[0109] Iout(est)=n·(2 / π)·IoutAC(peak)......(13)

[0110] thus, Figure 4 The process of estimating the load current Iout shown is now complete.

[0111] (C. Functions and Effects)

[0112] However, in LLC resonant switching power supply devices, in order to achieve constant current drop characteristics, it is generally necessary to correctly detect the value of the load current when performing overcurrent protection on the load current.

[0113] However, when using a current sensor, such as a resistive element, to detect such load current, power losses occur within the resistive element, leading to decreased efficiency and increased size of the switching power supply. Furthermore, when load current control is performed on the primary side of the switching power supply based on the detected load current, the following problems arise. Specifically, in this case, an isolation amplifier (or similar component) is needed to insulate and transmit the detected load current value (the result detected on the secondary side) to the primary side, increasing the number of components and the size of the switching power supply. Additionally, since it is a current detection value, it cannot be transmitted via an optocoupler. Therefore, it is evident that in a typical LLC resonant switching power supply, miniaturization is difficult while attempting to accurately detect the load current value.

[0114] In this embodiment of the switching power supply device 1, the control unit 60 of the load current estimation unit 6 controls the switching operation based on the voltage Vcr between the two ends of the resonant capacitor Cr and the voltage Vnp between the two ends of the primary winding 31. Therefore, compared to the general LLC resonant type switching power supply device described above, the switching power supply device 1 can be miniaturized in this embodiment. Specifically, in this embodiment, the load current estimation unit 6 calculates the primary-side converted value IoutAC of the load current Iout based on the resonant inductor current ILr and the transformer magnetizing current Im, and then calculates the load current estimation value Iout(est) based on this primary-side converted value IoutAC. Then, the control unit 60 performs switching control using the load current estimation value Iout(est) obtained in this way, thereby controlling the value of the load current Iout.

[0115] Therefore, in this embodiment, unlike the general LLC resonant switching power supply device described above, the following applies. Specifically, the estimated load current Iout(est) is obtained from the primary-side converted value IoutAC of the load current Iout obtained by a current sensor without using a resistor or similar element on the primary side, and the value of the load current Iout is controlled based on this estimated load current Iout(est). As a result, in this embodiment, the power loss caused by detecting the load current Iout using a resistor or similar element as described above can be avoided, and there is no need for components that insulate and transmit the detected load current Iout value to the primary side, thus reducing the number of components.

[0116] Furthermore, as an example, in a switching power supply device 1 with a DC input voltage Vin = 200~400V, a DC output voltage Vout = 48V, and a rated current = 21A, the load current Iout can be estimated within an error of about 10%.

[0117] As can be seen from the above, in this embodiment, compared with the general LLC resonant switching power supply device, the miniaturization of the switching power supply device 1 can be achieved.

[0118] Furthermore, as described above, power losses caused by the detection of load current Iout using resistive elements can be avoided, thus reducing the cost and improving the reliability of the switching power supply device 1. Moreover, as mentioned above, since there is no need for components that insulate and transmit the detected value of load current Iout to the primary side, the number of components is reduced, further reducing the cost and improving the reliability of the switching power supply device 1.

[0119] Furthermore, in this embodiment, since the load current estimate Iout(est) is obtained by utilizing the current phase difference φ1 between the resonant inductor current ILr and the transformer excitation current Im, the load current Iout can be easily estimated.

[0120] Furthermore, in this embodiment, since the current phase difference φ1 is obtained from the voltage Vcr between the two ends of the resonant capacitor Cr and the voltage Vnp between the two ends of the primary winding 31, the load current Iout can be estimated more easily.

[0121] Furthermore, in this embodiment, since the resonant inductor Lr of the inverter circuit 2 is composed of the leakage inductance of the transformer 3, it is not necessary to separately install the resonant inductor Lr, thus reducing the number of components. As a result, further miniaturization and cost reduction of the switching power supply device 1 can be achieved.

[0122] Furthermore, in this embodiment, since the switching elements S1 and S2 of the inverter circuit 2 are each composed of MOS-FETs, the switching frequency fsw can be increased, and the components can be miniaturized.

[0123] Furthermore, in this embodiment, since the rectifier circuit of the rectifier smoothing circuit 4 is a so-called "center-tapped" rectifier circuit, it is different from the so-called "bridge" rectifier circuit described in Variation 2 below. That is, the number of rectifier elements is reduced to two (rectifier diodes 41 and 42), resulting in miniaturization, low loss, and low cost of the rectifier circuit.

[0124] <2. Second Implementation Method>

[0125] Next, the second embodiment of the present invention will be described. Furthermore, in the following text, the same reference numerals will be used for constituent elements that are identical to those in the first embodiment described above, and their descriptions will be omitted as appropriate.

[0126] [structure]

[0127] Figure 5 This is a circuit diagram showing a schematic structural example of the switching power supply device (switching power supply device 1A) according to the second embodiment.

[0128] Furthermore, similar to the first embodiment, the system comprising the DC input power supply 10 and the switching power supply device 1A corresponds to a specific example of the "power supply system" of the present invention.

[0129] The second embodiment of the switching power supply device 1A is equivalent to the first embodiment of the switching power supply device 1, except that the load current estimation unit 6A is provided instead of the load current estimation unit 6, and the other structures are the same.

[0130] (Load current estimation unit 6A)

[0131] Like load current estimation unit 6, load current estimation unit 6A estimates the load current Iout (obtaining the estimated load current value Iout(est)). Additionally, like load current estimation unit 6, load current estimation unit 6A also obtains the primary-side converted value IoutAC of the load current Iout based on the resonant inductor current ILr and the transformer excitation current Im. Then, like load current estimation unit 6, load current estimation unit 6A also obtains the estimated load current value Iout(est) based on this primary-side converted value IoutAC.

[0132] Furthermore, particularly in this embodiment, the load current estimation unit 6A calculates the estimated load current Iout(est) using the difference between the current-time product (the integral of the current along the time axis: equivalent to the amount of charge) between the resonant inductor current ILr and the transformer excitation current Im. This will be described in detail later. More specifically, the load current estimation unit 6A modifies the estimation method for the load current Iout using this current-time product difference based on the relationship between the switching frequency fsw of the switching elements S1 and S2 and the resonant frequency fr of the resonant operation using the resonant inductor Lr and the resonant capacitor Cr. This will also be described in detail later.

[0133] exist Figure 5 In the example shown, such a load current estimation unit 6A has three detection units 61A, 62A, 63A and a control unit 60A.

[0134] Similar to detection unit 61, detection unit 61A is a circuit that detects the voltage Vcr between the two ends of the resonant capacitor Cr. Furthermore, detection unit 61A performs predetermined calculations based on the detected voltage Vcr, which will be described in detail later.

[0135] Similar to detection unit 62, detection unit 62A is a circuit that detects the voltage Vnp between the two ends of the primary winding 31. Furthermore, detection unit 62A performs predetermined calculations based on the detected voltage Vnp, which will be described in detail later.

[0136] The detection unit 63A is a circuit that detects the switching voltage Vsw corresponding to the potential at connection point P1. Furthermore, the detection unit 63A performs predetermined calculations based on the detected switching voltage Vsw, which will be described in detail later. Additionally, in Figure 5 In the example shown, the detection unit 63A is connected to the same ground (GND) as the second terminal side of the detection unit 61A and the resonant capacitor Cr. Additionally, in this... Figure 5In the example shown, the detection unit 63A is disposed between the primary side low voltage line L1L and the connection point P1, but it can also be disposed between the primary side high voltage line L1H and the connection point P1, for example.

[0137] Control unit 60A is a circuit that calculates the estimated load current value Iout(est) based on the calculation results of each detection unit 61A, 62A, and 63A, through the calculation process described later. Furthermore, similar to control unit 60, control unit 60A controls the switching drive of drive circuit 5 based on the calculated estimated load current value Iout(est). Thus, as mentioned above, the value of load current Iout is controlled within an appropriate range by drive circuit 5.

[0138] [Actions, Functions, and Effects]

[0139] (A. Estimation of load current Iout)

[0140] Next, refer to Figure 5 , Figures 6-10 The estimation operation of the load current Iout of the load current estimation unit 6A (the operation of deriving the estimated load current value Iout(est)) will be explained in detail below. Specifically, the estimation method (method A) for the case where the switching frequency fsw is higher than the resonant frequency fr (fsw>fr) and the estimation method (method B) for the case where the switching frequency fsw is lower than the resonant frequency fr (fsw<fr) will be explained in detail below.

[0141] Furthermore, the case of (fsw > fr, switching period Tsw (=1 / fsw) < resonant period Tr (=1 / fr)) is equivalent to the case where the primary-side equivalent value IoutAC of the load current Iout (along the time axis) is a continuous operating mode. On the other hand, the case of (fsw < fr, Tsw > Tr) is equivalent to the case where the primary-side equivalent value IoutAC of the load current Iout (along the time axis) is a discontinuous operating mode.

[0142] (A-1. Technique A: case where fsw > fr)

[0143] Figure 6 This is a timing diagram showing various waveform examples used in method A (the method where fsw > fr) of this embodiment. Specifically, Figure 6 (A) Figure 6 (B) Figure 6 (C) Figure 6(D) represents the waveforms of voltage Vnp, the primary-side equivalent value of load current Iout IoutAC (=ILr-Im), resonant inductor current ILr and transformer excitation current Im, and voltage Vcr, respectively. Furthermore, the horizontal axis represents time t, as is also true in the following timing diagrams.

[0144] Here, Figure 7 This is a flowchart illustrating an example of the load current Iout estimation method performed by the load current estimation unit 6A in this embodiment (method A described above).

[0145] First, in method A (where fsw > fr), the load current estimation unit 6A estimates the load current Iout using the aforementioned current-time product difference, based on the voltage Vcr between the resonant capacitor Cr and the voltage Vnp between the primary winding 31. In summary, in method A, the load current estimation unit 6A only utilizes the detection results of the aforementioned detection units 61A and 62A, and does not utilize the detection results of the detection unit 63A.

[0146] Specifically, in this method A, firstly, the detection unit 61A detects the voltage Vcr (refer to...) as described above. Figure 5 , Figure 6 (D)), respectively, find the zero-crossing point of the resonant inductor current ILr and the zero-crossing point of the resonant inductor current ILr. Figure 7 Step S21). Specifically, the detection unit 61A calculates the resonant inductor current ILr by differentiating the voltage Vcr.

[0147] Next, the detection unit 62A detects the voltage Vnp as described above (refer to...). Figure 5 , Figure 6 (A)), respectively, calculate the change ΔIm of the transformer excitation current Im (refer to) Figure 6 The zero-crossing point of the voltage Vnp (step S22). Specifically, the detection unit 62A calculates ΔIm based on the voltage Vnp using the following formula (14).

[0148] ΔIm=(Vnp·Tsw) / (4·Lm)……(14)

[0149] (Tsw: Switching cycle (reference)) Figure 6 Lm: Magnetizing inductance value of transformer 3

[0150] Next, the control unit 60A calculates the hysteresis time ΔT1 (refer to the zero-crossing points of the resonant inductor current ILr and voltage Vnp obtained in the detection units 61A and 62A) based on these zero-crossing points. Figure 6 (C) (step S23). Specifically, as Figure 6As shown, the control unit 60A obtains the hysteresis time ΔT1 from the time difference between the zero-crossing points of the current ILr and voltage Vnp of these resonant inductors.

[0151] Next, the control unit 60A uses the difference in the current-time product between the resonant inductor current ILr and the transformer excitation current Im to calculate the current-time product (a+b) of the primary-side converted value IoutAC (=ILr-Im) of the load current Iout (step S24: refer to Figure 6 (B) Specifically, the control unit 60A calculates the current-time product (a+b) using the following formula (15) based on the change ΔIm of the transformer excitation current Im and the hysteresis time ΔT1. Furthermore, the current-time products a and b are respectively as follows: Figure 6 As shown in (C).

[0152] (a+b)=(Cr·ΔVcr)-(ΔIm·ΔT1)......(15)

[0153] (Cr: capacitance value of resonant capacitor Cr, ΔVcr: difference between the maximum and minimum values ​​of voltage Vcr (refer to...) Figure 6 (D)))

[0154] Then, the control unit 60A calculates the estimated load current Iout(est) based on the current-time product (a+b) of the primary-side converted value IoutAC of the load current Iout obtained in this way (step S25). Specifically, the control unit 60A calculates the estimated load current Iout(est) using the following formula (16).

[0155] Iout(est)=(n·(a+b)) / (Tsw / 2)……(16)

[0156] (n = np / ns: the turns ratio of primary winding 31 to secondary windings 321 and 322)

[0157] thus, Figure 7 The estimation method for the load current Iout shown (method A above) is now complete.

[0158] (A-2. Technique B: Case where fsw < fr)

[0159] Figure 8 This is a timing diagram showing various waveform examples used in method B (the method where fsw < fr) of this embodiment. Specifically, Figure 8 (A) Figure 8 (B) Figure 8 (C) Figure 8 (D) Figure 8(E) represents the waveforms of the switching voltage Vsw, voltage Vnp, primary side converted value of load current IoutAC, resonant inductor current ILr, transformer excitation current Im, and voltage Vcr, respectively.

[0160] in addition, Figure 9 It is used for explanation Figure 8 The waveform example of the transformer excitation current Im shown is a schematic diagram illustrating the positive and negative asymmetry (asymmetry of positive and negative values). Figure 9 T3, as shown, represents the hysteresis time. Specifically, in this... Figure 9 (and Figure 8 In (D), regarding the waveform change (adjustment of approximate waveform) used to compensate for such positive and negative asymmetry, the transformer excitation current Im before such waveform change (shown as dashed line) and the transformer excitation current Im after waveform change (shown as solid line) are respectively represented.

[0161] In this method B (where fsw < fr), the load current estimation unit 6A considers the positive and negative asymmetry of the transformer excitation current Im to estimate the load current Iout. Specifically, as... Figure 9 As shown in the integral values ​​Sa,Sb (Sa≠Sb), the positive and negative values ​​of the transformer excitation current Im are asymmetrical in the case of (fsw<fr) compared to (fsw>fr). In this case of asymmetrical positive and negative values ​​of the transformer excitation current Im, since they cannot be canceled out, even if the detected value is integrated, the load current Iout cannot be separated. Therefore, in this method B, as described above, a waveform modification (approximate waveform adjustment) is implemented to compensate for the asymmetry of the positive and negative values ​​of the transformer excitation current Im (see reference). Figure 8 (D) Figure 9 (The arrow related to Im shown).

[0162] Here, Figure 10 This is a flowchart illustrating an example of the load current Iout estimation method performed by the load current estimation unit 6A in this embodiment (method B described above).

[0163] First, in method B (where fsw < fr), the load current estimation unit 6A, as described above, estimates the load current Iout using the aforementioned current-time product difference based on the aforementioned voltages Vcr and Vnp and the aforementioned switching voltage Vsw. In summary, unlike method A, in method B, the load current estimation unit 6A utilizes the detection results from the aforementioned detection units 61A, 62A, and 63A.

[0164] Specifically, in this method B, firstly, the process of step S21 is performed in the same manner as in method A. That is, the detection unit 61A detects the voltage Vcr as described above (refer to...). Figure 5 , Figure 8 (E)), respectively, find the zero-crossing point of the resonant inductor current ILr and the zero-crossing point of the resonant inductor current ILr. Figure 10 Step S21). Specifically, the detection unit 61A calculates the resonant inductor current ILr by differentiating the voltage Vcr.

[0165] Next, the detection unit 62A detects the voltage Vnp as described above (refer to...). Figure 5 , Figure 8 (B)), to obtain the change ΔIm of the transformer excitation current Im (refer to) Figure 8 (D) (step S32). Specifically, the detection unit 62A calculates the change ΔIm based on the voltage Vnp using the following formula (17).

[0166] ΔIm=(Vnp·Tr) / (4·Lm)……(17)

[0167] (Tr: Resonance period (=1 / fr: Reference) Figure 8 , Figure 9 ))

[0168] Secondly, the detection unit 63A detects the switching voltage Vsw (refer to) as described above. Figure 5 , Figure 8 (A)), find the zero-crossing point of the switching voltage Vsw (step S33).

[0169] Next, the control unit 60A calculates the hysteresis time ΔT2 (refer to the zero-crossing points of the resonant inductor current ILr and switching voltage Vsw obtained in the detection units 61A and 63A) based on these zero-crossing points. Figure 8 (D) (step S34). Specifically, as Figure 8 As shown, the control unit 60A obtains the hysteresis time ΔT2 from the time difference between the zero-crossing points of the resonant inductor currents ILr and the switching voltage Vsw.

[0170] Next, the control unit 60A uses the difference in the current-time product between the resonant inductor current ILr and the transformer excitation current Im to calculate the current-time product (a+b) of the primary-side converted value IoutAC (=ILr-Im) of the load current Iout (step S35: refer to Figure 8 (C)). Specifically, the control unit 60A uses the following formulas (18) and (19) to obtain the current-time product (a+b) based on the change ΔIm of the transformer excitation current Im and the hysteresis time ΔT2.

[0171] (a+b)=Cr·(ΔVcr-2·ΔVcr1)-(ΔIm·Tsw-Im2·Tr) / 2

[0172] =Cr·ΔVcr-(ΔIm·Tsw) / 2+Im2·(Tr-2·ΔT2) / 2......(18)

[0173] Im2=(2·Cr·ΔVcr1) / ΔT2……(19)

[0174] (ΔVcr1: The change in voltage Vcr during the hysteresis time ΔT2 (reference)) Figure 8 (E)))

[0175] Then, the control unit 60A calculates the estimated load current Iout(est) based on the current-time product (a+b) of the primary-side converted value IoutAC of the load current Iout obtained in this way (step S36). Specifically, the control unit 60A calculates the estimated load current Iout(est) using the following formula (20).

[0176] Iout(est)=(n·(a+b)) / (Tsw / 2)……(20)

[0177] thus, Figure 10 The process of estimating the load current Iout (method B above) is now complete.

[0178] (B. Functions and Effects)

[0179] In this embodiment of the switching power supply device 1A, the same effect can be obtained because it has essentially the same function as the switching power supply device 1 of the first embodiment.

[0180] Furthermore, especially in this embodiment, since the load current estimate Iout(est) is obtained by using the current-time product difference between the resonant inductor current ILr and the transformer excitation current Im, the load current Iout can be easily estimated.

[0181] Furthermore, in this embodiment, the estimation method for the load current Iout using the current-time product difference is modified based on the relationship between the switching frequency fsw and the resonant frequency fr. Specifically, when the switching frequency fsw is higher than the resonant frequency fr, the load current estimation unit 6A estimates the load current Iout using the current-time product difference based on the voltage Vcr across the resonant capacitor Cr and the voltage Vnp across the primary winding 31. Conversely, when the switching frequency fsw is lower than the resonant frequency fr, the load current estimation unit 6A estimates the load current Iout using the current-time product difference based on the voltages Vcr and Vnp and the aforementioned switching voltage Vsw. Therefore, in this embodiment, since an appropriate estimation method can be selected based on the relationship between these frequencies, the estimation accuracy of the load current Iout can be improved.

[0182] Furthermore, in this embodiment, since the switching frequency fsw is lower than the resonant frequency fr, the asymmetry of the positive and negative values ​​of the transformer excitation current Im is considered, and the load current Iout is estimated using the current-time product difference, as described below. In other words, even in the case of fsw < fr (as mentioned above, the primary-side converted value IoutAC of the load current Iout is in a discontinuous operating mode), the load current Iout can be estimated with high accuracy, thus further improving the estimation accuracy of the load current Iout.

[0183] Furthermore, in this embodiment, as an example, the load current Iout can be estimated with the following error.

[0184] • (fsw > fr) case (where the primary side conversion value of load current Iout, IoutAC, is in continuous operation mode): within an error of approximately 5%.

[0185] • (fsw < fr) case (where the primary-side equivalent value of load current Iout, IoutAC, is in discontinuous operation mode): within approximately 10% error.

[0186] <3. Variations>

[0187] Next, variations of the first and second embodiments described above (variations 1 to 4) will be described. Furthermore, in the following text, the same reference numerals will be added to constituent elements that are identical to those in the first or second embodiments, and their descriptions will be omitted as appropriate.

[0188] [Variation Example 1]

[0189] (structure)

[0190] Figure 11This is a circuit diagram showing a schematic structural example of the switching power supply device (switching power supply device 1B) of Modified Example 1.

[0191] Furthermore, similar to the first and second embodiments, the system comprising the DC input power supply 10 and the switching power supply device 1B corresponds to a specific example of the "power supply system" of the present invention.

[0192] The switching power supply device 1B of this modified example is equivalent to the switching power supply devices 1, 1A of the first and second embodiments, in which the following load current estimation unit 6B is provided instead of the load current estimation unit 6, 6A, and the rest of the structure is the same.

[0193] The load current estimation unit 6B estimates the load current Iout in essentially the same manner as the load current estimation unit 6 or the load current estimation unit 6A described above. However, the load current estimation unit 6B further estimates the load current Iout using predetermined parameters (described later) that are pre-corrected based on the measured value of the correspondence (output characteristics) between the load current Iout and the DC output voltage Vout.

[0194] Such pre-correction processing of parameters is performed, for example, during inspections such as when the switching power supply device 1B is shipped. In other words, the parameters are set to values ​​after such pre-correction processing (values ​​after fine-tuning in calculation). This is because the estimated value of the load current Iout (load current estimate Iout(est)) may vary due to individual deviations of various components within the switching power supply device 1B. Furthermore, examples of the aforementioned parameters include, for instance, the capacitance value of the resonant capacitor Cr, the magnetizing inductance value Lm of the transformer 3, and the detection values ​​in the load current estimation unit 6B (the aforementioned detection units 61, 62, 61A, 62A, 63A).

[0195] Here, Figure 12 This is a flowchart illustrating an example of the aforementioned pre-emptive corrective processing in Variation 1. Additionally, Figure 13 It is used for explanation Figure 12 The diagram illustrates an example of the correction process. Specifically, in this... Figure 13 In the example shown, the rated current and rated voltage are each taken as 100%, and the output characteristics of the switching power supply device 1B (the correspondence between the load current Iout and the DC output voltage Vout when the overcurrent decreases) are presented.

[0196] In this correction example, firstly, in the state where the load current Iout = 105% (105% of the rated current) (refer to...) Figure 13 Under these conditions, the DC output voltage Vout was measured. Figure 12Step S41). Then, determine whether the DC output voltage Vout measured in this state is 95% or more of the rated voltage (Vout≥95%) (step S42). Here, if Vout≥95% (refer to step S42: Y, Figure 13 If the "○" symbol at Iout = 105% is found, then proceed to step S44 as described later.

[0197] On the other hand, if Vout < 95% (refer to step S42: N, Figure 13 The "×" sign at Iout = 105% is removed, and the estimated load current value Iout(est) is reduced (step S43). Specifically, the predetermined parameters (e.g., the capacitance value of the resonant capacitor Cr, the magnetizing inductance value Lm of the transformer 3, etc.) are reduced. After step S43, the process returns to the aforementioned step S41.

[0198] Secondly, in step S44 above, in the state where the load current Iout = 110% (110% of the rated current) is set (refer to...) Figure 13 Under these conditions, the DC output voltage Vout is measured. Then, it is determined whether the DC output voltage Vout measured under these conditions is less than 95% of the rated voltage (Vout < 95%) (step S45). Here, if Vout < 95% (refer to step S45: Y, Figure 13 The "○" symbol at Iout=110% in the text. Figure 12 The series of correction processes shown has ended.

[0199] On the other hand, if Vout ≥ 95% (refer to step S45: N, Figure 13 The "×" sign at Iout = 110% is removed, and the estimated load current value Iout(est) is increased as a correction (step S46). Specifically, the above-mentioned parameters (e.g., the capacitance value of the resonant capacitor Cr, the magnetizing inductance value Lm of the transformer 3, etc.) are increased as corrections. After step S46, the process returns to the aforementioned step S44.

[0200] (Functions and Effects)

[0201] In this modified example 1 of the switching power supply device 1B, the same effect can be obtained because it has essentially the same function as the switching power supply devices 1, 1A of the first and second embodiments.

[0202] Furthermore, particularly in this modified example 1, the load current Iout is estimated using predetermined parameters that are pre-corrected based on the measured value of the correspondence between the load current Iout and the DC output voltage Vout, as described below. In other words, because such pre-correction is performed, for example, during inspections such as when the switching power supply device 1B is shipped, variations in the estimated load current Iout caused by individual deviations of various components can be suppressed in advance, thereby improving manufacturing yield. Furthermore, as a result, further cost reduction of the switching power supply device 1B can be achieved.

[0203] [Variation Example 2]

[0204] (structure)

[0205] Figure 14 This is a circuit diagram showing a schematic structural example of the switching power supply device (switching power supply device 1C) of Modified Example 2.

[0206] Furthermore, similar to the first and second embodiments, the system comprising the DC input power supply 10 and the switching power supply device 1C corresponds to a specific example of the "power supply system" of the present invention.

[0207] In this modified example 2, the switching power supply device 1C is equivalent to the switching power supply devices 1, 1A, and 1B described above, except that a transformer 3C and a rectifier smoothing circuit 4C are provided to replace the transformer 3 and the rectifier smoothing circuit 4, respectively, while the other structures are the same.

[0208] Transformer 3C has one primary winding 31 and one secondary winding 32. That is, transformer 3 has two secondary windings 321 and 322, while transformer 3C has only one secondary winding 32. In this secondary winding 32, the first end is connected to connection point P7 in the rectifier and smoothing circuit 4C (described later), and the second end is connected to connection point P8 in the rectifier and smoothing circuit 4C.

[0209] The transformer 3C, like transformer 3, performs voltage transformation on the voltage (rectangular pulse wave voltage) generated by inverter circuit 2 and outputs AC voltage from the end of secondary winding 32. Furthermore, in this case, the degree of voltage transformation of DC output voltage Vout to DC input voltage Vin is determined by the turns ratio of primary winding 31 to secondary winding 32 and the aforementioned switching frequency fsw.

[0210] The rectifier-smoothing circuit 4C has four rectifier diodes 41-44 and one output smoothing capacitor Cout. Specifically, the rectifier-smoothing circuit 4C includes a rectifier circuit with rectifier diodes 41-44 and a smoothing circuit with output smoothing capacitor Cout. In other words, the rectifier-smoothing circuit 4C is equivalent to changing the structure of the rectifier circuit in the rectifier-smoothing circuit 4.

[0211] Furthermore, these four rectifier diodes 41 to 44 correspond to a specific example of the "multiple rectifier elements" of the present invention.

[0212] The rectifier circuit in this modified example 2 differs from the rectifier circuits in the first, second, and modified examples 1 (so-called "center-tapped" rectifier circuits) in that it is a so-called "bridge" rectifier circuit. That is, the cathodes of rectifier diodes 41 and 43 are each connected to the output line LO, and the anode of rectifier diode 41 is connected at connection point P7 to the cathode of rectifier diode 42 and the aforementioned first terminal of the secondary winding 32. Furthermore, the anodes of rectifier diodes 42 and 44 are each connected to the ground line LG, and the cathode of rectifier diode 44 is connected at connection point P8 to the anode of rectifier diode 43 and the aforementioned second terminal of the secondary winding 32.

[0213] In the rectifier smoothing circuit 4C with such a structure, similar to the rectifier smoothing circuit 4, the AC voltage output from the transformer 3C is rectified and output in the rectifier circuit which is configured to include rectifier diodes 41 to 44.

[0214] (Functions and Effects)

[0215] In the modified example 2 switching power supply device 1C with such a structure, the same effect can be obtained because it has essentially the same function as the switching power supply devices 1, 1A, 1B described above.

[0216] Furthermore, particularly in this modified example 2, since the rectifier circuit of the rectifier smoothing circuit 4C is a bridge rectifier circuit, the number of windings (the number of secondary windings) of the transformer 3C is reduced to one (secondary winding 32) compared to, for example, the first and second embodiments and modified example 1. As a result, it is possible to achieve miniaturization and low loss of the transformer 3C.

[0217] [Variations 3, 4]

[0218] (structure)

[0219] In the switching power supply devices (switching power supply devices 1D, 1E) of the modifications 3 and 4, the rectifier circuits in the rectifier smoothing circuits 4 and 4C described above are respectively equivalent to the so-called synchronous rectifier circuits described below in the first and second embodiments and in the modifications 1 and 2.

[0220] Specifically, Figure 15 This is a circuit diagram showing a schematic structural example of the switching power supply device (switching power supply device 1D) of Modified Example 3.

[0221] In this modified example 3, the switching power supply device 1D is equivalent to the switching power supply devices 1, 1A, 1B in the first and second embodiments and the modified example 1, except that a rectifier smoothing circuit 4D is provided instead of a rectifier smoothing circuit 4, and the other structures are the same.

[0222] In the synchronous rectifier circuit (rectifier smoothing circuit 4D) of this modified example 3, as follows Figure 15 As shown, the aforementioned rectifier diodes 41 and 42 are each configured as MOS-FETs (MOS transistors M9 and M10) serving as switching elements. Furthermore, control is performed in this synchronous rectification circuit to synchronize with the conduction period of the parasitic diodes of each MOS transistor M9 and M10, while these MOS transistors M9 and M10 themselves are also in a conducting state (performing synchronous rectification). Specifically, the drive circuit 5 of this modified example 3 uses drive signals SG9 and SG10 to control the on / off operation of each MOS transistor M9 and M10 (see reference). Figure 15 ).

[0223] in addition, Figure 16 This is a circuit diagram showing a schematic structural example of the switching power supply device (switching power supply device 1E) of Modified Example 4.

[0224] The switching power supply device 1E in this modified example 4 is equivalent to the switching power supply device 1C in modified example 2, except that a rectifier smoothing circuit 4E is provided instead of a rectifier smoothing circuit 4C, and the rest of the structure is the same.

[0225] In the synchronous rectifier circuit (rectifier smoothing circuit 4E) of this modified example 4, as follows Figure 16 As shown, the rectifier diodes 41 to 44 described in Modification 2 are each configured as MOS-FETs (MOS transistors M11 to M14) serving as switching elements. Furthermore, in the synchronous rectification circuit of Modification 4, control is performed in the same manner as in Modification 3 to synchronize with the conduction period of the parasitic diodes of each MOS transistor M11 to M14, and these MOS transistors M11 to M14 themselves are also in a conducting state (performing synchronous rectification). Specifically, the drive circuit 5 of Modification 4 uses drive signals SG11 to SG14 to control the on / off operation of each MOS transistor M11 to M14 (see reference). Figure 16 ).

[0226] Furthermore, similar to the first and second embodiments, the system having a DC input power supply 10 and these switching power supply devices 1D or 1E corresponds to a specific example of the "power supply system" of the present invention.

[0227] (Functions and Effects)

[0228] In the modified examples 3 and 4 of the switching power supply device 1D and 1E, which are constructed with such a structure, the same effect can be obtained because they are essentially the same as those of the switching power supply devices 1, 1A, 1B and 1C described above.

[0229] Furthermore, especially in these variations 3 and 4, since the multiple rectifier elements of the rectifier circuit are each composed of a switching element, and the rectifier circuit is a synchronous rectifier circuit, as described below. That is to say, because the conduction loss during rectification is reduced by such a synchronous rectifier circuit, miniaturization and low loss of the rectifier circuit can be achieved. In addition to the MOS-FET mentioned above, other switching elements that can be used include, for example, HEMT (High Electron Mobility Transistor) = HFET (Heterostructure Field-Effect Transistor), IGBTs with diodes attached in parallel, or bipolar transistors.

[0230] <4. Other variations>

[0231] The present invention has been described above with examples of embodiments and variations, but the present invention is not limited to these embodiments and various changes can be made.

[0232] For example, although the inverter circuit structure has been specifically illustrated in the above embodiments, the inverter circuit is not limited to the examples described above. Other structures may be used as the inverter circuit. Specifically, the arrangement of the resonant inductor Lr, resonant capacitor Cr, and primary winding 31 connected in series is not limited to the arrangement described in the embodiments, and may be in any non-specific order. Furthermore, in the above embodiments, an example has been described in which the resonant inductor Lr, resonant capacitor Cr, and primary winding 31 are each arranged between the connection point P1 and the primary side low-voltage line L1L (input terminal T2), but this is not a limitation. That is, for example, the resonant inductor Lr, resonant capacitor Cr, and primary winding 31 may also be arranged between the connection point P1 and the primary side high-voltage line L1H (input terminal T1).

[0233] Furthermore, although the structure of a transformer (primary winding and secondary winding) has been specifically illustrated in the above embodiments, the invention is not limited to the examples described above. For example, other structures may be used as transformers (primary winding and secondary winding).

[0234] Furthermore, although the structure of the rectifier-smoothing circuit (rectifier circuit and smoothing circuit) has been specifically illustrated in the above embodiments, it is not limited to the examples described above. For example, other structures may be used as the rectifier-smoothing circuit (rectifier circuit and smoothing circuit).

[0235] Furthermore, although the above embodiments specifically illustrate a method for controlling the operation of each switching element by a drive circuit (switching drive), the method is not limited to the examples described above, and other methods can be used as the switching drive method. Additionally, the method for estimating the load current Iout is not limited to the methods described in the above embodiments, and other methods can be used. Moreover, although the structure of the load current estimation unit (detection unit and control unit) is specifically illustrated in the above embodiments, the method is not limited to the examples described above, and other structures can be used as the load current estimation unit (detection unit and control unit).

[0236] Furthermore, although a DC-DC converter has been described as an example of the switching power supply device of the present invention in the above embodiments, the present invention can also be applied to other types of switching power supply devices, such as AC-DC converters.

[0237] Furthermore, the above-mentioned structural examples can be applied in any combination.

Claims

1. A switching power supply device, comprising: Input terminal pair, input voltage; Output terminal pair, output voltage; A transformer has a primary winding and a secondary winding; An inverter circuit is disposed between the input terminal pair and the primary winding, and is configured to include a first switching element and a second switching element, a resonant inductor and a resonant capacitor. A rectifier smoothing circuit is disposed between the output terminal pair and the secondary winding, and is configured to include a rectifier circuit having multiple rectifier elements and a smoothing circuit having a smoothing capacitor. as well as The control unit controls the switching actions of the first and second switching elements of the inverter circuit, respectively. The first switching element and the second switching element are connected in series between a pair of connection lines that are individually connected to the input terminal pair. The resonant inductor, the resonant capacitor, and the primary winding are connected in series in an unspecified order between the connection point of the first and second switching elements and the input terminal of one of the input terminal pairs. The control unit controls the switching action based on the voltage between the two ends of the resonant capacitor and the voltage between the two ends of the primary winding. Equipped with a load current estimation unit, The load current estimation unit estimates the load current flowing through the load powered by the output voltage, and includes the control unit. The load current estimation unit calculates the primary-side converted value of the load current based on the resonant inductor current flowing through the resonant inductor and the transformer excitation current flowing through the primary winding, and then calculates the estimated value of the load current based on the primary-side converted value. The control unit uses the estimated value of the load current to control the switching action, thereby controlling the value of the load current. The load current estimation unit uses the difference between the current-time product of the resonant inductor current and the transformer excitation current to obtain an estimated value of the load current. The load current estimation unit modifies the method for estimating the load current using the current-time product difference based on the relationship between the switching frequencies of the first and second switching elements and the resonant frequency of the resonant operation using the resonant inductor and the resonant capacitor. When the switching frequency is higher than the resonant frequency, the load current estimation unit estimates the load current using the difference in the current-time product based on the voltage between the two ends of the resonant capacitor and the voltage between the two ends of the primary winding. Furthermore, when the switching frequency is lower than the resonant frequency, the load current is estimated using the current-time product difference based on the voltage between the two ends of the resonant capacitor, the voltage between the two ends of the primary winding, and the switching voltage corresponding to the potential of the connection point.

2. The switching power supply device according to claim 1, wherein, The load current estimation unit uses the current phase difference between the resonant inductor current and the transformer excitation current to obtain an estimated value of the load current.

3. The switching power supply device according to claim 2, wherein, The load current estimation unit calculates the current phase difference based on the voltage between the two ends of the resonant capacitor and the voltage between the two ends of the primary winding.

4. The switching power supply device according to claim 1, wherein, When the switching frequency is lower than the resonant frequency, the load current estimation unit considers the asymmetry of the positive and negative values ​​of the transformer excitation current and estimates the load current using the current-time product difference.

5. The switching power supply device according to any one of claims 1 to 4, wherein, The load current estimation unit estimates the load current using predetermined parameters that are pre-corrected based on the measured values ​​of the correspondence between the load current and the output voltage.

6. The switching power supply device according to claim 5, wherein, The specified parameters are the capacitance value of the resonant capacitor, the magnetizing inductance value of the transformer, or the detection value in the load current estimation unit.

7. The switching power supply device according to any one of claims 1 to 4, wherein, The resonant inductor is formed by the leakage inductance of the transformer.

8. The switching power supply device according to any one of claims 1 to 4, wherein, The first switching element and the second switching element are each composed of a MOS-FET.

9. The switching power supply device according to any one of claims 1 to 4, wherein, The rectifier circuit is a center-tapped rectifier circuit.

10. The switching power supply device according to any one of claims 1 to 4, wherein, The rectifier circuit is a bridge rectifier circuit.

11. The switching power supply device according to any one of claims 1 to 4, wherein, The plurality of rectifier elements are each composed of a switching element, and the rectifier circuit is a synchronous rectifier circuit.

12. A power supply system, comprising: The switching power supply device according to any one of claims 1 to 11; and A power supply provides the input voltage to the input terminals.

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