A method for calculating iron loss of an axial flux permanent magnet motor

Abstract

This invention discloses a method for calculating iron loss in an axial flux permanent magnet memory motor, aiming to solve the technical problem of calculation accuracy deviation in axial flux motors or some special motors. It includes: firstly, dividing the core structure of the axial flux motor into layers axially, rings radially, and blocks circumferentially; then, selecting representative points to plot the magnetic flux density variation law; performing harmonic decomposition of the magnetic flux density in three dimensions to obtain the peak values ​​of each order; and finally, calculating iron loss using the proposed model. This invention can consider the influence of multiple factors, including harmonic effects, the influence of circumferential rotational magnetization, small loops in the core hysteresis, eddy current effects, temperature field effects, and stress effects, etc., making it more accurate than traditional calculation methods. It is applicable to calculating iron loss of axial flux motors under different operating conditions, laying the foundation for further efficiency improvement and enhancement.

Description

Technical Field

[0001] This invention relates to a method for calculating the iron loss of an axial flux permanent magnet motor, belonging to the technical field of axial flux permanent magnet memory motors. Background Technology

[0002] With increasing emphasis on new energy sources, improving energy efficiency has become a pressing issue. Axial flux permanent magnet memory motors, characterized by high power density, short axial dimensions, adjustable air gap magnetic field, and ease of installation, possess promising application prospects and are expected to be widely used in aerospace, automotive, and intelligent manufacturing industries.

[0003] In recent years, axial flux permanent magnet motors have developed rapidly, but many axial motors, including those with field-tuning memory, are inefficient in applications. Besides copper losses accounting for a large proportion under no-load conditions, iron losses are gradually becoming a significant factor in high-power motors. Considering that there is currently no highly accurate iron loss measurement device, accurate calculation methods are needed for its analysis and calculation.

[0004] The earliest calculation of iron loss can be traced back to 1924 when Jordan proposed separating core losses and established an iron loss separation model. In 1988, Bertotti further expanded this model into a three-part separation calculation formula, expressed as follows:

[0005]

[0006] This model has good accuracy in calculating low-frequency core losses in a single magnetization direction, but its deviation is large when applied to rotating motors. Many experts and scholars have conducted extensive research on the iron losses of traditional radial motors, but research on permanent magnet memory motors with axial flux is extremely limited.

[0007] Currently, patent applications related to calculating motor iron loss, such as patent application number 202210510042.4, disclose a novel method for calculating the iron loss of a stator permanent magnet motor. This method considers the motor iron loss under the influence of multiple factors, including: elliptical rotational magnetization, DC bias, small hysteresis loop phenomenon, and PWM harmonic current. This model has high calculation accuracy and is simple to calculate, and is applicable to traditional radial motors under different operating conditions. However, it cannot be practically applied to the calculation of iron loss in special topology motors such as axial flux memory motors. Summary of the Invention

[0008] The purpose of this invention is to overcome the shortcomings of the prior art and provide a method for calculating the iron loss of an axial flux permanent magnet motor. This method solves the technical problem of accuracy deviation in the calculation of iron loss of axial flux permanent magnet motors by existing iron loss calculation methods. It is designed for axial flux permanent magnet motors and fully considers the influencing factors in the calculation process of motor iron loss, resulting in high calculation accuracy and close resemblance to actual values.

[0009] To achieve the above objectives, the present invention is implemented using the following technical solution:

[0010] This invention provides a method for calculating the iron loss of an axial flux permanent magnet motor, including:

[0011] The waveform of the change in magnetic flux density of the target permanent magnet motor is obtained, and the core of each part of the motor is divided into multiple core units according to the change in magnetic flux density.

[0012] Select a representative point of magnetic flux density in each of the divided iron core units;

[0013] The magnetic field of the motor was simulated using electromagnetic analysis software, and a three-dimensional magnetic flux density distribution map of each representative point was obtained.

[0014] The magnetic flux density at each representative point is orthogonally decomposed into three magnetic flux density vectors in three dimensions. Using the principle of harmonic analysis, the magnetic flux density in each direction is decomposed into the fundamental wave and its harmonics of each order, thereby obtaining the peak value B of the magnetic flux density of each order. m Then, the magnetic flux density peaks in each direction are superimposed to obtain the magnetic flux density amplitude B. mi (Z,R,θ);

[0015] The calculated magnetic flux density amplitude B mi Substituting (Z,R,θ) into the corrected iron loss calculation model yields the iron loss per unit mass, and the total loss of the motor is calculated based on the motor density. The corrected iron loss calculation model is the calculation model obtained by correcting the initial iron loss calculation model for temperature, stress, eddy current effect, and hysteresis loss.

[0016] Furthermore, the method of dividing the core of each part of the motor into multiple core units based on the changes and amplitudes of magnetic flux density is as follows: layering in the axial direction, dividing into rings in the radial direction, and dividing into blocks in the circumferential direction to form multiple core units.

[0017] Furthermore, the method for selecting representative points of magnetic flux density is to choose the geometric center point in each divided core element as the representative point for calculating the magnetic flux density of that core element. Then, the time-stepping finite element method can be used to simulate the magnetic field of the motor, obtaining a three-dimensional diagram of the magnetic flux density distribution at each point.

[0018] Furthermore, the magnetic flux density in each direction is decomposed into fundamental waves and their harmonics of various orders using the principle of harmonic analysis, thereby obtaining the peak magnetic flux density B of each order. mThen, the magnetic flux density peaks in each direction are superimposed to obtain the magnetic flux density amplitude B. mi (Z,R,θ) includes:

[0019]

[0020]

[0021]

[0022] In the formula: B(t) z This represents the extracted magnetic flux density waveform in the Z direction as a function of time; the calculation methods for the radial and tangential directions are similar; c0 represents the DC component; c i This represents the amplitude of each order of magnetic flux density. The phase angle is represented by i = 1, 2, ..., which represent different orders of magnetic flux density. When i = 1, it is the fundamental magnetic flux density after decomposition; when i = 2, 3, 4, ..., it is the harmonic magnetic flux density after decomposition. B mi (Z,R,θ) represents the magnetic flux density amplitude after superposition in all directions. The peak values ​​of the magnetic flux density in the Z direction are for each order. These represent the peak values ​​of the magnetic flux density at each radial order. denoted as the peak magnetic flux density of each order in the tangential direction, and c as the coefficient of the corresponding loss term.

[0023] Furthermore, the corrected iron loss calculation model is as follows:

[0024]

[0025] Among them, P h For hysteresis ratio loss, P c For eddy current ratio loss, P a For abnormal specific loss, i is the harmonic order, f is the fundamental frequency of the alternating magnetic field, l is the highest harmonic order considered, K(B) is the hysteresis loss correction coefficient, and K h To account for the hysteresis loss coefficient under stress, K c K is the eddy current loss coefficient. a K is the additional loss factor. c (t) is the correction factor considering the effect of temperature, K c (f) is the eddy current correction coefficient; B mi (Z,R,θ) represents the magnetic flux density amplitude after superposition in all directions.

[0026] For motors with asymmetrical magnetic pole structures, the magnetic flux frequencies on the stator and rotor are inconsistent, which can be determined using the following formula:

[0027]

[0028]

[0029]

[0030] In the formula: f s f rm f rc These represent the stator magnetic flux density frequency, the rotor combined magnetic flux density frequency, and the component magnetic flux density frequency, respectively; n is the motor speed; p r p is the number of rotor poles. s This is the stator pole number.

[0031] Furthermore, the iron loss calculation method described in this application takes into account the mechanical stress on the silicon steel sheet, which significantly increases the hysteresis loss term of the silicon steel sheet. Therefore, its coefficient is stress-corrected. The hysteresis loss coefficient K considering the influence of stress... h The calculation formula is:

[0032] K h =K h0 +K σ ;

[0033] Among them, K h0 To disregard the hysteresis loss coefficient caused by stress, K σ The hysteresis loss coefficient is increased to account for stress variations.

[0034] Furthermore, the method also includes first fitting a number of parameters to the iron loss coefficient, then optimizing the fitted iron loss coefficient using a sparrow search algorithm to obtain the iron loss coefficient at different frequencies; and optimizing the hysteresis loss coefficient K under rated operating conditions according to this method. h0 Eddy current loss coefficient K c and additional loss coefficient K a They are 0.027, 1.052e-4, and 1.561e-5, respectively.

[0035] Furthermore, the iron loss calculation method described in this application takes into account the temperature field effect caused by the increase in material resistivity due to the temperature rise during motor operation. Therefore, its coefficients are temperature-corrected. The correction coefficient K that considers the temperature effect is... c The formula for calculating (t) is:

[0036]

[0037] Where d is the thickness of the silicon steel sheet, ρ is the density of the silicon steel sheet, and k t K is the temperature change coefficient, Δt is the temperature change, and k is the temperature change coefficient. c0 ρ is the eddy current loss coefficient at the reference temperature. e0 This refers to the resistivity of the metal at room temperature.

[0038] ρ e(t) represents the resistivity of the metal at different temperatures, and its calculation formula is:

[0039] ρ e (t)=ρ e0 (1+k t ·Δt).

[0040] Furthermore, the iron loss calculation method described in this application takes into account the influence of the eddy current effect generated in the silicon steel lamination on the iron loss of the motor, and uses the eddy current correction coefficient to correct the eddy current effect, and calculates the eddy current correction coefficient K. c (f) The eddy current correction function K c The formula for calculating (f) is:

[0041]

[0042] Among them, K c denoted as eddy current loss coefficient, and f is the fundamental frequency of the alternating magnetic field.

[0043] D is the loss constant considering the skin effect, and its calculation formula is:

[0044]

[0045] Where: μ is the average magnetic permeability, σ f The conductivity of silicon steel sheet.

[0046] Furthermore, the iron core experiences alternating magnetic fields. Due to the influence of higher harmonics, the magnetic flux waveform is distorted, resulting in small fluctuations. This corresponds to small hysteresis loops in the hysteresis loop, thus requiring hysteresis loss correction for its coefficient. The formula for calculating the hysteresis loss correction coefficient K(B) is as follows:

[0047]

[0048] Among them, t m Take a constant of 0.65, ΔB λ α represents the magnetic flux density amplitude produced by the hysteresis loop effect, S is the total number of hysteresis loops in one period, and λ is the hysteresis loop number; h This is the correction constant for hysteresis loss, which is typically 1 at lower frequencies; B m The peak value of the magnetic flux density for each order is denoted as .

[0049] Compared with the prior art, the beneficial effects achieved by the present invention are as follows:

[0050] The iron loss calculation method described in this invention is designed for axial flux permanent magnet motors. It divides the motor core in multiple dimensions and fully considers the influencing factors in the iron loss calculation process. The calculation accuracy is high and close to the actual value. This method has wide applications and is suitable for calculating the iron loss of permanent magnet motors with different topologies.

[0051] This invention takes into account the influence of various factors, including: the influence of high-order magnetic flux density harmonics, the influence of axial alternating magnetization, small loops of core hysteresis, skin effect of core eddy currents, influence of temperature field, and stress, etc., and provides technical guidance for the design and optimization of axial flux permanent magnet memory motors.

[0052] The iron loss calculation method described in this invention has a wide range of applications and is suitable for various types of motors and application scenarios. For example, when the motor excitation is PWM controlled, the generated magnetic flux density harmonic content is relatively large, and this calculation method can effectively account for the influence of higher-order magnetic flux density harmonics. This invention considers magnetic flux density in three dimensions, and the magnetic flux density distribution of the motor and the magnitude of waveform distortion can be analyzed using the spatial rotating magnetization trajectory. Because this invention considers the skin effect of the iron core, it can be applied to the iron loss calculation of high-speed rotating motors. This invention considers the influence of hysteresis effect, and can be applied to low-speed, high-torque motors. Because this invention considers temperature and stress fields, it can be applied to the iron loss calculation of motors under various special working environments, such as deep water and high-temperature steam engines. Attached Figure Description

[0053] Figure 1 This is a flowchart of a method for calculating the iron loss of an axial flux permanent magnet motor provided in an embodiment of the present invention;

[0054] Figure 2 This is a schematic diagram showing the core division results of the motor as described in the embodiment;

[0055] Figure 3 This is a tangential magnetic flux density harmonic decomposition diagram at a certain point on the lower stator tooth as described in the embodiment;

[0056] Figure 4 This is an axial magnetic flux density harmonic decomposition diagram of a certain point on the lower stator tooth as described in the embodiment;

[0057] Figure 5 These are the four magnetic flux density representative points of the N pole of the B phase of the lower stator as described in the embodiment;

[0058] Figure 6 yes Figure 5 Magnetic flux density distribution at point A in the middle;

[0059] Figure 7 yes Figure 5 Magnetic flux density distribution at point B;

[0060] Figure 8 yes Figure 5 Magnetic flux density distribution at point C;

[0061] Figure 9 yes Figure 5 Magnetic flux density distribution at point D;

[0062] Figure 10 This refers to the typical magnetic flux density waveform and the small hysteresis loop in its hysteresis loop as described in the embodiment.

[0063] Figure 11 This is the lower stator magnetic flux density vector diagram described in the embodiment;

[0064] Figure 12 This is a magnetic flux density trajectory diagram of different representative points on the lower stator as described in the embodiment.

[0065] In the diagram: 1. Lower stator tooth tip; 2. Lower stator tooth-yoke junction; 3. Lower stator yoke; 4. Rotor tooth tip; 5. Rotor tooth middle; 6. Rotor tooth root; 7. Upper stator inner ring; 8. Upper stator middle ring; 9. Upper stator outer ring; A is a representative point of the lower stator yoke; B is a representative point of the lower stator tooth-yoke junction; C is a representative point of the lower stator tooth middle; D is a representative point of the lower stator tooth tip. Detailed Implementation

[0066] The technical solution of the present invention will be described in detail below with reference to the accompanying drawings and specific embodiments. It should be understood that the embodiments and specific features in the embodiments are detailed descriptions of the technical solution of the present application, rather than limitations thereof. In the absence of conflict, the embodiments and technical features in the embodiments can be combined with each other.

[0067] Example 1

[0068] Figure 1 This flowchart illustrates a method for calculating iron losses in an axial flux permanent magnet motor, as provided in this embodiment. This flowchart merely shows the logical sequence of the method described in this embodiment; however, in other possible embodiments of the invention, different methods may be used, provided there are no conflicts. Figure 1 Complete the steps shown or described in the order indicated.

[0069] The iron loss calculation method for axial flux permanent magnet motors provided in this embodiment can be applied to a terminal and executed by an iron loss calculation device. This device can be implemented in software and / or hardware and can be integrated into the terminal, such as any smartphone, tablet, or computer device with communication capabilities. See also... Figure 1 The method in this embodiment specifically includes the following steps.

[0070] Step 1: Model Partitioning

[0071] The motor structure described in this embodiment is as follows: Figure 2As shown, the upper stator disk of the motor is made of radially magnetized NdFeB35 and soft magnetic material spliced ​​together. The soft magnetic material is bonded with a magnetic adjustment block made of AlNiCo, and the magnetic adjustment block is wound with a magnetic adjustment winding. The rotor is made of 5 fan-shaped iron cores spliced ​​circumferentially on a non-magnetic ring. The lower stator has a concentrated armature winding.

[0072] The magnetic field of the target permanent magnet motor was simulated using the time-stepping finite element method to obtain the waveform of the motor's magnetic flux density changing over time. Considering the amplitude and distribution of the motor's magnetic flux density, and the fact that the lower stator, rotor, and upper stator disks all contain iron cores, each part was divided into layers axially, rings radially, and blocks circumferentially. Each of these blocks is considered a core unit. Specifically, the lower stator tooth tip 1, the lower stator tooth yoke junction 2, and the lower stator yoke 3 are divided into an outer ring, a middle ring, and an inner ring in the radial direction, respectively. In the axial direction, the lower stator tooth tip 1 is divided into two layers, and the lower stator yoke 3 is divided into five layers. The rotor tooth tip 4, the rotor tooth middle 5, and the rotor tooth root 6 are all divided into two layers in the axial direction. The rotor tooth is divided into three rings at equal intervals in the radial direction, and into equal fan-shaped blocks with central angles in the circumferential direction. The blocks numbered 10 and 11 on the rotor are rotor skew poles. The upper stator is divided into an inner ring 7, a middle ring 8, and an outer ring 9 at equal intervals in the radial direction. Considering the thickness of the soft magnetic side, the soft magnetic side is not divided in the axial direction, but is divided into an inner ring and a middle ring in the radial direction.

[0073] Step 2: Select representative points

[0074] In such Figure 2 In each of the divided core units shown, the geometric center point is selected as the representative point for calculating the magnetic flux density of that core unit. In this embodiment, a total of 46 representative points are selected, including 21 points in the lower stator yoke and teeth, 22 points in the upper and lower layers of the rotor, and 3 points in the soft magnetic material.

[0075] Step 3: Extract the magnetic flux density of representative points

[0076] Using the time-stepping finite element method, the waveforms of the magnetic flux density at representative points in the iron core in the axial, radial, and circumferential directions as a function of time are calculated in Maxwell's two-dimensional transient electromagnetic field, and three-dimensional images of the magnetic flux density distribution at each representative point are extracted.

[0077] Step 4: Extract the peak value Bm of the decomposed magnetic flux density.

[0078] like Figure 3 As shown in the figure, the tangential component of the magnetic flux density at a certain point of the lower stator tooth is decomposed using Fourier decomposition. The magnetic flux density waveform is decomposed to the 5th harmonic. The decomposition to the 3rd harmonic is more obvious, while the harmonic amplitude is very small when it reaches the 4th or 5th harmonic.

[0079] like Figure 4 As shown, this is a Fourier decomposition waveform of the axial magnetic flux density at a certain point on the lower stator tooth, with the axial magnetic flux density component B.z In axial flux permanent magnet memory motors, the magnetic flux flows from NdFeB on both sides into the soft magnetic material, then to the AlNiCo material it connects to, passes through the double air gap and the central iron core rotor, enters the lower stator B-phase N pole, and then disperses into the lower stator yoke back to NdFeB. Therefore, the negative sequence component accounts for the majority of the magnetic flux density. The magnetic flux density waveform shows significant second harmonics, while higher-order harmonics have smaller amplitudes.

[0080] like Figure 5 As shown, four representative points of magnetic flux density of the N pole of phase B of the lower stator were selected and three-dimensional magnetic flux density decomposition was performed respectively. Among them, A is the representative point of the lower stator yoke; B is the representative point of the lower stator tooth-yoke junction; C is the representative point of the middle part of the lower stator tooth; and D is the representative point of the lower stator tooth tip. The magnetic flux density distribution diagram is shown in the figure. Figures 6 to 9 As shown, the closer the magnetic flux density position is to the tip of the lower stator tooth, the amplitude of each component of magnetic flux density gradually increases, and the distortion of the magnetic flux density waveform becomes more obvious. The magnetic flux density waveform at the tooth tip position becomes a flat-topped wave.

[0081] Therefore, the iron loss calculation method described in this invention considers the influence of higher-order magnetic flux density harmonics and takes into account magnetic flux density in three dimensions. The specific method is as follows:

[0082] The magnetic flux density at each representative point is orthogonally decomposed into three magnetic flux density vectors in three dimensions. Using the principle of harmonic analysis, the magnetic flux density in each direction is decomposed into the fundamental wave and its harmonics of each order, thereby obtaining the peak value B of the magnetic flux density of each order. m Then, the magnetic flux density peaks in each direction are superimposed to obtain the magnetic flux density amplitude B. mi (Z,R,θ). The calculation method is as follows:

[0083]

[0084]

[0085]

[0086] In the formula: B(t) z This represents the extracted magnetic flux density waveform in the Z direction as a function of time; the calculation methods for the radial and tangential directions are similar; c0 represents the DC component; c i This represents the amplitude of each order of magnetic flux density. The phase angle is represented by i = 1, 2, ..., which represent different orders of magnetic flux density. When i = 1, it is the fundamental magnetic flux density after decomposition; when i = 2, 3, 4, ..., it is the harmonic magnetic flux density after decomposition. B mi (Z,R,θ) represents the magnetic flux density amplitude after superposition in all directions. The peak values ​​of the magnetic flux density in the Z direction are for each order. These represent the peak values ​​of the magnetic flux density at each radial order. denoted as the peak magnetic flux density of each order in the tangential direction, and c as the coefficient of the corresponding loss term.

[0087] Step 5: Calculate iron loss using the model

[0088] The calculated magnetic flux density amplitude B mi Substituting (Z,R,θ) into the corrected iron loss calculation model yields the iron loss per unit mass. Finally, the total loss of the motor is calculated based on the motor density.

[0089] The corrected iron loss calculation model is as follows:

[0090]

[0091] Among them, P h For hysteresis ratio loss, P c For eddy current ratio loss, P a For abnormal specific loss, i is the harmonic order, f is the fundamental frequency of the alternating magnetic field, l is the highest harmonic order considered, K(B) is the hysteresis loss correction coefficient, and K h To account for the hysteresis loss coefficient under stress, K c K is the eddy current loss coefficient. a K is the additional loss factor. c (t) is the correction factor considering the effect of temperature, K c (f) is the eddy current correction coefficient; B mi (Z,R,θ) represents the magnetic flux density amplitude after superposition in all directions.

[0092] For motors with asymmetrical magnetic pole structures, the magnetic flux frequencies on the stator and rotor are inconsistent, which can be determined using the following formula:

[0093]

[0094]

[0095]

[0096] In the formula: f s f rm f rc These represent the stator magnetic flux density frequency, the rotor combined magnetic flux density frequency, and the component magnetic flux density frequency, respectively; n is the motor speed; p r p is the number of rotor poles. s This is the stator pole number.

[0097] like Figure 10 As shown, the iron core experiences alternating magnetic fields. Due to the influence of higher harmonics, the magnetic flux waveform is distorted, resulting in small fluctuations. This corresponds to the formation of small hysteresis loops within the hysteresis loop. For example... Figure 11As shown, this is a magnetic flux density vector diagram of the lower stator section. The magnetic vector is dispersed from the B-phase N pole into the A-phase S pole and C-phase S pole on both sides, and finally returns to NdFeB through the double-layer air gap and the rotor. Figure 12 As shown, spatial magnetic flux density trajectory diagrams corresponding to each point are selected. Since the harmonic content in the air gap gradually increases as it gets closer to the air gap and affects the magnetic flux density at the lower stator, the ellipticity of the magnetic flux density trajectory gradually decreases and the distortion becomes more and more obvious as the axial position gradually increases.

[0098] Therefore, the iron loss calculation method described in this embodiment takes into account the influence of various factors, including harmonic effects, the effects of circumferential rotational magnetization and axial alternating magnetization, small loops of core hysteresis, eddy current skin effect, temperature field effects, and stress effects, and corrects various loss coefficients.

[0099] The formula for calculating the hysteresis loss correction coefficient K(B) is as follows:

[0100]

[0101] Among them, t m Take a constant of 0.65, ΔB λ α represents the magnetic flux density amplitude produced by the hysteresis loop effect, S is the total number of hysteresis loops in one period, and λ is the hysteresis loop number; h This is the correction constant for hysteresis loss, which is typically 1 at lower frequencies; B m The peak value of the magnetic flux density for each order is denoted as .

[0102] The iron loss calculation method described in this embodiment takes into account the mechanical stress on the silicon steel sheet, which significantly increases the hysteresis loss term. Therefore, its coefficient is stress-corrected. The hysteresis loss coefficient K considering the stress effect is... h The calculation formula is:

[0103] K h =K h0 +K σ ;

[0104] Among them, K h0 To disregard the hysteresis loss coefficient caused by stress, K σ The hysteresis loss coefficient is increased to account for stress variations.

[0105] The iron loss calculation method described in this embodiment takes into account the temperature field effect caused by the increase in material resistivity due to the temperature rise during motor operation. Therefore, its coefficients are corrected for temperature. The correction coefficient K that considers the temperature effect is... c The formula for calculating (t) is:

[0106]

[0107] Where d is the thickness of the silicon steel sheet, ρ is the density of the silicon steel sheet, and k t K is the temperature change coefficient, Δt is the temperature change, and k is the temperature change coefficient. c0 ρ is the eddy current loss coefficient at the reference temperature. e0 This refers to the resistivity of the metal at room temperature.

[0108] ρ e (t) represents the resistivity of the metal at different temperatures, and its calculation formula is:

[0109] ρ e (t)=ρ e0 (1+k t ·Δt).

[0110] The iron loss calculation method described in this embodiment takes into account the influence of the eddy current effect generated in the silicon steel lamination on the iron loss of the motor. The eddy current effect is corrected using an eddy current correction coefficient, and the eddy current correction coefficient K is calculated. c (f) The eddy current correction function K c The formula for calculating (f) is:

[0111]

[0112] Among them, K c denoted as eddy current loss coefficient, and f is the fundamental frequency of the alternating magnetic field.

[0113] D is the loss constant considering the skin effect, and its calculation formula is:

[0114]

[0115] Where: μ is the average magnetic permeability, σ f The conductivity of silicon steel sheet.

[0116] The method described in this embodiment further includes first fitting a number of parameters to the iron loss coefficient, then optimizing the fitted iron loss coefficient using a sparrow search algorithm to obtain the iron loss coefficient at different frequencies; and then optimizing the hysteresis loss coefficient K under rated operating conditions according to this method. h0 Eddy current loss coefficient K c and additional loss coefficient K a They are 0.027, 1.052e-4, and 1.561e-5, respectively.

[0117] Those skilled in the art will understand that embodiments of this application can be provided as methods, systems, or computer program products. Therefore, this application can take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, this application can take the form of a computer program product embodied on one or more computer-usable storage media (including but not limited to disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code.

[0118] This application is described with reference to flowchart illustrations and / or block diagrams of methods, apparatus (systems), and computer program products according to embodiments of this application. It will be understood that each block of the flowchart illustrations and / or block diagrams, and combinations of blocks in the flowchart illustrations and / or block diagrams, can be implemented by computer program instructions. These computer program instructions can be provided to a processor of a general-purpose computer, special-purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, generate instructions for implementing the flowchart... Figure 1 One or more processes and / or boxes Figure 1 A device that provides the functions specified in one or more boxes.

[0119] These computer program instructions may also be stored in a computer-readable storage medium that can direct a computer or other programmable data processing device to function in a particular manner, such that the instructions stored in the computer-readable storage medium produce an article of manufacture including instruction means, which are implemented in a process Figure 1 One or more processes and / or boxes Figure 1 The function specified in one or more boxes.

[0120] These computer program instructions may also be loaded onto a computer or other programmable data processing equipment to cause a series of operational steps to be performed on the computer or other programmable equipment to produce a computer-implemented process, thereby providing instructions that execute on the computer or other programmable equipment for implementing the process. Figure 1 One or more processes and / or boxes Figure 1 The steps of the function specified in one or more boxes.

[0121] The above description is only a preferred embodiment of the present invention. It should be noted that for those skilled in the art, several improvements and modifications can be made without departing from the technical principles of the present invention, and these improvements and modifications should also be considered within the scope of protection of the present invention.

Claims

1. A method for calculating the iron loss of an axial flux permanent magnet motor, characterized in that, include: The waveform of the change in magnetic flux density of the target permanent magnet motor is obtained, and the core of each part of the motor is divided into multiple core units according to the change in magnetic flux density. Select a representative point of magnetic flux density in each of the divided iron core units; The magnetic field of the motor was simulated using electromagnetic analysis software, and a three-dimensional magnetic flux density distribution map of each representative point was obtained. The magnetic flux density at each representative point is orthogonally decomposed into three magnetic flux density vectors in three dimensions. Using the principle of harmonic analysis, the magnetic flux density in each direction is decomposed into the fundamental wave and its harmonics of each order, thereby obtaining the peak value B of the magnetic flux density of each order. m Then, the magnetic flux density peaks in each direction are superimposed to obtain the magnetic flux density amplitude. ; The calculated magnetic flux density amplitude The iron loss per unit mass is obtained by substituting the corrected iron loss calculation model into the motor, and the total loss of the motor is calculated based on the motor density. The corrected iron loss calculation model is the calculation model obtained by correcting the initial iron loss calculation model for temperature, stress, eddy current effect, and hysteresis loss. The method of dividing the core of each part of the motor into multiple core units based on the changes and amplitudes of magnetic flux density is as follows: layering in the axial direction, dividing into rings in the radial direction, and dividing into blocks in the circumferential direction to form multiple core units; Using the principle of harmonic analysis, the magnetic flux density in each direction is decomposed into fundamental waves and their harmonics of various orders, thereby obtaining the peak magnetic flux density B of each order. m Then, the magnetic flux density peaks in each direction are superimposed to obtain the magnetic flux density amplitude. ,include: ； ； ； In the formula: The extracted magnetic flux density waveform in the Z direction varies with time; the calculation methods for the radial and tangential directions are similar. Indicates the DC component; Indicates the phase angle. Representing different orders of magnetic flux density, The time is the fundamental magnetic flux density after decomposition. The harmonic magnetic flux density after decomposition. This represents the magnitude of the magnetic flux density after superposition in all directions. The peak values ​​of the magnetic flux density in the Z direction are for each order. These represent the peak values ​​of the magnetic flux density at each radial order. Let c be the peak magnetic flux density of each order in the tangential direction, and c be the coefficient of the corresponding loss term; the corrected iron loss calculation model is as follows: ； Among them, P h For hysteresis ratio loss, P c For eddy current ratio loss, P a For abnormal specific loss, i is the harmonic order, f is the fundamental frequency of the alternating magnetic field, l is the highest harmonic order considered, K(B) is the hysteresis loss correction coefficient, and K h To account for the hysteresis loss coefficient under stress, K a K is the additional loss factor. c (t) is the correction factor considering the effect of temperature, K c (f) represents the eddy current correction coefficient; This represents the magnetic flux density amplitude after superposition in all directions.

2. The method for calculating iron loss of an axial flux permanent magnet motor according to claim 1, characterized in that, The hysteresis loss coefficient K considering the influence of stress h The calculation formula is: ； Among them, K h0 To disregard the hysteresis loss coefficient caused by stress, K σ The hysteresis loss coefficient is increased to account for stress variations.

3. The method for calculating iron loss of an axial flux permanent magnet motor according to claim 2, characterized in that, The correction factor K that takes into account the effect of temperature c The formula for calculating (t) is: ； Where d is the thickness of the silicon steel sheet, ρ is the density of the silicon steel sheet, and k t K is the temperature change coefficient, Δt is the temperature change, and k is the temperature change coefficient. c0 ρ is the eddy current loss coefficient at the reference temperature. e0 This refers to the resistivity of the metal at room temperature. ρ e (t) represents the resistivity of the metal at different temperatures, and its calculation formula is: 。 4. The method for calculating iron loss of an axial flux permanent magnet motor according to claim 3, characterized in that, The eddy current correction coefficient K c The formula for calculating (f) is: ； Among them, K c denoted as eddy current loss coefficient, and f is the fundamental frequency of the alternating magnetic field. D is the loss constant considering the skin effect, and its calculation formula is: ； Where: μ is the average magnetic permeability. The conductivity of silicon steel sheet.

5. The method for calculating iron loss of an axial flux permanent magnet motor according to claim 4, characterized in that, The formula for calculating the hysteresis loss correction coefficient K(B) is as follows: ； Among them, t m Take a constant of 0.65, Let λ be the magnetic flux density amplitude generated by the hysteresis loop effect, S be the total number of hysteresis loops in one period, and λ be the hysteresis loop number. B is the correction constant for hysteresis loss. m The peak value of the magnetic flux density for each order is denoted as .

6. The method for calculating iron loss of an axial flux permanent magnet motor according to claim 5, characterized in that, This also includes first fitting a number of parameters to the iron loss coefficient, then using a sparrow search algorithm to optimize the fitted iron loss coefficient to obtain the iron loss coefficient at different frequencies; and using this method to optimize the hysteresis loss coefficient K under rated operating conditions. h0 Eddy current loss coefficient K c and additional loss coefficient K a .

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