A current equalization control method for a hybrid-structure dual-winding motor

Abstract

The application discloses a kind of mixed structure's double-winding steering motor current equalization control method and system, this method comprehensively the advantage of vector space decoupling coordinate transformation and double dq coordinate transformation, can make the current of two windings keep balance, simultaneously eliminate the coupling phenomenon between winding, to improve the control performance of line control steering system;Receive reference current signal and convert actual current signal into reference voltage signal by transformation matrix, decoupling converter further processes reference voltage signal, obtains actual voltage signal;By Park inverse transformation and SVPWM, actual voltage signal is converted into driving current, and drives double-winding motor to rotate.The application eliminates the current imbalance phenomenon of double-winding steering motor, improves steering angle tracking control performance, and has better generality and control performance.The application is expected to be applied in line control steering system and other fields, to improve the working efficiency and stability of system.

Description

Technical Field

[0001] This invention belongs to the field of automotive steering system technology, specifically relating to a current balancing control method for a hybrid dual-winding motor. Background Technology

[0002] Current automotive steer-by-wire systems typically use a single-winding three-phase permanent magnet synchronous motor as the actuator. A failure in one winding will cause the entire steering motor to fail, severely reducing the reliability of the steer-by-wire system. A dual-winding motor, on the other hand, consists of two sets of three-phase windings that are electrically independent. A failure in one winding does not affect the normal operation of the other, providing excellent fault tolerance and reliability.

[0003] However, the current imbalance caused by the difference in resistance and inductance between the two windings of a dual-winding motor, or the power supply voltage imbalance caused by the difference between the two inverters, often leads to an uneven current between the two windings. This, in turn, causes uneven heating of the dual-winding motor, which may damage the insulation structure of the windings. The winding with a larger current may also be at risk of overload and burnout. Chinese invention patent application number CN201711054340.2, entitled "A Current Balancing Control Method for Multiphase Motors Based on Generalized Symmetrical Component Theory," discloses a method of decomposing a multiphase motor into n symmetrical m-phase systems and using a PI control algorithm for current balancing control. However, it does not consider the influence of parameter uncertainties and external disturbances on the current balancing control of a dual-winding motor. Chinese invention patent application number CN202111353115.5, entitled "A Current Balancing Control Method for Dual-Winding Motors Based on a Dual dq Coordinate Transformation Structure," designs a sliding mode controller based on an RBF neural network and superimposes the output of the sliding mode controller with the output of the PI current controller to jointly act on the dual-winding motor, achieving current balancing between the two sets of windings in the dual-winding steering motor. However, this method is based on a dual dq coordinate transformation structure, which presents a coupling problem between the two sets of windings. Summary of the Invention

[0004] To address the shortcomings of the prior art, the present invention aims to provide a hybrid structure dual-winding motor current balancing control method to solve the problem of current balancing control of dual-winding motors that does not consider the coupling effect between the two sets of windings in the prior art. The control method of the present invention can combine the advantages of vector space decouping (VSD) coordinate transformation and dual dq coordinate transformation to keep the current of the two sets of windings balanced, while eliminating the coupling phenomenon between the two sets of windings and improving the control performance of the steer-by-wire system.

[0005] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0006] The present invention provides a current balancing control method for a hybrid dual-winding motor, comprising the following steps:

[0007] Step 1: Receive the reference current i for the D1 and Q1 axes D1,ref i Q1,ref The reference current i for the D2 axis and Q2 axis D2,ref i Q2,ref , where the reference current i D2,ref i Q2,ref All are zero;

[0008] Step 2: Acquire the actual phase current i of the dual-winding motor using a current sensor. A i B i C i U i V i W And through the first transformation matrix T DQ Converted into actual current i for D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 The actual rotor speed ω and actual rotor angle θ are collected by the motor position sensor. r ;

[0009] Step 3: Subtract the reference current from the actual current of axis D1, the reference current from the actual current of axis Q1, the reference current from the actual current of axis D2, and the reference current from the actual current of axis Q2. Input the resulting differences into the PI controllers respectively. The PI controllers output the reference voltage u of the undecoupled axes D1, Q1, D2, and Q2. D1* ,u Q1* ,u D2* ,u Q2* ;

[0010] Step 4: Set the reference voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1* ,u Q1* ,u D2* ,u Q2* The actual current i of the D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 The actual rotor speed ω is input to the decoupling converter, which outputs the actual voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1 ,u Q1 ,u D2 ,u Q2 ;

[0011] Step 5: Convert the actual voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1 ,u Q1 ,u D2 ,u Q2 Input the second transformation matrix T dq Output the original voltage u of the d1-q1 subspace and the d2-q2 subspace. d1 ,u q1 ,u d2 ,u q2 ;

[0012] Step 6: Actual current i of D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 After the second transformation matrix T dq The actual current i converted into the d1-q1 subspace and the d2-q2 subspace d1 i q1 i d2 i q2 ;

[0013] Step 7: Convert the actual current i d1 i q1 i d2 i q2 Input current equalization controller, output current equalization compensation u d1+ ,u q1+ ,u d2+ ,u q2+ ;

[0014] Step 8: Equalize and compensate the current u d1+ ,u q1+ ,u d2+ ,u q2+ With the original voltage u d1 ,u q1 ,u d2 ,u q2 The summation result is used to obtain the actual voltage u in the α1-β1 and α2-β2 subspaces through the inverse Park transform. α1 ,u β1 ,u α2 ,u β2 ;

[0015] Step 9: Convert the actual voltage u α1 ,u β1 ,u α2 ,u β2 The input is fed into SVPWM to obtain the voltage signal S that can drive a three-phase inverter. 1-6 and S 7-12The three-phase inverter outputs drive current to the dual-winding motor, driving the dual-winding motor to rotate.

[0016] Furthermore, the first transformation matrix T in step 2 DQ The expression is as follows:

[0017]

[0018]

[0019] The first transformation matrix T DQ Applying this to phase current, we can obtain:

[0020]

[0021] In the formula, i D1 i Q1 These are the actual currents of the D1 axis and the Q1 axis, respectively; i D2 i Q2 These are the actual currents of the D2 axis and the Q2 axis, respectively.

[0022] Furthermore, the expression for the PI controller in step 3 is as follows:

[0023] u(t) = K p e(t)+K i ∫e(t)dt

[0024] In the formula, u(t) is the output signal of the PI controller, and K p K is the proportionality coefficient. i Here, is the integral coefficient, and e(t) is the input error signal.

[0025] Furthermore, the expression for the decoupling converter in step 4 is as follows:

[0026]

[0027]

[0028] In the formula, L D1 ,L Q1 ,L D2 L Q2 For the synchronous inductance of a two-winding motor, ψ PM ω is the flux linkage of the permanent magnet, ω is the rotor speed, and p is the number of pole pairs.

[0029] Furthermore, the expression for the second transformation matrix in step 5 is:

[0030]

[0031] Furthermore, the current balancing controller in step 7 performs the following steps:

[0032] The actual current i q1 with i q2 The difference is input to the PI controller, and the PI controller outputs a compensation signal u. q1+ The compensation signal u is added to the output signal of the q-axis PI controller of the first winding. q2+ The output signal of the second winding q-axis PI controller is added to balance the currents of the q1 and q2 axes; the current i d1 and i d2 The difference is input to the PI controller, and the PI controller outputs a compensation signal u. d1+ The compensation signal u is added to the output signal of the PI controller on the d-axis of the first winding. d2+ The output signal of the second winding d-axis PI controller is added to achieve a balance between the d1-axis and d2-axis currents;

[0033] Compensation signal u d1+ ,u q1+ ,u d2+ ,u q2+ The following relationship must be satisfied:

[0034]

[0035] Furthermore, the inverse Park transform in step 8 is expressed as:

[0036]

[0037] In the formula, θ r This represents the actual angle of the rotor.

[0038] The beneficial effects of this invention are:

[0039] This invention effectively eliminates the current imbalance between the two windings of a dual-winding motor, ensuring equal current in both windings, reducing the degree of current imbalance, improving the overall performance of the motor, and extending its service life. Furthermore, the hybrid coordinate transformation structure proposed in this invention eliminates the coupling phenomenon between the two windings of the dual-winding motor. When designing control algorithms for the dual-winding motor, it can be treated as a typical three-phase motor, making it more general and offering better control performance compared to the dual dq coordinate transformation structure. Attached Figure Description

[0040] Figure 1 This is a control block diagram of the method of the present invention.

[0041] Figure 2 This is a block diagram of the current balancing controller. Detailed Implementation

[0042] To facilitate understanding by those skilled in the art, the present invention will be further described below with reference to embodiments and accompanying drawings. The content mentioned in the embodiments is not intended to limit the present invention.

[0043] Reference Figure 1 , Figure 2 As shown, the current balancing control method for a hybrid dual-winding motor of the present invention comprises the following steps:

[0044] Step 1: Receive the reference current i for the D1 and Q1 axes D1,ref i Q1,ref The reference current i for the D2 axis and Q2 axis D2,ref i Q2,ref , where the reference current i D2,ref i Q2,ref All are zero.

[0045] Step 2: Acquire the actual phase current i of the dual-winding motor using a current sensor. A i B i C i U i V i W And through the first transformation matrix T DQ Converted into actual current i for D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 The actual rotor speed ω and actual rotor angle θ are collected by the motor position sensor. r ;

[0046] First transformation matrix T DQ The expression is as follows:

[0047]

[0048]

[0049] The first transformation matrix T DQ Applying this to phase current, we can obtain:

[0050]

[0051] In the formula, i D1 i Q1 These are the actual currents of the D1 axis and the Q1 axis, respectively; i D2 i Q2 These are the actual currents of the D2 axis and the Q2 axis, respectively.

[0052] Step 3: Subtract the reference current from the actual current of axis D1, the reference current from the actual current of axis Q1, the reference current from the actual current of axis D2, and the reference current from the actual current of axis Q2. Input the difference results into the PI controller respectively. The PI controller outputs the reference voltage u of the undecoupled axes D1, Q1, D2, and Q2. D1* ,u Q1* ,u D2* ,u Q2* ;

[0053] The expression for the PI controller is as follows:

[0054] u(t) = K p e(t)+K i ∫e(t)dt

[0055] In the formula, u(t) is the output signal of the PI controller, and K p K is the proportionality coefficient. i Here, is the integral coefficient, and e(t) is the input error signal.

[0056] Step 4: Set the reference voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1* ,u Q1* ,u D2* ,u Q2* The actual current i of the D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 The actual rotor speed ω is input to the decoupling converter, which outputs the actual voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1 ,u Q1 ,u D2 ,u Q2 ;

[0057] The expression for the decoupling converter is as follows:

[0058]

[0059]

[0060] In the formula, L D1 ,L Q1 ,L D2 L Q2 For the synchronous inductance of a two-winding motor, ψ PM ω is the flux linkage of the permanent magnet, ω is the rotor speed, and p is the number of pole pairs.

[0061] Step 5: Convert the actual voltage u of the D1 axis, Q1 axis, D2 axis, and Q2 axis. D1 ,uQ1 ,u D2 ,u Q2 Input the second transformation matrix T dq Output the original voltage u of the d1-q1 subspace and the d2-q2 subspace. d1 ,u q1 ,u d2 ,u q2 ;

[0062] The expression for the second transformation matrix is:

[0063]

[0064] Step 6: Actual current i of D1 axis, Q1 axis, D2 axis, and Q2 axis D1 i Q1 i D2 i Q2 After the second transformation matrix T dq The actual current i converted into the d1-q1 subspace and the d2-q2 subspace d1 i q1 i d2 i q2 .

[0065] Step 7: Convert the actual current i d1 i q1 i d2 i q2 Input current equalization controller, output current equalization compensation u d1+ ,u q1+ ,u d2+ ,u q2+ ;

[0066] The current equalization controller performs the following steps:

[0067] The actual current i q1 with i q2 The difference is input to the PI controller, and the PI controller outputs a compensation signal u. q1+ The compensation signal u is added to the output signal of the q-axis PI controller of the first winding. q2+ The output signal of the second winding q-axis PI controller is added to balance the currents of the q1 and q2 axes; the current i d1 and i d2 The difference is input to the PI controller, and the PI controller outputs a compensation signal u. d1+ The compensation signal u is added to the output signal of the PI controller on the d-axis of the first winding. d2+ The output signal of the second winding d-axis PI controller is added to achieve a balance between the d1-axis and d2-axis currents;

[0068] Compensation signal u d1+ ,u q1+ ,u d2+ ,u q2+ The following relationship must be satisfied:

[0069]

[0070] Step 8: Equalize and compensate the current u d1+ ,u q1+ ,u d2+ ,u q2+ With the original voltage u d1 ,u q1 ,u d2 ,u q2 The summation result is used to obtain the actual voltage u in the α1-β1 and α2-β2 subspaces through the inverse Park transform. α1 ,u β1 ,u α2 ,u β2 ;

[0071] The inverse Park transform is represented as:

[0072]

[0073] In the formula, θ r This represents the actual angle of the rotor.

[0074] Step 9: Convert the actual voltage u α1 ,u β1 ,u α2 ,u β2 The input is fed into SVPWM (Space Vector Pulse Width Modulator) to obtain the voltage signal S that can drive the three-phase inverter. 1-6 and S 7-12 The three-phase inverter outputs drive current to the dual-winding motor, driving the dual-winding motor to rotate.

[0075] This invention has many specific applications. The above description is only a preferred embodiment of this invention. It should be noted that for those skilled in the art, several improvements can be made without departing from the principle of this invention, and these improvements should also be considered within the scope of protection of this invention.

Claims

1. A hybrid structured dual-winding motor current equalization control method, characterized by, The steps are as follows: Step 1: Receive reference currents for D1 and Q1 axes reference currents for D2 and Q2 axes where the reference currents are all zero; Step 2: Collect actual phase currents of the double-winding motor and convert into D1, Q1, D2, Q2 actual currents by the first transformation matrix ; Collect actual rotor speed and actual rotor angle ;​ Step 3: difference between the reference current and the actual current of the D1 axis, difference between the reference current and the actual current of the Q1 axis, difference between the reference current and the actual current of the D2 axis, difference between the reference current and the actual current of the Q2, and the difference results are input into the PI controller respectively, and the PI controller outputs the reference voltage of the uncoupled D1 axis, Q1 axis, D2 axis, and Q2 axis ; Step 4: Set the reference voltages for the D1, Q1, D2, and Q2 axes. Actual currents of D1 axis, Q1 axis, D2 axis, and Q2 axis and actual rotor speed The input is fed into the decoupling converter, which outputs the actual voltages of the D1 axis, Q1 axis, D2 axis, and Q2 axis. ; Step 5: Convert the actual voltages of the D1 axis, Q1 axis, D2 axis, and Q2 axis. Input the second transformation matrix Output the original voltages of the d1-q1 subspace and the d2-q2 subspace. ; Step 6: Actual currents of D1 axis, Q1 axis, D2 axis, and Q2 axis After the second transformation matrix Transformed into actual currents in the d1-q1 and d2-q2 subspaces ; Step 7: Convert the actual current Input current equalization controller, output current equalization compensation ; Step 8: Perform current equalization compensation With the original voltage The result of the summation is obtained by inverse Park transform. subspace and Actual voltage of subspace ; Step 9: Convert the actual voltage The input is fed into SVPWM to obtain the voltage signal S that can drive a three-phase inverter. 1-6 and S 7-12 The three-phase inverter outputs drive current to the dual-winding motor, driving the dual-winding motor to rotate; The first transformation matrix in step 2 The expression is as follows: （1）； （2）； The first transformation matrix Applying this to phase current, we can obtain: （3）； In the formula, , These are the actual currents of the D1 axis and the Q1 axis, respectively. , These are the actual currents of the D2 axis and the Q2 axis, respectively. The expression for the decoupling converter in step 4 is as follows: （4）； （5）； In the formula, , , , For a dual-winding motor, the synchronous inductor It is a permanent magnet flux linkage. The rotor speed, It is the extreme logarithm; The expression for the second transformation matrix in step 5 is: （6）。 2. The current balancing control method for a hybrid dual-winding motor according to claim 1, characterized in that, The expression for the PI controller in step 3 is as follows: ； where u(t) is the output signal of the PI controller, K p is the proportional coefficient, K i is the integral coefficient, and e(t) is the input error signal.

3. The current balancing control method for a hybrid dual-winding motor according to claim 1, characterized in that, The current balancing controller in step 7 performs the following steps: The actual current i q1 with i q2 The difference is input to the PI controller, and the PI controller outputs a compensation signal. The compensation signal is added to the output signal of the q-axis PI controller of the first winding. The output signal of the second winding q-axis PI controller is added to balance the currents of the q1 and q2 axes; the current i d1 and i d2 The difference is input to the PI controller, and the PI controller outputs a compensation signal. The compensation signal is added to the output signal of the PI controller on the d-axis of the first winding. The output signal of the second winding d-axis PI controller is added to achieve a balance between the d1-axis and d2-axis currents; Compensation signal The following relationship must be satisfied: 。 4. The current balancing control method for a hybrid dual-winding motor according to claim 1, characterized in that, The inverse Park transform in step 8 is represented as follows: （7）； In the formula, This represents the actual angle of the rotor.

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