A high power factor control method of electrolytic capacitorless permanent magnet motor driving system

Abstract

The application discloses a high power factor control method of an electrolytic capacitor-free permanent magnet motor driving system, which comprises the following steps: constructing a three-input three-output model of inverter power about motor dq-axis current, utilizing a relative gain matrix to analyze the coupling degree of the model, and selecting three maximum coupling degree paired relationships as optimal control paired relationships; constructing inverter power instruction values according to the power relationship of the driving system; constructing an inverter power closed-loop control loop based on the optimal control paired relationships to complete the following of the inverter power instruction values; obtaining inverter power feedforward compensation values by subtracting feedback values from the inverter power instruction values, converting the inverter power feedforward compensation values into alpha-beta-axis voltage compensation values, and compensating the alpha-beta-axis voltage compensation values into alpha-beta-axis voltage instructions; and then performing space vector modulation to realize the control of the inverter. Therefore, high grid input power factors and low grid input current harmonics can be obtained, and the grid-side power quality of the driving system is improved.

Description

Technical Field

[0001] This invention belongs to the field of variable frequency drive for motors, and more specifically, relates to a high power factor control method for a permanent magnet motor drive system without electrolytic capacitors. Background Technology

[0002] In recent years, variable frequency motor drive technology has been widely applied in various fields. Traditional variable frequency drive technology adopts an AC-DC-AC topology, mainly composed of a single-phase uncontrolled rectifier bridge, a power factor correction (PFC) circuit, bus capacitors, and a three-phase inverter. However, the large-capacity aluminum electrolytic capacitors used in traditional variable frequency drive systems affect the lifespan and reliability of the entire drive system, and the introduction of additional PFC circuits is not conducive to the development of systems towards low cost, miniaturization, and high reliability. To solve these problems, small-capacity film capacitors can be used to replace aluminum electrolytic capacitors, eliminating the need for PFC circuits. This new drive system is called an "electrolytic capacitor-free drive system." Film capacitors have advantages such as low loss, good high-temperature stability, and long lifespan, and offer higher reliability compared to electrolytic capacitors, effectively preventing system failures caused by capacitor failure. Furthermore, eliminating the PFC circuit further reduces the size and cost of the drive system.

[0003] The film capacitors used in electrolytic capacitor-free drive systems typically have capacitance values ​​ranging from a few microfarads to tens of microfarads, resulting in limited energy storage capacity. The instantaneous power of the grid-side input and the inverter output is no longer decoupled. By rationally controlling the instantaneous power output of the inverter, the instantaneous power of the grid-side input can be indirectly adjusted, thereby correcting the grid-side power factor without using a PFC circuit. This is one of the unique advantages of electrolytic capacitor-free drive systems. Researching high power factor control methods for electrolytic capacitor-free permanent magnet motor drive systems contributes to the development of frequency converter drive products towards lower cost, higher power density, and higher reliability, possessing significant theoretical and practical value.

[0004] Currently, high power factor control methods for capacitorless permanent magnet motor drive systems mainly fall into two categories:

[0005] (1) Inverter power open-loop control method: This type of control method establishes open-loop control based on the mathematical relationship between inverter power and q-axis current. However, this type of control method is easily affected by parameter changes, current loop bandwidth and external interference, and it is difficult to obtain a sufficiently high grid-side power factor.

[0006] (2) Inverter power closed-loop control method: This type of control method establishes a closed-loop control loop between inverter power and q-axis current, ignoring the effect of d-axis current on inverter power control. Compared with open-loop control method, closed-loop control method can achieve better disturbance rejection performance and grid-side power factor. However, the d-axis current command needs to be manually adjusted or other complex control strategies need to be adopted, and there is still room for improvement. Summary of the Invention

[0007] To address the shortcomings and improvement needs of existing technologies, this invention provides a high power factor control method for a capacitor-free permanent magnet motor drive system, aiming to obtain a high grid input power factor and low grid input current harmonics, thereby improving the grid-side power quality of the drive system.

[0008] To achieve the above objectives, according to a first aspect of the present invention, a high power factor control method for a capacitor-free permanent magnet motor drive system is provided, comprising:

[0009] S1. Construct a three-input three-output model of inverter power with respect to the dq-axis current of the motor. Use the relative gain matrix to perform coupling analysis on the three-input three-output model and select the three pairing relationships with the largest coupling as the optimal control pairing relationships.

[0010] S2. Construct the inverter power command value based on the power relationship of the drive system;

[0011] S3. Based on the optimal control pairing relationship, construct the inverter power closed-loop control loop to complete the following of the inverter power command value;

[0012] S4. The inverter power feedforward compensation value is obtained by subtracting the inverter power command value from the feedback value, and the inverter power feedforward compensation value is converted into the αβ axis voltage compensation value and compensated into the αβ axis voltage command; then space vector modulation is performed to realize the control of the inverter.

[0013] Furthermore, in S1, a three-input, three-output model of the inverter power with respect to the dq-axis current of the motor is constructed, including:

[0014] The inverter power is decomposed into DC and 100Hz components, with the amplitudes of each component being: DC amplitude P0 and 100Hz sinusoidal amplitude P. a 100Hz cosine amplitude P b The motor's d-axis current is decomposed into DC and 100Hz components, with the amplitudes of each component as follows: d-axis current DC amplitude I d0 q-axis current DC amplitude I q0 q-axis current 100Hz sinusoidal amplitude I qa Thus, we obtain P0 and P a and Pε with I d0 I q0 and I qa A three-input, three-output model, where the intermediate variable P... ε For P0 and P b sum.

[0015] Furthermore, in S1, the optimal control pairing relationship is: I q0 Control P0, I qa Control P a I d0 Control P ε .

[0016] Further, S2 includes:

[0017] Calculate grid-side power command

[0018]

[0019] Among them, u g To drive the grid-side voltage of the system, θ is the desired fundamental amplitude of the grid-side current. g The phase angle of the phase-locked loop;

[0020] Calculate the power of an ideal capacitor

[0021]

[0022] Where, ω g C is the fundamental angular frequency of the grid-side voltage. dc U is the capacitance value of the bus. g This represents the fundamental amplitude of the grid-side voltage.

[0023] Based on the instantaneous power relationship of the drive system, the inverter power command value for:

[0024]

[0025] Further, S3 includes:

[0026] S31. Calculate the inverter power error Δp inv :

[0027]

[0028] Where, p inv Here, ΔP0 represents the inverter power feedback value, and ΔP represents the inverter power DC amplitude error. a The inverter power 100Hz sinusoidal amplitude error, ΔP bThe inverter power 100Hz cosine amplitude error;

[0029] S32. The inverter power error in S31 is filtered by a moving average filter to extract the DC part ΔP0.

[0030] S33, Multiply the inverter power error in S31 by 2sin(2θ) g Demodulation yields:

[0031] 2Δp inv sin(2θ g )=ΔP a +2ΔP0sin(2θ g )+ΔP b sin(4θ g )-ΔP a sin(4θ g )

[0032] S34. A moving average filter is used to filter the inverter power error after demodulation in S33, and the DC part ΔP is extracted. a ;

[0033] S35, Multiply the inverter power error in S31 by 2cos(2θ) g Demodulation yields:

[0034] 2Δp inv cos(2θ g )=ΔP b +2ΔP0cos(2θ g )+ΔP a sin(4θ g )-ΔP b sin(4θ g )

[0035] S36. A moving average filter is used to filter the inverter power error after demodulation in S35, and the DC part ΔP is extracted. b ;

[0036] S37. Based on the optimal control pairing relationship in S1, construct the inverter power component amplitudes P0 and P1 respectively. a and P ε The closed-loop control loop, the DC component amplitude of the q-axis current command and the amplitude of the 100Hz sine component respectively composed of P0 and P a The DC component amplitude of the d-axis current command output of the PI regulator in the control loop. By P ε The output of the PI controller in the control loop has the following expressions:

[0037]

[0038] Where, k p0 and k i0 These are the proportional and integral coefficients of the PI controller in the P0 control loop, respectively, k pa and k ia P respectively a The proportional and integral coefficients, k, of the PI controller in the control loop pε and k iε P respectively ε The proportional and integral coefficients of the PI controller in the control loop;

[0039] S38. Calculate the dq axis current commands according to the following expressions:

[0040]

[0041] Further, S4 includes:

[0042] S41. Calculate the inverter power feedforward compensation value p. cmp :

[0043]

[0044] S42. The inverter power feedforward compensation value in S41 is filtered using a first-order low-pass filter. The transfer function of the filter is:

[0045]

[0046] Where, ω c This is the filter cutoff frequency;

[0047] S43. Calculate the stator voltage compensation value Δu αβ Its expression is:

[0048]

[0049] Among them, i α i β These are the α-axis and β-axis currents, respectively.

[0050] S44. According to the direction of the αβ axis current, the stator voltage compensation value in S43 is decomposed into α-axis and β-axis components, and its expression is:

[0051]

[0052] S45. Adjust the stator voltage compensation values ​​Δu of the α and β axes. α , Δu β Superimposed on the α and β axis stator voltage commands and Space vector modulation is then performed to control the inverter.

[0053] Secondly, the present invention provides a high power factor control system for a capacitor-free permanent magnet motor drive system, comprising:

[0054] processor;

[0055] A memory storing a computer-executable program, which, when executed by the processor, causes the processor to perform the high power factor control method for the electrolytic capacitor-free permanent magnet motor drive system as described in the first aspect.

[0056] In summary, the above-described technical solutions conceived in this invention can achieve the following beneficial effects:

[0057] This invention constructs a three-input, three-output model of inverter power with respect to the motor's dq-axis current from the perspective of the magnitude of the main components. This model clarifies the control relationship between the dq-axis current and inverter power, guiding the design of the inverter power control loop. The inverter power closed-loop control loop can dynamically adjust the command value of the dq-axis current based on the inverter power feedback, making it highly practical. The inverter power feedforward compensation can effectively eliminate the influence of grid voltage distortion on inverter power control, giving the drive system better adaptability. Attached Figure Description

[0058] Figure 1 Topology diagram of a capacitorless drive system with single-phase AC input;

[0059] Figure 2 This is a block diagram of the high power factor control of the electrolytic capacitor-free permanent magnet motor drive system of the present invention;

[0060] Figure 3 This is a control model diagram of the inverter power and dq-axis current of the present invention;

[0061] Figure 4 This is a specific implementation diagram of the inverter power closed-loop control strategy of the present invention;

[0062] Figure 5 This is an experimental waveform diagram of the present invention;

[0063] Figure 6 The power analyzer of this invention records the grid-side current waveform and power analysis results. Detailed Implementation

[0064] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention. Furthermore, the technical features involved in the various embodiments of this invention described below can be combined with each other as long as they do not conflict with each other.

[0065] In this invention, the terms "first," "second," etc. (if present) in the invention and the accompanying drawings are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence.

[0066] The topology of a single-phase AC input electrolytic capacitor-free drive system is as follows: Figure 1 As shown, it mainly includes an uncontrolled rectifier bridge, thin-film capacitors, a three-phase inverter, and a permanent magnet synchronous motor. The rectifier bridge input is a single-phase 220V / 50Hz AC power supply, denoted as the grid-side voltage u. g The input current to the rectifier bridge is denoted as the grid-side current i. g Thin film capacitor C dc The two ends are the DC bus voltage, denoted as u. dc Based on the above-mentioned drive system, the inverter power control block diagram of the electrolytic capacitor-free permanent magnet synchronous motor drive system proposed in this invention is as follows: Figure 2 As shown. The specific implementation steps of the present invention include:

[0067] Step S1: Construct a three-input three-output model of inverter power with respect to the dq-axis current of the motor. Use the relative gain matrix to perform coupling analysis on the three-input three-output model and select the three pairing relationships with the largest coupling as the optimal control pairing relationships.

[0068] Before step S1, the following is also included:

[0069] S01, sampling motor three-phase stator current i a i b and i c Grid-side voltage u g DC bus voltage u dc Motor rotor electrical angle θ e The mechanical angular velocity ω of the electronic rotor m ;

[0070] S02, regarding the three-phase stator current i a i b and i c Perform Clark transformation to obtain the αβ axis current i α and i β ; for the αβ axis current i α and i βPerform the Park transformation to obtain the dq-axis current i. d and i q ;

[0071] S03, according to the speed command and speed feedback ω m For closed-loop speed control, the output of the speed PI regulator is the fundamental amplitude of the grid-side current.

[0072] Further, in step S1, a three-input three-output model of the inverter power with respect to the dq-axis current of the motor is constructed, and the coupling degree of the three-input three-output model is analyzed using the relative gain matrix. Specifically, this includes the following steps:

[0073] S11. Analyze the frequency composition of the inverter power, ignoring components with frequencies higher than 100Hz. The inverter power can be expressed as:

[0074] p inv =P0+P a sin(2θ g )+P b cos(2θ g )

[0075] =1.5(u d i d +u q i q )

[0076] Among them, u d and u q These are the stator voltages along the d and q axes, respectively.

[0077] S12, dq axis current i d and i q They can be represented as follows:

[0078]

[0079] Among them, I d0 I is the DC amplitude of the d-axis current. q0 I is the DC amplitude of the q-axis current. qa The amplitude of the q-axis current is a 100Hz sine wave.

[0080] The stator voltages of the S13 and dq axes can be expressed as follows:

[0081]

[0082] Among them, R s L is the stator resistance. d and L q For the dq axis inductance, ω eLet ψ be the electric angular velocity of the motor. f It is a permanent magnet flux linkage.

[0083] S14. Substituting the dq-axis stator voltage and current from S12 and S13 into the inverter power expression in S11, we can obtain:

[0084]

[0085]

[0086]

[0087] The coefficient of the last term in both S15 and S14 is 1.5ω. e ψ f Its relationship with electric angular velocity ω e The correlation is significantly greater than that of other items. Therefore, we can preliminarily conclude that: P a Mainly affected by I qa Control, P0 and P b All are mainly affected by I q0 To avoid two controlled variables being controlled by the same control variable, making it impossible to construct independent control loops, based on P0 and P... b The characteristics of the expression allow for the construction of a new state variable P. ε Its expression is:

[0088]

[0089] S16. Construct a closed-loop control circuit for the inverter power amplitude, such as Figure 3 As shown.

[0090] S17. For P0 and P respectively... a and P ε Linearization yields the following small-signal model:

[0091]

[0092] S18. Calculate the relative gain matrix R of the inverter power control system corresponding to the small-signal model in S17. RGA :

[0093]

[0094] S19. Calculate R under each working condition. RGA By selecting the relative gain closest to unit 1 in each row as the pairing scheme, the optimal control pairing scheme can be obtained as follows: using I q0 Control P0, I qa Control P a I d0 Control P εAt this point, the internal coupling of the control loop is at its maximum.

[0095] Step S2: Construct the inverter power command value based on the power relationship of the drive system.

[0096] Specifically, step S2, which involves constructing the inverter power command, includes the following steps:

[0097] S21. Let the physical meaning of the speed loop output be the desired fundamental amplitude of the grid-side current.

[0098] S22. Detect the grid-side voltage u of the drive system using a sensor. g ;

[0099] S23. Use a software phase-locked loop to perform phase locking and obtain its phase angle θ. g ;

[0100] S24. Calculate the grid-side power command.

[0101]

[0102] S25, Assuming bus voltage u dc The power of the ideal capacitor is calculated based on the absolute value of the grid-side voltage.

[0103]

[0104] Where, ω g C is the fundamental angular frequency of the grid-side voltage. dc U is the capacitance value of the bus. g This represents the fundamental amplitude of the grid-side voltage.

[0105] S26. Based on the instantaneous power relationship of the drive system, the inverter power command is:

[0106]

[0107] Step S3: Based on the optimal control pairing relationship, construct the inverter power closed-loop control loop to complete the following of the inverter power command value.

[0108] Specifically, step S3 constructs the inverter power closed-loop control circuit, including the following steps:

[0109] S31. Calculate the inverter power error Δp inv :

[0110]

[0111] Where, p inv Here, ΔP0 represents the inverter power feedback value, and ΔP represents the inverter power DC amplitude error.a The inverter power 100Hz sinusoidal amplitude error, ΔP b This represents the 100Hz cosine amplitude error of the inverter power.

[0112] S32, according to Figure 4 The inverter power error in S31 is filtered using a moving average filter (MAF) with a sliding window size of 0.01s to extract the DC component, i.e., ΔP0.

[0113] S33, Multiply the inverter power error in S31 by 2sin(2θ) g Demodulation yields:

[0114] 2Δp inv sin(2θ g )=ΔP a +2ΔP0sin(2θ g )+ΔP b sin(4θ g )-ΔP a sin(4θ g )

[0115] S34. A moving average filter is used to filter the error in S33. The moving average window size is 0.01s, and the DC component, ΔP, is extracted. a ;

[0116] S35, Multiply the inverter power error in S31 by 2cos(2θ) g Demodulation yields:

[0117] 2Δp inv cos(2θ g )=ΔP b +2ΔP0cos(2θ g )+ΔP a sin(4θ g )-ΔP b sin(4θ g )

[0118] S36. A moving average filter is used to filter the error in S35. The moving average window size is 0.01s, and the DC component, ΔP, is extracted. b ;

[0119] S37. Based on the optimal control pairing relationship in S1, construct the inverter power component amplitudes P0 and P1 respectively. a and P ε The closed-loop control loop, the DC component amplitude of the q-axis current command and the amplitude of the 100Hz sine component respectively composed of P0 and P a The DC component amplitude of the d-axis current command output of the PI regulator in the control loop. By P ε The output of the PI controller in the control loop has the following expressions:

[0120]

[0121] Where, k p0 and k i0 These are the proportional and integral coefficients of the PI controller in the P0 control loop, respectively, k pa and k ia P respectively a The proportional and integral coefficients, k, of the PI controller in the control loop pε and k iε P respectively ε The proportional and integral coefficients of the PI controller in the control loop.

[0122] S38. Calculate the dq axis current commands according to the following expressions:

[0123]

[0124] Step S4: Subtract the inverter power command value from the feedback value to obtain the inverter power feedforward compensation value, and convert the inverter power feedforward compensation value into the αβ axis voltage compensation value, and compensate it into the αβ axis voltage command; then perform space vector modulation to realize the control of the inverter.

[0125] Specifically, step S4 includes the following steps:

[0126] S41. Calculate the inverter power feedforward compensation value p. cmp :

[0127]

[0128] S42. The inverter power feedforward compensation value in S41 is filtered using a first-order low-pass filter with a cutoff frequency of 450Hz. The transfer function of the filter is:

[0129]

[0130] Where, ω c This is the filter cutoff frequency;

[0131] S43. Calculate the stator voltage compensation value Δu αβ Its expression is:

[0132]

[0133] Among them, i α i β These are the α-axis and β-axis currents, respectively.

[0134] S44. According to the direction of the αβ axis current, the stator voltage compensation value in S43 is decomposed into α-axis and β-axis components, and its expression is:

[0135]

[0136] S45. Adjust the stator voltage compensation values ​​Δu of the α and β axes. α , Δu β Superimposed on the α and β axis stator voltage commands and Space vector modulation is then performed. The resulting PWM signal is used to drive the inverter, ultimately achieving power control of the inverter.

[0137] Analysis of Result of Example:

[0138] Figure 5 This is an experimental waveform diagram of the present invention. Figure 5 It can be seen that the inverter power feedback p inv It follows its commands well; the d-axis current command is in DC form, and the q-axis current command is a combination of DC and sinusoidal signals. The dq-axis current control is good; the sinusoidal nature of the grid-side current is good.

[0139] Figure 6 This shows the grid-side current waveform and power analysis results recorded by the power analyzer. Figure 6 It can be seen that the grid-side current has good sinusoidal properties, the grid-side power factor is as high as 0.99119, and the grid-side current harmonics meet the IEC61000-3-2 standard. By applying the high power factor inverter control method of the present invention, the electrolytic capacitor-free permanent magnet synchronous motor drive system has achieved good grid-side power quality.

[0140] Those skilled in the art will readily understand that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A high power factor control method for a capacitor-free permanent magnet motor drive system, characterized in that, Includes the following steps: S1, Constructing inverter power relative to motor dq The three-input three-output model of shaft current is analyzed for coupling degree using the relative gain matrix. The three pairing relationships with the highest coupling degree are selected as the optimal control pairing relationships. S2. Construct the inverter power command value based on the power relationship of the drive system; S3. Based on the optimal control pairing relationship, construct the inverter power closed-loop control loop to complete the following of the inverter power command value; S4. Subtract the inverter power command value from the feedback value to obtain the inverter power feedforward compensation value, and convert the inverter power feedforward compensation value into... Shaft voltage compensation value, compensated to In the shaft voltage command; Then, space vector modulation is performed to control the inverter; In step S1, the inverter power is constructed relative to the motor. dq The three-input, three-output model of shaft current includes: The inverter power is decomposed into DC and 100Hz components, with the amplitudes of each component as follows: DC amplitude 100Hz sine wave amplitude 100Hz cosine amplitude ; the motor dq The shaft current is decomposed into DC and 100Hz components, with the amplitudes of each component as follows: d DC amplitude of shaft current , q DC amplitude of shaft current , q 100Hz sine wave amplitude of shaft current ; thereby obtaining , and and , and The three-input, three-output model, where the intermediate variable... for and sum; S3 includes: S31. Calculate inverter power error : in, This is the inverter power feedback value. This is the inverter power command value. Let be the phase angle of the phase-locked loop. For inverter power DC amplitude error, For the inverter power 100Hz sine wave amplitude error, The inverter power 100Hz cosine amplitude error; S32. The inverter power error in S31 is filtered using a moving average filter to extract the DC component. ; S33, Multiply the inverter power error in S31 by Demodulation yields: S34. A moving average filter is used to filter the inverter power error after demodulation in S33, and the DC part is extracted. ; S35, Multiply the inverter power error in S31 by Demodulation yields: S36. A moving average filter is used to filter the inverter power error after demodulation in S35, and the DC part is extracted. ; S37. Based on the optimal control pairing relationship in S1, construct the inverter power component amplitude respectively. , and The closed-loop control loop, q DC component amplitude of shaft current command and the amplitude of the 100Hz sine component Each by and The output of the PI regulator in the control loop, d DC component amplitude of shaft current command Depend on The output of the PI controller in the control loop has the following expressions: in, and They are respectively The proportional and integral coefficients of the PI controller in the control loop. and They are respectively The proportional and integral coefficients of the PI controller in the control loop. and They are respectively The proportional and integral coefficients of the PI controller in the control loop; S38. Calculate the following expressions respectively. dq Shaft current command: 。 2. The high power factor control method for a capacitor-free permanent magnet motor drive system according to claim 1, characterized in that, In S1, the optimal control pairing relationship is: control , control , control .

3. The high power factor control method for a capacitor-free permanent magnet motor drive system according to claim 2, characterized in that, S2 includes: Calculate grid-side power command : in, To drive the grid-side voltage of the system, The desired fundamental amplitude of the grid-side current. The phase angle of the phase-locked loop; Calculate the power of an ideal capacitor : in, The fundamental angular frequency of the grid-side voltage. This is the capacitance value of the busbar. This represents the fundamental amplitude of the grid-side voltage. Based on the instantaneous power relationship of the drive system, the inverter power command value for: 。 4. The high power factor control method for a capacitor-free permanent magnet motor drive system according to claim 1, characterized in that, S4 includes: S41. Calculate the inverter power feedforward compensation value. : S42. The inverter power feedforward compensation value in S41 is filtered using a first-order low-pass filter. The transfer function of the filter is: in, This is the filter cutoff frequency; S43. Calculate the stator voltage compensation value. Its expression is: in, , They are respectively , shaft current; S44, according to The direction of the shaft current decomposes the stator voltage compensation value in S43 into shaft and The axis component, its expression is: S45, will , Shaft stator voltage compensation value , Overlay , Shaft stator voltage command and And perform space vector modulation to control the inverter.

5. A high power factor control system for a capacitor-free permanent magnet motor drive system, characterized in that, include: processor; A memory storing a computer-executable program, which, when executed by the processor, causes the processor to perform a high power factor control method for a capacitorless permanent magnet motor drive system as described in any one of claims 1-4.

Images