Self-capacitance sensing system and method

Abstract

A system and method for self-capacitance sensing in a touch panel array integrated with a display panel are described. Each channel of the array has a self-capacitance (Ci) that varies in response to the presence or absence of a local touch event. The channel pairs are differentially read out by coupling them to branches of an AC-mode bridge. The AC-mode bridge includes a current source that drives each branch (and thus each channel) with a sinusoidal current, generating a branch voltage on each branch based on the branch's self-capacitance. The branch voltage is used to generate an output voltage. The sinusoidal current is controlled by comparing a drive signal with feedback from the branch, such that common-mode noise on the channel becomes a common-mode component of the sinusoidal current and is suppressed during the generation of the output voltage.

Description

Technical Field

[0001] This document relates to sensing circuits, and more specifically, to a self-capacitance sensing method utilizing the continuous time of an AC-mode bridge, for example, in a large capacitive touch panel. Background Technology

[0002] Many modern electronic applications include integrated touch panels, such as touchscreen displays. Typically, the touch sensing layer of a touchscreen display uses capacitive sensing to determine the time and location of a user's touch. Display noise can couple into the touch sensing layer, manifesting as noise in the readout of capacitive touch sensing information. Over time, this display noise coupling tends to increase, making it increasingly challenging to provide sufficiently low-noise readout circuitry for such applications.

[0003] Typically, the touch sensing layer of a display comprises an array of "mutual capacitors" and "self capacitors." For example, each row and column of the array has a self capacitor, and each row and column intersection has a mutual capacitor. Mutual capacitors in a touch panel tend to be the primary sensing elements because they tend to provide more accurate information about the location of a touch (e.g., a finger). Furthermore, self-capacitance sensing can provide a useful alternative (or supplementary) source of touch sensing information, especially in situations where mutual capacitance sensing is often inaccurate (e.g., when the user's finger is wet).

[0004] However, self-capacitance sensing can be more challenging because the signal level obtained using self-capacitance sensing is much smaller than that obtained using mutual capacitance sensing. The capacitance change induced in a self-capacitor during a touch event is often only a fraction of its underlying capacitance value. As display size increases, the sensitivity of self-capacitance sensing may decrease further. Reliably sensing such minute capacitance changes requires designing high-performance sensing circuitry. Summary of the Invention

[0005] The embodiments disclosed herein include systems and methods for self-capacitance sensing using an AC-mode bridge in a capacitive touch panel, such as a display integrated into a touchscreen electronic device. For example, a touch panel array is integrated with a display panel and has multiple touch sensing channels. Each channel has a corresponding channel self-capacitance (Ci), which includes: a corresponding base self-capacitance (Cs) corresponding to display noise coupled from the display panel capacitance to the channel; and a corresponding touch capacitance (Ctouch) that varies in response to the presence of a local touch event in the channel. The channel pairs are differentially read out by coupling them to branches of the AC-mode bridge. For example, channels i and j are coupled to two branches of the AC-mode bridge. The AC-mode bridge includes a current source that drives each branch (and thus each channel) using a sinusoidal current, generating a branch voltage in each branch based on the branch's self-capacitance. These two branch voltages are used to generate a differential output voltage. The sinusoidal current is controlled by comparing the drive signal with feedback from the branch, so that the common-mode noise on the channel becomes the common-mode component of the sinusoidal current and is suppressed during the generation of the output voltage.

[0006] According to a first set of embodiments, a system for performing self-capacitance sensing in a touch panel array integrated with a display panel is provided. The system includes: an AC-mode bridge having: an error amplifier for generating a loop control voltage in response to a sinusoidal drive signal and an error feedback signal; a first branch and a second branch, the first branch being coupled to an i-th channel of a touch panel array having an i-th channel self-capacitance (Ci_i), and the second branch being coupled to a j-th channel of the touch panel array having a j-th channel self-capacitance (Ci_j); a set of adjustable current sources configured to output a first sinusoidal current to the first branch and a second sinusoidal current to the second branch based on the loop control voltage, wherein the first branch generates a first branch voltage in response to applying the first sinusoidal current to Ci_i, and the second branch generates a second branch voltage in response to applying the second sinusoidal current to Ci_j, the error feedback signal being a function of the differential voltage between the first branch voltage and the second branch voltage; and an output amplifier for generating an i-th output voltage (Vout_i) based on the difference between the first branch voltage and the second branch voltage.

[0007] According to another set of embodiments, a method for performing self-capacitance sensing in a touch panel array integrated with a display panel is provided. The method includes: (a) coupling a first branch of an AC-mode bridge to an i-th channel of the touch panel array having an i-th channel self-capacitance (Ci_i), and coupling a second branch of the AC-mode bridge to a j-th channel of the touch panel array having a j-th channel self-capacitance (Ci_j); (b) generating a loop control voltage in response to a sinusoidal drive signal and an error feedback signal, the error feedback signal being a function of a first branch voltage on the first branch and a second branch voltage on the second branch; (c) generating a first sinusoidal current and a second sinusoidal current based on the loop control voltage; (d) driving the first branch with the first sinusoidal current to generate a first branch voltage based on Ci_i; (e) driving the second branch with the second sinusoidal current to generate a second branch voltage based on Ci_j; and (f) generating an i-th output voltage (Vout_i) based on the difference between the first branch voltage and the second branch voltage.

[0008] The following figures, description and claims provide a more detailed description of the above-mentioned contents, implementation and features of the disclosed technology. Attached Figure Description

[0009] The accompanying drawings, which are mentioned and form part of this document, illustrate embodiments of this disclosure. The drawings and description together serve to illustrate the principles of the invention.

[0010] Figure 1 A simplified portion of an illustrative touch panel array with rows and columns is shown.

[0011] Figure 2 A simplified portion of the touch sensing environment for further illustration of self-capacitance sensing is shown.

[0012] Figure 3 A high-level block diagram of a self-capacitance sensing environment for discrete-time sensing is shown.

[0013] Figure 4 Simplified illustrative waveforms and timings are shown at the three operational phases.

[0014] Figure 5 A high-level block diagram is shown to illustrate a self-capacitance sensing environment for discrete-time sensing using a conventional pre-charged capacitor (PCC) with a discharge block.

[0015] Figure 6A high-level block diagram is shown for a self-capacitance sensing environment that performs discrete-time sensing using conventional resistance-to-time conversion (RTC) with a discharge block.

[0016] Figure 7 A simplified circuit diagram of an illustrative self-capacitance-based touch sensing system using a voltage-mode bridge is shown.

[0017] Figure 8 A simplified circuit diagram of an illustrative self-capacitance-based touch sensing system using a novel AC-mode bridge, according to an embodiment described herein, is shown.

[0018] Figure 9 A more detailed circuit diagram of the illustrative error amplifier and a pair of adjustable current sources for an AC-mode bridge according to an embodiment described herein is shown.

[0019] Figure 10A and Figure 10B An illustrative touch panel array configured in an illustrative first readout frame and an illustrative second readout frame are shown respectively.

[0020] Figure 11 A simplified circuit diagram of an illustrative self-capacitance-based touch sensing system using a novel AC-mode bridge with a local current rotator, according to an embodiment described herein, is shown.

[0021] Figure 12 A simplified circuit diagram of another illustrative self-capacitance-based touch sensing system is shown, which shares an error amplifier among several instances of an AC-mode bridge according to an embodiment described herein.

[0022] Figure 13 A flowchart illustrating an illustrative method for performing self-capacitance sensing in a touch panel array according to embodiments described herein is shown.

[0023] In the accompanying drawings, similar parts and / or features may have the same reference numerals. Furthermore, various parts of the same type can be distinguished by using a second reference numeral after the reference numerals to differentiate similar parts. If only the first reference numeral is used in the description, the description applies to any similar parts having the same first reference numeral, regardless of the second reference numeral. Detailed Implementation

[0024] In the following description, numerous specific details are provided to provide a thorough understanding of the invention. However, those skilled in the art will understand that the invention can be practiced without one or more of these details. In other instances, features and techniques known in the art will not be described for the sake of brevity.

[0025] Many modern electronic applications include integrated touch panels, such as touchscreen displays. Typically, the touch sensing layer of a touchscreen display uses capacitive sensing to determine the time and location of a user's touch. Display noise can couple into the touch sensing layer, manifesting as noise in the readout of capacitive touch sensing information. Over time, this display noise coupling tends to increase, making it increasingly challenging to provide sufficiently high-performance readout circuitry for such applications.

[0026] Figure 1 A simplified portion of an illustrative touch panel array 100 with rows and columns is shown. Generally, the touch panel array 100 is configured to sense the time and location of a user touch the array by sensing capacitance changes and associating the sensed changes with specific row-column intersections. Mutual capacitors (Cm) 110 (shown as 110aa-110nn) are located at each row-column intersection. Additionally, each row and each column of the array 100 is coupled to a corresponding self-capacitor (Cs) 105 (shown as Csrx 105ra-105rn and Cstx 105ta-105tn). Row-oriented self-capacitors are labeled "Csrx," and column-oriented self-capacitors are labeled "Cstx."

[0027] As used herein, a touch event is considered any touch interaction with touch panel array 215, which can be detected by any one or more touch sensing circuits 120. A touch event is considered "local" for that particular row or column line when it is sufficiently close to a specific row or column line to manifest as a change in capacitance (mutual capacitance and / or self-capacitance), wherein such change can be detected at least by touch sensing circuit 120 coupled to that particular row or column line. Similarly, a touch event is considered "local" for that particular self-capacitance when it is sufficiently close to a specific row or column line coupled to a particular self-capacitance 105, wherein such change can be detected at least by touch sensing circuit 120 coupled to that particular row or column line; and a touch event is considered "local" for that particular self-capacitance 105 when it is sufficiently close to a particular mutual capacitor 110, wherein such change can be detected at least by touch sensing circuit 120 receiving a signal driven by mutual capacitor 110. Similarly, when a touch event is close enough to any part of the touch panel array 100 to manifest as a change in mutual capacitance and / or self capacitance, wherein such change can be detected at least by a specific touch sensing circuit 120, the touch event is considered herein to be local to that specific touch sensing circuit 120.

[0028] For example, a touch event occurring at the circled row-column intersection 115 (e.g., a placed finger) can cause a detectable capacitance change associated with the mutual capacitor 110bc, the row-direction self-capacitor 105rb, and the column-direction self-capacitor 105tc. Therefore, this touch event can be considered local to at least: the third column line, the second row line, the row-direction self-capacitor 105rb, the column-direction self-capacitor 105tc, the mutual capacitor 110bc, the touch sensing circuit 120rb, and the touch sensing circuit 120tc. In some cases, the same touch event may be local to multiple adjacent row lines, column lines, self-capacitors 105, and / or mutual capacitors 110 (i.e., and therefore detectable relative to multiple adjacent row lines, column lines, self-capacitors 105, and / or mutual capacitors 110).

[0029] Although not explicitly shown, the touch panel array 100 can be integrated as part of a display, such as a touchscreen display for an electronic device. The grid of row and column lines effectively provides multiple touch sensing channels. Mutual capacitors 110 in the touch panel array 100 tend to be primary sensing elements because they tend to provide more accurate information about the location of a touch (e.g., a finger). The mutual capacitance of a mutual capacitor 110 is typically measured by driving a signal through the column and row lines coupled to that mutual capacitor 110 and measuring the output. For example, measuring the capacitance of mutual capacitor 110bc can involve coupling a driver (not shown) to a column line corresponding to a self-capacitor 105tc in the column direction. The driver can send a signal through the column line, and this signal is coupled via mutual capacitor 110bc to a row line corresponding to a self-capacitor 105rb in the row direction. This signal can then be received at a touch sensing circuit 120rb coupled to that row line and measured to detect any change in capacitance indicating the presence of a touch event at mutual capacitor 110bc.

[0030] In addition to mutual capacitance 110 sensing, self-capacitance 105 sensing can also provide a useful alternative (or supplementary) source of touch sensing information, especially where mutual capacitance 110 sensing is often inaccurate (e.g., when the user's fingers are wet). Although self-capacitance 105... Figure 1 The components are shown as discrete devices, but each self-capacitor 105 represents the aggregated parasitic capacitance on the corresponding row or column line, shown as coupled to the self-capacitor 105. For example, as described above, the touch panel array 100 can be integrated with a display, and the parasitic capacitance of the self-capacitor 105 can be manifested from capacitively coupled display noise. Measuring the self-capacitor 105 typically does not involve driving signals traversing the row or column lines of the touch panel array 100. Instead, as... Figure 1As shown, each row line and each column line is coupled to a corresponding touch sensing circuit 120 (shown as touch sensing circuits 120ra-120rn for row lines and touch sensing circuits 120ta-120tn for column lines). As described herein, a touch event near a particular row or column line can be represented as a change in self-capacitance sensed by the touch sensing circuit 120 coupled to that row or column line.

[0031] Figure 2 A simplified portion of the touch sensing environment 200 for further illustration of self-capacitance sensing is shown. The environment shown illustrates a touch panel array 215 integrated with a display panel 220 (e.g., by...). Figure 1 (Implemented by touch panel array 100). Typically, the common cathode layer of display panel 220 (e.g., this common cathode layer can be considered as a local ground plane) faces touch panel array 215. The capacitively coupled display noise 225 from display panel 220 manifests at touch panel array 215 as self-capacitance (i.e., parasitic capacitance) on the row and column lines of touch panel array 215, which can be represented (e.g., as...). Figure 1 (This refers to the self-capacitor 105 coupled to those row and column lines. To avoid making the figures too complex, only one self-capacitor 105 is shown.)

[0032] For reference Figure 1 Each row and column line discussed can be coupled to a corresponding touch sensing circuit 120 (only one is shown). Embodiments of the touch sensing circuit 120 typically include an analog front-end (AFE) 230 that generates a channel sensing output voltage (Vout) 235. It is desirable to include an up-conversion stage in the AFE 230 to up-convert the input (DC) signal to a higher frequency, helping to eliminate low-frequency noise in the AFE. The AFE 230 path then includes a subsequent down-converting mixer (DCM) 245 to shift the signal back to DC. A filter 240 following the DCM 245 can help maintain the DC signal while eliminating higher-frequency glitches. For example, without up-conversion, the output of the AFE 230 is typically a DC amplitude (e.g., generated by the average amplitude of a series of unipolar pulses). The upconversion stage may result in the AFE 230 output being a series of pulses with alternating polarities, causing the detected signal to be represented as an amplitude at a specific frequency (i.e., not DC, and with effectively removed low-frequency noise). The DCM 245 can then mix the AFE 230 output with a sine wave of the same frequency to produce a DC output signal corresponding to the DC input signal.

[0033] When no touch event is present in a localized area of ​​the touch sensing circuit 120, the touch sensing circuit 120 is configured to generate Vout 235 based on the channel capacitance corresponding to the base capacitance value of the corresponding self-capacitor 105. When a touch event is present, the amount of self-capacitance is manifested through the change in self-capacitor 105. For example, as shown, a finger 210 touching the touch panel array 215 can act as a touch capacitor 205 providing a parallel capacitance path to ground. This can effectively increase the apparent self-capacitance of any self-capacitor 105 in the localized area of ​​the touch event. Therefore, the touch sensing circuit 120 is configured to generate Vout 235 based on the increased channel capacitance corresponding to the base capacitance value of the corresponding self-capacitor 105 plus the additional parallel capacitance provided by the touch event (i.e., channel capacitance = Cs 105 + Ctouch 205).

[0034] While this type of self-capacitor 105 sensing may be effective, it can be more challenging than mutual capacitor 110 sensing, at least because self-capacitor sensing often involves signal levels much smaller than those obtained by mutual capacitor 110 sensing. The capacitance change induced in the self-capacitor 105 during a touch event is typically only a fraction of its underlying capacitance value. For example, the difference in capacitance measured between touch and non-touch conditions may typically be less than 0.1%. To reliably sense such small capacitance changes, the sensing circuitry can be designed to effectively cancel out the underlying capacitance value with very low readout noise.

[0035] Figure 3 A high-level block diagram of a self-capacitance sensing environment 300 for discrete-time sensing is shown. Environment 300 typically includes an input stage 301, a discharge stage 302, and a sensing stage 303. Input stage 301 may represent a touch panel array seen at the input of coupled touch sensing circuitry 120 (e.g., ...). Figure 1 100 touch panel arrays Figure 2 Specific row lines or column lines (channels) of a touch panel array 215, etc. For example... Figure 2 As described herein, the input to the touch sensing circuit 120 may be an AFE 230, which may be implemented as a discharge stage 302 and a sensing stage 303 (i.e., as used herein, AFE 230 is considered to include both discharge stage 302 and sensing stage 303). Although specific components are shown in a particular arrangement and / or stage for clarity, alternative embodiments may rearrange and / or recombine components without departing from the scope of the embodiments described herein.

[0036] Input stage 301 represents a specific channel of the capacitive touch sensing array, as seen in AFE 230. As described above, the base capacitance of the channel's self-capacitor 105 corresponds to the display noise 225 from the capacitive coupling of the integrated display panel. The total self-capacitor (Ci) of the channel can be simply expressed as the self-capacitor 105 in parallel with the touch capacitor 205 (i.e., Ci = Cs + Ctouch). The amount of increase in touch capacitance 205 can be zero when there is no touch event locally on the self-capacitor 105, or the amount of increase in touch capacitance 205 can be a detectable value (e.g., Ctouch > 0) when there is a touch event locally on the self-capacitor 105. Input stage 301 is also shown to have impedance, such as Figure 3 The output of input stage 301 can be represented by a resistor. The output of input stage 301 can be represented as the input voltage level at input voltage node (Vin) 310 of AFE 230.

[0037] Figure 3 The operation of each stage typically follows three phases corresponding to the timing of the three switches K1 330, K2 335, and K3 340. The timing of these phases is controlled by the phase-controlled switch controller 360. To enable... Figure 3 The description is clearer. Figure 4 A simplified illustrative waveform and timing diagram at point 402 is shown for the three operating phases. Signals used to control specific switches are labeled using the switch's tag (e.g., ...). Figure 4 The signal marked "K1 330" is used for control. Figure 3 (Signal of switch K1 330).

[0038] In the first stage 402a, K1 330 closes for a charging period. For example... Figure 3 As shown, closing K1 330 couples the capacitor input 301 to the source voltage (Vcc) via a resistor (Rp), thereby charging the channel self-capacitance (Ci). Closing K1 330 also effectively couples Vin 310 to Vcc (e.g., via the input impedance, as shown), such that in the first stage 402a, the voltage drop between Vin 310 and ground increases to Vcc (or, based on the input impedance, increases to a predetermined level below Vcc) through the capacitor input 301 connected in series with Rp. After Ci is charged, K1 330 is opened. Figure 4 An illustrative graph of the Vin 310 charged to its full charge voltage level is shown. The thicker solid line represents the behavior of the Vin 310 in the absence of a touch event (Ci = Cs), while the thinner dashed line represents the behavior of the Vin 310 in the presence of a touch event (Ci = Cs + Ctouch).

[0039] In the second stage 402b, K2 335 closes for a predetermined discharge time (T) 405 (K1 330 and K3 340 are both open). The predetermined discharge time (T) 405 is also referred to herein as the "discrete discharge time," therefore, the self-capacitance sensing method described herein can be considered a type of discrete-time sensing method. Figure 3 As shown, closing K2 335 couples Vin 310 to the discharge block 325. The voltage across Ci and Rp (i.e., related to Vin 310) is discharged for a predetermined amount of time via the discharge current (Iout) 320 provided by the discharge block 325. The discharge rate through the discharge block 325 is inversely proportional to Ci. Therefore, the difference in Ci between the presence and absence of a local touch event results in a difference in the discharge rate through the discharge block 325. Figure 4 An illustrative graph of Vin 310 illustrates this effect. Because the capacitance Ci from the additional Ctouch 205 increases when a partial touch event is present, the discharge exhibited at Vin 310 is slower in the presence of a partial touch event (dashed line) than in the absence of a partial touch event (solid line). The components (at least discharge stage 302) are configured to draw a certain amount of charge from Vin 310 such that at the end of T 405 (i.e., after the discrete discharge time): in the absence of a partial touch event (i.e., when Ci = Cs), Vin has decreased to the discharge reference level (Vcm) 315; or in the presence of a partial touch event, Vin has decreased to a level detectably different from (e.g., higher than) Vcm 315. Therefore, after T 405, when K2 335 is turned off, the remaining charge on Ci (and the corresponding level of Vin 310) is detectably different between touch event and non-touch event conditions.

[0040] In stage 402c, K3 340 is closed (K1 330 and K2 335 are open). For example... Figure 3 As shown, closing K3 340 couples Vin 310 to amplifier block 350. Amplifier block 350 is configured to amplify the difference in Vin 310 between touch event and non-touch event conditions, such that the difference can be read as a reliable detectable difference in channel sense output voltage (Vout) 235 and output by AFE 230. Figure 4 An illustrative graph of Vout 235 is shown, and the illustrative difference of Vout 235 between touch event and non-touch event conditions is labeled Vsense 410. For a typical amplifier block 350 with gain (α), Vout 235 can be expressed as:

[0041] .

[0042] In some implementations, amplifier block 350 compares Vin 310 with a discharge reference level (Vcm) 315. For example, as described above, parameters (e.g., T 405, Iout 320, etc.) are set such that in the absence of a local touch event, Vin 235 decays to a level substantially equal to Vcm 315 in the second stage 402b, or in the presence of a local touch event, Vin 310 decays to a level detectably different from (e.g., greater than) Vcm 315. For a capacitor, the capacitor current (Ic) is known to be related to its capacitance and the voltage change over time: Ic = C*(dV / dt). In the context of this example implementation, this relationship can be reformulated as: Ic*T = (Vcc - Vcm)*Cs. Amplifier block 350 can amplify the difference between Vin310 and Vcm315 in the third stage 402c, so that the generated Vout235 is essentially zero when there is no touch event (where Vin≈Vcm), or the generated Vout235 exhibits a non-zero Vsense410 when there is a touch event (where Vin>Vcm).

[0043] As shown in the figure, the embodiment may include a phase-controlled switch controller 360, or may communicate with a phase-controlled switch controller 360. The phase-controlled switch controller 360 may output control signals to set the state of switches such as K1 330, K2 335, and K3 340. For example, the switch may be a transistor, and the control signals may be used to turn the transistor on or off. The phase-controlled switch controller 360 may include its own timing control (e.g., a clock, counter, etc.), or the phase-controlled switch controller 360 may communicate with additional components that control the timing of the signals output by the phase-controlled switch controller 360.

[0044] As described above, the signal level may be very low when the self-capacitor 105 senses a touch event. For example, the level difference of Vin 310 at the end of the second stage 402b between touch and non-touch conditions may be very small. Detection in the third stage 402c depends on the discrimination between touch and non-touch levels, which can depend on the reliably canceled underlying capacitance value Cs 105. For example, the presence of additional noise on Vin 310 or Vcm 315 may reduce the dynamic margin available for reliably distinguishing between touch and non-touch conditions.

[0045] Various types of discrete-time operations are widely used in touch sensing based on self-capacitance. Such methods generally begin by charging Ci (e.g., and the corresponding Vin 310) of each channel to a charging voltage level (e.g., Vcc). In the presence of a touch event, a portion of the charge represents the basic self-capacitance value (Cs 105), and a portion represents the additional touch capacitance (Ctouch 205). The technique attempts to cancel out the basic (Cs 105) portion, such that the residual charge after cancellation represents only the touch (Ctouch 205) portion. The technique can then convert the residual charge into an output signal that represents only the touch information of the channel. Using such a method, any basic portion remaining after cancellation tends to reduce the detection sensitivity. Therefore, various conventional methods have been explored to cancel out the basic value.

[0046] To supplement the context, Figure 5 and Figure 6 Two conventional methods for offsetting the underlying self-capacitance to facilitate touch sensing are shown. Figure 5 A high-level block diagram is shown of a self-capacitance sensing environment 500 for discrete-time sensing using a conventional pre-charged capacitor (PCC) with a discharge block. Environment 500 includes an input stage 501, a discharge stage 502, and a sensing stage 503. The input stage 501 can be connected to... Figure 3 The input stage 301 is essentially the same. For example, input stage 501 includes a self-capacitor 105 (which manifests as capacitively coupled display noise 225) connected in parallel with touch capacitor 205 to form a total channel self-capacitance (Ci). Touch capacitor 205 is represented as a variable capacitor; when there is no touch event in a localized area of ​​the channel, touch capacitor 205 is essentially zero, and when a touch event is present in a localized area of ​​the channel, touch capacitor 205 is a detectable non-zero quantity. The input stage also includes an impedance (Rp).

[0047] The discharge stage 502 is implemented as a conventional PCC discharge block 510 with a charging capacitor (Cc) 505. The sensing stage 503 is generally shown as including an operational amplifier 520. Figure 3 Similarly, the operation at each level is controlled by various switches, and the timing associated with controlling these switches can be referenced. Figure 4 The timing sequence is described similarly. For example, the switch is controlled such that: in the first stage (e.g., corresponding to...) Figure 4 In stage 402a), switch K1 330 (shown as 330a-330c) is set to charge Ci (and Vin 310) to the charging voltage level and precharge Cc 505; in the second stage (e.g., corresponding to Figure 4In stage 402b), switch K2 335 (shown as 335a-335c) is set to discharge Ci (and Vin 310) to the discharge voltage level using pre-charged Cc 505; in the third stage (e.g., corresponding to Figure 4 In stage 402b), switch K3 340 is set to convert the discharge level into an output voltage (Vout 235) representing touch information of the channel.

[0048] Specifically, in the first stage, switch K1 330a is closed to couple Vin 310 to Vcc, thereby coupling Ci to Vcc via Rp. Simultaneously, switches K1 330b and 330c are closed to couple Cc 505 between Vcc 310 and ground. Switch K2 335 is open, isolating Cc 505 from Vin 310. Therefore, when Ci is charging, the PCC discharge block 510 pre-charges Cc 505. In the second stage, switch K1 330 is open and switch K2 335 is closed. This decouples Cc 505 from its pre-charge path and couples Cc 505 to Vin 310. The capacitance of Cc 505 is substantially smaller than that of Cs 105, causing the coupling of Cc 505 to Vin 310 to draw charge from Ci. As described above, it is desirable to configure the discrete-time discharge cycle (e.g., T 405) and Cc 505 such that the amount of charge (Qd) pulled from Cs 105 substantially brings Vin 310 as close as possible to Vcm 315. Overall, Qd = Id * T, where Id is the discharge current. For proper operation, the capacitance of Cc 505 is typically chosen to be approximately one-third the capacitance of Cs 105 (e.g., if the capacitance of Cs 105 is 1 nF, then Cc 505 could be approximately 330 pF). In the third stage, switch K3 340 closes to couple the discharged Vin 310 to operational amplifier 620, thereby converting the discharge level (i.e., corresponding to the residual charge on Ci) into an output voltage (Vout 235) representing touch information of the channel.

[0049] In some applications, the use of the PCC discharge block 510 offers various characteristics, such as low sensitivity to clock jitter (especially in the second stage) due to the complete stability of operation in each phase. However, implementing the PCC discharge block 510 involves providing a Cc 505 for each channel (e.g., each instance of Cs 105 can have a corresponding instance of Cc 505). Especially in cases where there are dozens or more channels in a touch panel, the Cc 505 instances can consume a relatively large amount of silicon area, which may be undesirable for many applications.

[0050] To avoid the large space loss associated with the PCC discharge block 510 method, some conventional implementations use a resistive method to discharge Ci for each channel in a discrete time quantity. Figure 6 A high-level block diagram is shown of a self-capacitance sensing environment 600 for discrete-time sensing using conventional resistance-time conversion (RTC) with a discharge block. Environment 600 includes an input stage 601, a discharge stage 602, and a sensing stage 603. Input stage 601 can be connected to... Figure 3 Input level 301 and Figure 5 The input stage 501 is essentially the same. For example, input stage 601 includes a self-capacitor 105 (which manifests as capacitively coupled display noise 225) connected in parallel with touch capacitor 205 to form a total channel self-capacitance (Ci). Touch capacitor 205 is represented as a variable capacitor; when there is no touch event in a localized area of ​​the channel, touch capacitor 205 is essentially zero, and when a touch event is present in a localized area of ​​the channel, touch capacitor 205 is a detectable non-zero quantity. The input stage also includes an impedance (Rp).

[0051] Discharge stage 602 is implemented as a conventional RTC discharge block 610 with a discharge resistor (Rd) 605 (shown as a variable resistor). Sensing stage 603 is generally shown as including operational amplifier 620. Figure 3 and Figure 5 Similarly, the operation at each level is controlled by various switches, and the timing associated with controlling these switches can be referenced. Figure 4 The described timing is similar. For example, in the first stage (e.g., corresponding to...) Figure 4 In stage 402a), switch K1 330 is closed to charge Ci (and Vin 310) to the charging voltage level. In the second stage (e.g., corresponding to...), Figure 4 In stage 402b), switch K1 330 is open and switch K2 335 is closed to discharge Ci (and Vin 310) to the discharge voltage level via Rd 605. In the third stage (e.g., corresponding to...), Figure 4 In stage 402b), switch K3 340 is closed to couple Vin 310 to operational amplifier 620, thereby converting the discharge level (i.e., the residual charge on Ci) into an output voltage (Vout 235) representing the touch information of the channel.

[0052] As described above, the RTC discharge block 610 achieves multiple instances (such as the Cc 505 instance) that do not rely on large capacitors, and thus can save significantly more space. However, because current and voltage are inversely proportional in a resistor, the amount of charge discharged through Rd 605 varies with Vin 310 in the second stage. Therefore, the discharge provided by the RTC discharge block 610 can produce a very large (e.g., about 40%) signal loss. Furthermore, the RTC discharge block 610 is highly sensitive to clock jitter during discharge. For example, although the discharge period (e.g., T 405) is intended to be a predetermined, discrete time amount, clock noise can cause slight variations in the pulse width used to control the turn-on or turn-off timing of K2 335, which can effectively alter the discharge period. It is known that the capacitor current (Ic) is related to its capacitance and the voltage change over time: Ic = C*(dV / dt). If there is an increased pulse width due to clock jitter (Tj), then for the discharge current (Id), the voltage error at Vin 310 due to jitter (Vin_e) can be described as: Vin_e = Tj*Id / (Cs + Ctouch).

[0053] To avoid limitations imposed by PCC and RTC-based methods on self-capacitance-based sensing, voltage-mode bridges have been proposed. Figure 7 A simplified circuit diagram 700 of an illustrative self-capacitance-based touch sensing system using a voltage-mode bridge is shown. As shown, the voltage-mode bridge method relies on a differential input stage 701. The differential input stage 701 represents two touch sensing channels, the i-th channel and the j-th channel, of a touch panel array. Each channel has a corresponding total channel capacitance Ci 705, labeled Ci_i 705i and Ci_j 705j, respectively. As described herein, each Ci 705 may include a corresponding basic self-capacitance (Cs 105, labeled 105i and 105j for the i-th and j-th channels, respectively), representing the display noise 225 coupled from the integrated display panel capacitance to the channel. Each Ci 705 may also include a corresponding touch capacitance (Ctouch 205, labeled 205i and 205j for the i-th and j-th channels, respectively), which varies in response to the presence or absence of a local touch event in the associated channel.

[0054] It is generally assumed that although the display noise 225 can vary on the display panel, it tends to have very small local variance. Figure 7The contextual assumptions presented herein assume that the i-th and j-th channels are generally any adjacent channels. The term "adjacent" is used herein to generally refer to any channel or associated component (e.g., and any associated metal row or column lines, mutual capacitors, etc.) that is directly adjacent to each other on the touch panel, within a few rows of each other on the touch panel, or otherwise sufficiently close to each other such that display noise coupled to these channels can be considered common-mode noise for the cancellation purposes described herein. In some embodiments, the i-th and j-th channels are directly adjacent channels of the touch panel array (e.g., directly adjacent row lines or directly adjacent column lines). In other embodiments, the i-th and j-th channels are not directly adjacent, but are close enough to experience common-mode display noise sufficient for cancellation by the sensing circuitry described herein.

[0055] Since it is assumed that the i-th and j-th channels are adjacent, it is assumed that the display noise 225 is the common-mode noise of this pair of channels. Therefore, the Cs 105 of the two channels of the differential input stage 701 is presented as coupled to the same representation of the display noise 225. On the same basis, it can be assumed that Cs 105i and Cs 105j have substantially the same (e.g., or very close) fundamental self-capacitance, such that any difference between Ci_i 705i and Ci_j 705j is primarily attributable to the difference between Ctouch 205i and Ctouch 205j, which represent touch information.

[0056] A differential input stage 701 is coupled to a voltage-mode bridge comprising an alternating current (AC) voltage source 715, a driver amplifier 710, an output amplifier 720, and two variable resistors 730 (labeled resistors 730i and 730j for the i-th and j-th channels, respectively). The AC voltage source 715 outputs a sinusoidal drive signal, which is buffered (e.g., amplified) by the driver amplifier 710. Each variable resistor 730 is associated with a corresponding branch of the voltage-mode bridge. In one branch, variable resistor 730i is coupled on one side to the output of the driver amplifier 710 (i.e., the buffered version of the sinusoidal drive signal) and on the other side to Ci_i 705i and one of the two differential inputs (e.g., the positive input) of the output amplifier 720. In another branch, variable resistor 730j is coupled to the output of driver amplifier 710 on one side and to the other (e.g., the negative input) of Ci_j 705j and the other of the two differential inputs of output amplifier 720 on the other side. Therefore, the differential input of output amplifier 720 represents the differential input voltage Vin 725. To maximize the output signal in each branch, the resistance of each variable resistor 730 is adjusted to match the impedance of its corresponding Ci 705 (i.e., variable resistor 730i is impedance-matched to Ci_i 705i, and variable resistor 730j is impedance-matched to Ci_j 705j).

[0057] During operation, a sinusoidal drive signal from the drive amplifier 710 sinusoidally drives each branch of the voltage-mode bridge, thereby generating a corresponding channel response signal in each branch corresponding to Ci 705 associated with that branch. Each channel response signal includes a fundamental component due to Cs 105 of the branch and a touch component due to Ctouch 205 of the branch (e.g., the touch component may be absent if no local touch event occurs). As assumed above, if Cs 105i and Cs 105j are substantially the same, the corresponding fundamental components of the channel response signals are also substantially the same. At the input of the output amplifier, the substantially matched fundamental components manifest as the common-mode portion of Vin 725 and are often ignored (e.g., suppressed) by the output amplifier 720. Therefore, the output Vout 735 generated by the amplifier represents the difference between the channel response signals. This difference is primarily due to the touch component of the signal, thus representing local touch information.

[0058] While this method effectively cancels out some capacitively coupled display noise, it has certain limitations. One limitation is that impedance matching each variable resistor 730 with its associated Ci 705 essentially creates a voltage divider at each input of the output amplifier. Therefore, approximately half of the response signal for each channel is lost at the corresponding variable resistor 730, and the voltage swing at Ci 705 is only half that applied by the sinusoidal drive signal. Another limitation is that approximately half of the display noise is transmitted to the output amplifier (e.g., to the tap point that couples the differential input stage 701 to the voltage-mode bridge). Even though the display noise manifests as a common-mode signal on the bridge and is eventually suppressed by the output amplifier 720, it still consumes the input dynamic range of the output amplifier 720. This reduces the available dynamic range for detecting the differential signal, thereby decreasing the sensitivity of the output amplifier 720 to touch information.

[0059] The various methods described above, including traditional PCC and RTC methods and voltage-mode bridge methods, can be successful in some applications. However, they are often highly susceptible to display noise, which can be much larger than the signal variations in the channel caused by localized touch events. Therefore, implementing these and other methods in high-performance applications typically relies on timing synchronization of touch event sensing with display control signals to reduce the impact of capacitively coupled display noise (i.e., attempting sensing operations only when display noise is relatively low and avoiding sensing operations when display noise is relatively high). Designing implementations to handle this synchronization increases system complexity and reduces system flexibility. Furthermore, as capacitively coupled display noise continues to increase, it can be difficult to find time windows with sufficiently low display noise to support reliable touch sensing using such traditional methods.

[0060] The embodiments described herein include various novel techniques for self-capacitance-based sensing using a current-mode alternating current (AC) bridge to improve sensitivity and thereby enhance touch sensing performance. Such embodiments can operate without synchronization with display control signals. One resulting feature is that the embodiments described herein can support self-capacitance sensing in asynchronous operation modes. Another resulting feature is that the embodiments described herein can support stimulus frequency hopping. Yet another resulting feature is that the embodiments described herein can support scenarios with higher display noise, such as those where insufficient display noise can be compensated simply by synchronizing the sensing with the display signal.

[0061] Figure 8 A simplified circuit diagram 800 of an illustrative self-capacitance-based touch sensing system using a novel AC-mode bridge, according to embodiments described herein, is shown. Figure 7Similarly, the AC-mode bridge method relies on a differential input stage 801 representing two touch sensing channels (the i-th and j-th channels) of the touch panel array. Each channel has a corresponding total channel capacitance Ci 805, denoted as Ci_i 805i and Ci_j 805j, respectively. As described herein, each Ci 805 may include a corresponding basic self-capacitance (Cs 105, denoted as 105i and 105j for the i-th and j-th channels, respectively), representing the display noise 225 coupled from the integrated display panel capacitance to the channel. Each Ci 805 may also include a corresponding touch capacitance (Ctouch 205, denoted as 205i and 205j for the i-th and j-th channels, respectively), which varies in response to the presence or absence of a local touch event in the associated channel.

[0062] The AC-mode bridge includes a sinusoidal voltage source 815, an error amplifier 810, an output amplifier 820, and two adjustable current sources 830. The AC-mode bridge has two branches, each coupled to a corresponding channel of the differential input stage 801 (i.e., each branch of the AC-mode bridge is effectively coupled to a corresponding Ci 805). In the first branch, the first adjustable current source 830a is coupled between the local source voltage (Vdd) and the positive branch voltage (Vinp) node 825p, which is coupled to the i-th Ci 805i. In the second branch, the second adjustable current source 830b is coupled between the local source voltage (Vdd) and the negative branch voltage (Vinn) node 825n, which is coupled to the j-th Ci 805j. Vinp node 825p and Vinn node 825n are also coupled to the corresponding differential inputs of output amplifier 820, such that the voltage difference between Vinp node 825p and Vinn node 825n is the differential input voltage of output amplifier 820.

[0063] The embodiments are designed such that the two adjustable current sources 830 are nominally identical. Because it is recognized that in the actual implementation of circuits or other manufactured products, truly identical components are impractical or impossible, the phrase "nominally identical" (and its variations) is used for design purposes of equivalence, interchangeability, etc. For example, if two current sources are designed or intended to be copies of each other (i.e., as two instances of the same component), then these two current sources are considered "nominally identical" herein, regardless of whether process variations and other practical considerations actually tend to prevent the current sources from being truly identical. Some circuits described herein attempt at least in part to compensate for the practical implementation differences that arise between nominally identical components.

[0064] Error amplifier 810 is configured as a feedback loop to control two adjustable current sources 830 such that, following a sinusoidal drive signal output from AC voltage source 815, each adjustable current source 830 generates the same sinusoidal current 835 (labeled Iac_i 835i and Iac_j 835j for the i-th and j-th channels, respectively). As shown, the first (e.g., positive) input of error amplifier 810 is coupled to sinusoidal voltage source 815 to receive the sinusoidal drive signal. The second (e.g., negative) input of error amplifier 810 is coupled to two branches (e.g., via corresponding resistors) to convert the sinusoidal current 835 into a voltage at the second input of error amplifier 810. The output of error amplifier 810 is used for voltage control of the two adjustable current sources 830.

[0065] Using the error amplifier 810 in this feedback configuration, the sinusoidal current 835 is forced to be equal. Therefore, any differential voltage between Vinp node 825p and Vinn node 825n is due to the difference between Ci 805i and Cj 805j. Thus, the output Vout 730i of the output amplifier represents the difference in touch information between the i-th and j-th channels. For example, if Ctouch 205 is essentially zero in both branches (i.e., there is no touch condition locally in either the i-th or j-th channel) and Cs 105 is essentially equal in both branches (i.e., essentially all the display noise 225 is common to both branches), then there is essentially no differential voltage at the input of the output amplifier 820, and Vout 735i is essentially zero. If Ctouch 205 in any channel is non-zero (i.e., a touch condition exists locally in the i-th or j-th channel) and Cs 105 remains substantially equal in the two branches, then there is a differential voltage 820 at the input of the output amplifier 820, which will generate the corresponding Vout 735i signal.

[0066] As described above, the associated display noise 225 at Vinp node 825p and Vinn node 825n manifests as common-mode signal components. From the perspective of these common-mode signals, both Vinp node 825p and Vinn node 825n are low-impedance nodes due to the feedback loop of error amplifier 810. Therefore, the display noise 225 is significantly reduced at the input of output amplifier 820. Thus (e.g., with...) Figure 7Unlike a voltage-mode bridge, display noise 225 does not tend to reduce the dynamic range of output amplifier 820 and does not cause significant noise. Since the effect of display noise 225 on the sensitivity of output amplifier 820 is significantly reduced, the AC-mode bridge method can be implemented without relying on synchronization with the display control signal. For example, the AC-mode bridge can be implemented without relying on additional complex signals and circuitry dedicated to limiting the operation of the sensing circuit only during low display noise time windows. Furthermore, because the operation of the AC-mode bridge does not depend on this synchronization with the display signal, the frequency of the sinusoidal drive signal can be set independently of any particular display noise 225 profile.

[0067] Figure 9 A more detailed circuit diagram 900 shows an illustrative error amplifier 910 and a pair of adjustable current sources 920 for an AC-mode bridge according to an embodiment described herein. The error amplifier 910 is... Figure 8 An illustrative implementation of the error amplifier 810, the adjustable current source 920 is Figure 8 An illustrative implementation of the adjustable current source 830 is shown. As illustrated, the error amplifier 910 employs voltages Vinp 825p and Vinn 825n (corresponding to...). Figure 8 The voltages at Vinp node 825p and Vinn node 825n are used as inputs. The error amplifier 910 is implemented as a complementary metal-oxide-semiconductor (CMOS) operational amplifier, controlled by six bias signals: vbp1, vbp2, vbp3, vbn1, vbn2, and vbn3. The bias signals can be generated by bias generator 940, causing the error amplifier 910 to output a sine wave corresponding to the difference between Vinp 825p and Vinn 825n.

[0068] In the illustrated implementation, both adjustable current sources 920 are implemented identically. To conserve quiescent power, the adjustable current sources 920 are implemented as "Class AB" current sources. During each positive half-cycle of the sine wave, the PMOS portion of each adjustable current source 920 supplies current; during each negative half-cycle of the sine wave, the NMOS portion of each adjustable current source 920 sinks current. The Class AB current sources are designed to share the same control and have transistors of the same size, such that each current source produces the same output current.

[0069] Due to process variations and / or other practical considerations, a certain mismatch exists between the adjustable current sources 920 (e.g., differences in the performance between corresponding transistors in the two adjustable current sources 920) and so-called "flicker noise" (e.g., causing a mismatch between the PMOS and NMOS portions of each adjustable current source 920). To mitigate these types of noise, choppers 930 (labeled choppers 930a and 903b for adjustable current sources 920a and 920b, respectively) are added to the adjustable current source 920 to switch the current between the two outputs. As shown, the two outputs are two currents, which can be... Figure 8 The sinusoidal currents 835a and 835b are generated. Chopper 930 is controlled by chopper clock signals (swp1, swp2, swn1, and swn2), which can be generated by chopper clock generator 950. As shown, the chopper clock signals can generally be synchronous square waves, where swp1 and swn2 are in phase with each other, and swp2 and swn1 are in phase with each other but 180 degrees out of phase with swp1 and swn2. Chopper clock generator 950 is configured to generate chopper clock signals such that switching occurs at a chopping frequency higher than (e.g., significantly higher) than the sinusoidal signal frequency. This allows any mismatch and flicker noise to be shifted out of the frequency band of the signal of interest (i.e., the sinusoidal signal), resulting in good in-band matching between the sinusoidal currents 835 generated by adjustable current source 920.

[0070] When differentially reading all channels of a touch panel array, it is desirable to form differential pairs with any channels that are directly adjacent to each other; for most channels, there are directly adjacent channels on both sides. For example, in addition to differentially pairing channels 1 and 2, 3 and 4, 5 and 6, it is also desirable to pair channels 2 and 3, 4 and 5, etc. While the described implementation can eliminate mismatch and flicker noise between adjustable current sources 920 for its differential output, this assumes that the channels differentially read in the same AC mode bridge are coupled together with a pair of adjustable current sources 920. However, the same may not hold if the differential output is taken from two channels from different pairs of adjustable current sources 920. To differentially read all adjacent pairs, some embodiments execute each readout cycle as two readout frames. In each first readout frame, each channel is paired with one adjacent channel, and the differential reading is acquired between each pair of multiple channels. In each second readout frame, each channel is paired with its other adjacent channel, and the differential reading is acquired between each pair of multiple channels.

[0071] Figure 10A and Figure 10BAn illustrative touch panel array configured in illustrative first readout frame 1000a and illustrative second readout frame 1000b are shown, respectively. The touch panel array is shown to have several RX lines (labeled RX0-RX11) and several TX lines (labeled TX0-TX11). Switches are configured to selectively couple the lines to a sensing circuit in a differential manner. In each readout frame, adjacent pairs of lines are coupled to the sensing circuit to read out the pairs of lines differentially. For example, each readout of the array may include two or more readout frames. As shown in the figure, in the first readout frame 1000a (… Figure 10A In the second read frame 1000b (shown), the differential pairs are {RX0, RX1}, {RX2, RX3}, {RX4, RX5}, {RX6, RX7}, {RX8, RX9}, {RX10, RX11}, {TX0, TX1}, {TX2, TX3}, {TX4, TX5}, {TX6, TX7}, {TX8, TX9}, and {TX10, TX11}. Figure 10B In the figure shown, the switch is reconfigured to read each line with its adjacent line on the other side. As shown, the differential pairs in the second read frame are {RX1, RX2}, {RX3, RX4}, {RX5, RX6}, {RX7, RX8}, {RX9, RX10}, {TX1, TX2}, {TX3, TX4}, {TX5, TX6}, {TX7, TX8}, and {TX9, TX10}. In some implementations, RX or TX lines at the edge of the touch panel array may not be read in both read frames (e.g., RX0 line is differentially paired with RX1 line in the first read frame but is not part of any differential pair in the second read frame). In other implementations, RX or TX lines at the edge of the touch panel array may be differentially paired with a reference voltage (e.g., a common-mode voltage reference) to be read in one read frame.

[0072] In the implementation shown, the TX and RX lines can be read out simultaneously. This saves time and power during the readout operation. The same sinusoidal excitation can be applied regardless of which direction the TX / RX channel is sensed. A corresponding mutual capacitor (Cm 110, as shown) is coupled between each TX and RX line. Figure 1 As shown, this ensures that the two ends of Cm have the same voltage, and any interference caused by changes in Cm is eliminated. Therefore, power is actually wasted when driving in the non-sensing direction. Reading both the TX and RX lines simultaneously recovers this power.

[0073] Figure 11 A simplified circuit diagram 1100 of an illustrative self-capacitance-based touch sensing system using a novel AC-mode bridge with a local current rotator 1120, according to embodiments described herein, is shown. Unless otherwise stated, Figure 11The AC mode bridge can be used with Figure 8 The AC-mode bridge operates in the same or similar manner. The AC-mode bridge method is coupled to a differential input stage 801 representing two touch sensing channels (channel i and channel j) of the touch panel array. Each channel has a corresponding total channel capacitance Ci 805, denoted as Ci_i 805i and Ci_j 805j; the total channel capacitance Ci 805 may include a corresponding basic self-capacitance (Cs 105, denoted as 105i and 105j for channels i and j, respectively), corresponding to the display noise 225 coupled from the integrated display panel capacitance to the channel, and a corresponding touch capacitance (Ctouch 205, denoted as 205i and 205j for channels i and j, respectively), in response to the presence or absence of a local touch event in the associated channel.

[0074] As described above, some embodiments operate based on the assumption that the fundamental self-capacitances of channels i and j 105i and 105j are very close (e.g., assuming any difference can be easily canceled out by common-mode noise suppression at output amplifier 820, etc.). For example, in Figure 8 In the illustrated system, assuming the channels feeding the two branches have substantially the same fundamental self-capacitance (i.e., they capacitively couple out substantially the same display noise), reliable operation can be achieved by applying the same sinusoidal current 835 to these branches. However, the fundamental capacitances of the i-th and j-th channels may not be close enough for these embodiments to operate reliably in all cases. For example, especially when the edge region has camera holes, notches, or other features, even directly adjacent channels near the edge of the touch panel array may exhibit a large mismatch in their respective fundamental self-capacitance values. In some such cases, the ratio of the fundamental self-capacitances between directly adjacent touch sensing channels can be approximately 2:1. If the same sinusoidal current is applied to two branches with such a large fundamental self-capacitance mismatch, the AC-mode bridge method may not be able to effectively differentially cancel the fundamental self-capacitance values.

[0075] Figure 11The illustrated embodiment uses a local current rotator 1120 to set the AC current ratio to match the ratio of the fundamental self-capacitances between the branches of the AC-mode bridge. As shown, the AC-mode bridge includes an AC current source 815, an error amplifier 810, an output amplifier 820, a local current rotator 1120, and N adjustable current sources 1130 (i.e., labeled current sources 1130a-1130n). In some implementations, N can be any suitable integer greater than 2. In some implementations, N is at least ten. The local current rotator 1120 includes N rotator inputs and two rotator outputs. Each of the N rotator inputs is coupled to a corresponding adjustable current source 1130 among the N adjustable current sources 1130, such that each input receives the current output of the adjustable current source 1130 coupled thereto. For example, each adjustable current source 1130 outputs approximately the same unit current (Iunit), such that the total current from all N adjustable current sources 1130 is approximately N*Iunit.

[0076] At each of the two rotator outputs, the local current rotator 1120 outputs a corresponding sinusoidal current 1135 (labeled Iac_i 1135i and Iac_j 1135j). The first sinusoidal current Iac_i 1135i is generated from a first group of NM adjustable current sources 1130, and the second sinusoidal current Iac_j 1135j is generated from a second group of M adjustable current sources 1130. The adjustable current sources 1130 are rotated by the local current rotator 1120 such that, with each rotation, the first and second groups of adjustable current sources 1130 consist of different adjustable current sources 1130. For example, in the first rotation, the first group is {1130a, 1130b, 1130c}, and the second group is {1130d, 1130e} (i.e., N = 5, M = 2); in the second rotation, the first group is {1130b, 1130c, 1130d}, and the second group is {1130e, 1130a}.

[0077] M can be adjusted in any suitable manner. In one implementation, M is hard-coded in the local current rotator 1120. In another implementation, M is set (e.g., programmable) by one or more control signals, for example, by a control processor (not shown). M is set such that (NM): the ratio of M to the ratio of the i-th basic self-capacitance 105i to the j-th basic self-capacitance 105j. M can be any suitable integer between 1 and N-1. For example, if there are 20 adjustable current sources 1130 (i.e., N = 20), then Iac_i 1135i can be generated from 11 of the adjustable current sources 1130, and Iac_j 1135j can be generated from the remaining 9 adjustable current sources 1130; or Iac_i 1135i can be generated from 6 of the adjustable current sources 1130, and Iac_j 1135j can be generated from the remaining 14 adjustable current sources 1130. The number (N) of adjustable current sources 1130 can effectively determine the solution for the local current rotator 1120. For example, a larger N will increase the complexity of the circuit and consume more area, power, etc.; but a larger N is also beneficial for more finely adjusting the ratio between Iac_i 1135i and Iac_j 1135j.

[0078] Rotation of the adjustable current sources 1130 provides several advantages. First, semiconductor devices generate so-called "flicker" noise (or "1 / f" noise), and rotation among multiple adjustable current sources 1130 effectively mitigates the flicker noise from each device. Second, while all N adjustable current sources 1130 can be configured to nominally produce the same unit current, differences naturally exist between current sources (e.g., due to process variations between transistors), and rotation among multiple adjustable current sources 1130 effectively eliminates these variations at both ends of the two branches of the AC-mode bridge. Further use of the local current rotator 1120 allows setting the ratio of sinusoidal currents to compensate for the mismatch in the underlying self-capacitance between the i-th and j-th channels, making the same circuit easily reconfigurable to operate with different channel pairs having different mismatches. Notably, the feedback to the error amplifier 810 comes from both branches of the AC-mode bridge, making it based on the total current N*Iunit, independent of the selected ratio.

[0079] The above describes various embodiments with different features and limitations. For example, such as Figure 8 The two-current-source method of the illustrated embodiment is often less complex, has lower area consumption, and lower power consumption, but often relies on the fact that there is essentially no mismatch in the underlying self-capacitance between the differential sensing channel pairs. In contrast, such as Figure 11The rotating current source method of the illustrated embodiments is often complex, consumes more area, and consumes more power, but it often operates with a large mismatch in the basic self-capacitance between the differential sensing channel pairs. Some embodiments can use a combination of rotating current source methods to implement sensing circuitry for different channels of a touch panel array. For example, it is known that for the 15% of touch sensing channels closest to the edge of a given touch panel array, the basic self-capacitance values ​​between adjacent touch sensing channels often differ by at least 10% (e.g., this ratio is often greater than 1.1:1); while for the remaining 85% of touch sensing channels, the basic self-capacitance values ​​between adjacent touch sensing channels often differ by less than 10%. For such touch panel arrays, the rotating current source method can be used to implement the sensing circuitry for the 15% of touch sensing channels closest to the edge, and the two current source method can be used to implement the sensing circuitry for all other touch sensing channels.

[0080] Figure 12 A simplified circuit diagram 1200 of another illustrative self-capacitance-based touch sensing system is shown, illustrating a shared error amplifier 810 among several instances of an AC-mode bridge according to an embodiment described herein. As shown, a set of K+1 instances of the AC-mode bridge are each used to differentially sense a corresponding pair of touch sensing channels, as described herein. This set of K+1 AC-mode bridges includes a master AC-mode bridge 1210 and K slave AC-mode bridges 1220 (i.e., shown as slave AC-mode bridges 1120-1 to 1120-K). The master AC-mode bridge 1210 can be implemented according to any embodiment described herein. In the illustrated embodiment, the master AC-mode bridge 1210 is based on... Figure 8 This is implemented using an embodiment (i.e., a method based on two current sources). Alternatively, the main AC mode bridge 1210 can be implemented using a method based on a rotating current source, such as... Figure 11 As described in [the document], to achieve [the desired outcome].

[0081] As shown in the figure, the main AC mode bridge 1210 has a pair of branches, each branch being coupled to a corresponding touch sensing channel in the first pair of touch sensing channels (channels a and b) that generate the first pair of self-capacitance values ​​(Ci_a805a and Ci_b805b). For simplicity, assume that the main AC mode bridge 1210 is coupled to a reference... Figure 8The same manner of operation is described. Each of the first pair of adjustable current sources 830a1 and 830b1 generates a corresponding (nominally identical) sinusoidal current 835 (shown as 835a and 835b) for a corresponding branch. The error amplifier 810 outputs a loop control voltage 1215, which controls the adjustable current sources 830a1 and 830b1 based on the AC current source 815 and feedback from the branches. In each branch, the corresponding sinusoidal current 835 is applied to the corresponding self-capacitance, generating a corresponding branch input voltage (825p1 and 825n1) of the first pair of branch input voltages at the input of the first output amplifier 820-1. The first output amplifier 820-1 thus outputs a first Vout 735-1 corresponding to the difference between Ci_a 805a and Ci_b 805b.

[0082] As shown in the figure, each slave AC-mode bridge 1220 is configured to share the error feedback loop of the master AC-mode bridge 1210. Specifically, each slave AC-mode bridge 1220 receives the loop control voltage 1215 output by the error amplifier 810 of the master AC-mode bridge 1210, without including its own instance of error amplifier 810. Therefore, in each slave AC-mode bridge 1220, the sinusoidal current 835 in the branch is controlled based on the error feedback in the master AC-mode bridge 1210, rather than based on the feedback in the slave AC-mode bridge 1220.

[0083] For example, the first slave AC-mode bridge 1220-1 (i.e., the second AC-mode bridge in this group) has a pair of branches, each branch being coupled to a corresponding touch-sensing channel in a second pair of touch-sensing channels (e.g., channel c and channel d) that generate the second pair of self-capacitance values ​​(805c and 805d). Each of the second pair of adjustable current sources 830a2 and 830b2 generates a corresponding sinusoidal current 835c and 835d for a corresponding branch of the first slave AC-mode bridge 1220-1 based on a shared loop control voltage 1215. Therefore, the sinusoidal currents 835c and 835d are nominally identical to each other, and also nominally identical to the sinusoidal currents 835a and 835b in the branches of the master AC-mode bridge 1210. In each branch, a corresponding sinusoidal current 835 is applied to the corresponding self-capacitance, generating a corresponding branch voltage in the second pair of branch voltages (825p2 and 825n2) at the input of the second output amplifier 820-2, causing the second output amplifier 820-1 to output a second Vout 735-2 corresponding to the difference between 805c and 805d.

[0084] The implementation can utilize any number of AC-mode bridges greater than 1 (i.e., K ≥ 2). In one implementation, adjacent AC-mode bridge pairs share the error amplifier 810 (i.e., K = 2). See, for example, [link to relevant documentation]. Figure 10A and Figure 10B Approximately T / 2 sensing circuits (i.e., T / 2 AC mode bridges) can be used to sense T touch sensing channels (i.e., T ≥ 4). Figure 10A and Figure 10B In one implementation (T = 24), approximately T / 4 shared error amplifiers 810 are included. In another implementation, multiple sets of AC-mode bridges (i.e., K = 5) share error amplifiers 810 as a group of five AC-mode bridges. For example, T / 2 sensing circuits can be implemented using T / 10 shared error amplifiers 810. A characteristic of such embodiments is that sharing error amplifiers 810 among multiple AC-mode bridges in a group can reduce power, area, complexity, etc. The trade-off is that each AC-mode bridge 1220 lacks its own error feedback, making it possible that differences between AC-mode bridges in the group may not be compensated for, eliminated, etc. For example, such differences may arise from common-mode differences in the differential channels coupled to different pairs of different AC-mode bridges, and / or from process differences in the components of different AC-mode bridges.

[0085] Figure 13 A flowchart illustrating an illustrative method 1300 for performing self-capacitance sensing in a touch panel array according to embodiments described herein is shown. As described herein, the embodiments operate in the context of a touch panel array integrated with a display panel. The touch panel array has multiple channels (e.g., touch sensing channels, such as row lines and / or column lines). Each channel has a corresponding channel self-capacitance (Ci), which includes: a corresponding base self-capacitance (Cs) corresponding to display noise coupled from the display panel capacitance to the channel; and a corresponding touch capacitance (Ctouch) that varies in response to the presence of a local touch event in the channel. Method 1300 may represent a channel readout cycle involving differential readout of two channels (e.g., two adjacent channels). For example, an embodiment of method 1300 may begin at a stage 1302 labeled “Start Readout Cycle”.

[0086] An embodiment of method 1300 may continue at stage 1304, whereby a first branch of the AC mode (AC mode) bridge is coupled to the i-th channel of the touch panel array having an i-th channel self-capacitance (Ci_i), and a second branch of the AC mode bridge is coupled to the j-th channel of the touch panel array having a j-th channel self-capacitance (Ci_j). As described above, Ci_i includes an i-th basic self-capacitance (Cs_i) and an i-th touch capacitance (Ctouch_i), the i-th basic self-capacitance (Cs_i) corresponding to the i-th display noise coupled from the display panel capacitance to the i-th channel, and the i-th touch capacitance (Ctouch_i) changing in response to the presence of a local touch event in the i-th channel; Ci_j includes a j-th basic self-capacitance (Cs_j) and a j-th touch capacitance (Ctouch_j), the j-th basic self-capacitance (Cs_j) corresponding to the j-th display noise coupled from the display panel capacitance to the j-th channel, and the j-th touch capacitance (Ctouch_j) changing in response to the presence of a local touch event in the j-th channel. Some embodiments assume that at least a portion of the noise displayed on the i-th and j-th channels is common-mode noise on the i-th and j-th channels, such that the first branch voltage and the second branch voltage include common-mode components corresponding to the common-mode noise.

[0087] In stage 1308, embodiments may generate a loop control voltage in response to a sinusoidal drive signal and an error feedback signal, the error feedback signal being a function of a first branch voltage on a first branch and a second branch voltage on a second branch. Some embodiments of method 1300 include generating a sinusoidal drive signal in stage 1303.

[0088] In stage 1312, embodiments may generate a first sinusoidal current and a second sinusoidal current based on the loop control voltage generated in stage 1308. In some embodiments, the first sinusoidal current is generated by a first adjustable current source based on the loop control voltage; and the second sinusoidal current is generated by a second adjustable current source based on the loop control voltage. In such embodiments, the first adjustable current source and the second adjustable current source may be nominally the same, and the first sinusoidal current and the second sinusoidal current may be nominally the same.

[0089] In some embodiments, generation at stage 1312 involves switching between a first adjustable current source and a second adjustable current source based on a set of chopper clock signals. In the first configuration, generation at stage 1312 may involve generating a first sinusoidal current through a first CMOS portion (e.g., a PMOS portion) of the first adjustable current source and a second CMOS portion (e.g., an NMOS portion) of the second adjustable current source, and generating a second sinusoidal current through the second CMOS portion (e.g., an NMOS portion) of the first adjustable current source and the first CMOS portion (e.g., a PMOS portion) of the second adjustable current source. In the second configuration, generation at stage 1312 may involve generating a second sinusoidal current through the first CMOS portion of the first adjustable current source and the second CMOS portion of the second adjustable current source, and generating a first sinusoidal current through the second CMOS portion of the first adjustable current source and the first CMOS portion of the second adjustable current source.

[0090] In some embodiments, the generation at stage 1312 involves generating N instances of a rotating unit current (Iunit) generated by N nominally identical adjustable current sources. Each current source is configured to output a corresponding instance of Iunit based on a loop control voltage. In such embodiments, a first sinusoidal current can be generated by combining NM instances of the N instances of the rotating Iunit, and a second sinusoidal current can be generated by combining M instances of the N instances of the rotating Iunit, the M instances being different from the NM instances. Some such embodiments also include setting M such that the ratio between NM and M corresponds to the ratio between Cs_i and Cs_j.

[0091] In stage 1316, the embodiment can drive the first branch with a first sinusoidal current to generate a first branch voltage based on Ci_i. In stage 1320, the embodiment can drive the second branch with a second sinusoidal current to generate a second branch voltage based on Ci_j. In stage 1324, the embodiment can generate the i-th output voltage (Vout_i) based on the difference between the first branch voltage and the second branch voltage.

[0092] As shown in the figure, some embodiments of method 1300 may cycle through the various stages, for example, by looping through at least stages 1304-1324. For example, a single readout cycle may include multiple (e.g., two) readout frames. In each readout frame, different pairs of channels of the touch panel array may be read out differentially. In one such embodiment, method 1300 includes reading out the touch panel array by cycling between a first readout frame and a second readout frame. In each first readout frame, at least stages 1308-1324 are performed after stage 1304 by coupling a first branch to a first channel of the touch panel array and coupling a second branch to a second channel of the touch panel array. In each second readout frame, at least stages 1308-1324 are performed after stage 1304 by coupling a first branch to a second channel and coupling a second branch to a third channel of the touch panel array.

[0093] It should be understood that when an element or component is referred to herein as being "connected to" or "coupled to" another element or component, it may be connected to or coupled to another element or component, or there may be intermediate elements or components. Conversely, when an element or component is referred to herein as being "directly connected to" or "directly coupled to" another element or component, there are no intermediate elements or components between them. It should be understood that although the terms "first," "second," "third," etc., may be used herein to describe various elements, components, and regions, these elements, components, and regions should not be limited by these terms. These terms are only used to distinguish an element or component from another element or component. Therefore, without departing from the teachings of the invention, the first element or component discussed below may be technically referred to as the second element or component. As used herein, the terms "logic low," "low state," "low level," "logic low level," "low," or "0" are used interchangeably. The terms "logic high," "high state," "high level," "logic high level," "high," or "1" are used interchangeably.

[0094] As used herein, the terms “a,” “an,” and “the” can include both singular and plural references. It should also be understood that the terms “comprising,” “including,” “having,” and variations thereof, when used in this specification, indicate the presence of the stated features, steps, operations, elements, and / or components, but do not exclude the presence or addition of one or more other features, steps, operations, elements, components, and / or groups thereof. Conversely, the term “comprising,” when used in this specification, indicates the stated features, steps, operations, elements, and / or components, and excludes additional features, steps, operations, elements, and / or components. Furthermore, as used herein, the word “and / or” can refer to and cover any possible combination of one or more associated listed items.

[0095] While the invention has been described herein with reference to illustrative embodiments, this description is not intended to be limiting. Rather, the illustrative embodiments are intended to enable those skilled in the art to better understand the spirit of the invention. Many details of well-known processes and manufacturing techniques have been omitted so as not to obscure the scope of the invention. Various modifications to the illustrative embodiments and other embodiments will be apparent to those skilled in the art upon referring to this description. Therefore, the appended claims are intended to cover any such modifications.

[0096] Furthermore, some features of the preferred embodiments of the invention can be advantageously used without correspondingly using other features. Therefore, the foregoing description should be considered merely as an illustration of the principles of the invention, and not as a limitation thereof. Those skilled in the art will understand variations of the above embodiments that fall within the scope of the invention. Therefore, the invention is not limited to the specific embodiments and descriptions discussed above, but is limited by the appended claims and their equivalents.

Claims

1. A self-capacitance sensing system for performing self-capacitance sensing in a touch panel array integrated with a display panel, the system comprising: AC mode bridge circuits, including: An error amplifier is used to generate a loop control voltage in response to a sinusoidal drive signal and an error feedback signal. The first branch and the second branch are used to couple with the i-th channel of the touch panel array having the i-th channel self-capacitance (Ci_i), and the second branch is used to couple with the j-th channel of the touch panel array having the j-th channel self-capacitance (Ci_j). A set of adjustable current sources is configured to output a first sinusoidal current to the first branch and a second sinusoidal current to the second branch based on the loop control voltage. Wherein, the first branch responds to applying the first sinusoidal current to the self-capacitance of the i-th channel to generate a first branch voltage, the second branch responds to applying the second sinusoidal current to the self-capacitance of the j-th channel to generate a second branch voltage, and the error feedback signal is a function of the first branch voltage and the second branch voltage; and An output amplifier is used to generate the i-th output voltage (Vout_i) based on the difference between the first branch voltage and the second branch voltage.

2. The system according to claim 1, wherein: The i-th channel self-capacitance includes the i-th basic self-capacitance (Cs_i) and the i-th touch capacitance (Ctouch_i). The i-th basic self-capacitance (Cs_i) corresponds to the i-th display noise coupled from the display panel capacitance to the i-th channel. The i-th touch capacitance (Ctouch_i) changes in response to the presence of a local touch event in the i-th channel. The j-th channel self-capacitance includes the j-th basic self-capacitance (Cs_j) and the j-th touch capacitance (Ctouch_j), the j-th basic self-capacitance (Cs_j) corresponding to the j-th display noise coupled from the display panel capacitance to the j-th channel, and the j-th touch capacitance (Ctouch_j) changing in response to the presence of a local touch event in the j-th channel; and At least a portion of the i-th display noise and the j-th display noise are common-mode noise on the i-th channel and the j-th channel, such that the first branch voltage and the second branch voltage include a common-mode component corresponding to the common-mode noise.

3. The system according to claim 1, wherein, The AC mode bridge also includes: An AC source is coupled to the error amplifier and configured to generate the sinusoidal drive signal.

4. The system according to claim 1, wherein, The set of adjustable current sources includes: A first adjustable current source is configured to output a first sinusoidal current based on the loop control voltage; and The second adjustable current source is configured to output the second sinusoidal current based on the loop control voltage. The first adjustable current source and the second adjustable current source are nominally the same, and the first sinusoidal current and the second sinusoidal current are nominally the same.

5. The system according to claim 1, wherein, The set of adjustable current sources includes: A first adjustable current source, comprising a first chopper, a first CMOS section, and a second CMOS section; and The second adjustable current source has a second chopper, a third CMOS section, and a fourth CMOS section, wherein the first chopper and the second chopper are configured to switch between a first configuration and a second configuration based on a set of chopper clock signals, such that: In the first configuration, the first sinusoidal current is generated by the first CMOS portion and the fourth CMOS portion, and the second sinusoidal current is generated by the third CMOS portion and the second CMOS portion; and In the second configuration, the second sinusoidal current is generated by the first CMOS portion and the fourth CMOS portion, and the first sinusoidal current is generated by the third CMOS portion and the second CMOS portion.

6. The system according to claim 1, wherein: The readout of the touch panel array cycles between the first readout frame and the second readout frame; In each first readout frame, the first branch is coupled to the first channel of the touch panel array, and the second branch is coupled to the second channel of the touch panel array; as well as In each second readout frame, the first branch is coupled to the second channel of the touch panel array, and the second branch is coupled to the third channel of the touch panel array.

7. The system according to claim 1, wherein, The set of adjustable current sources includes: N adjustable current sources, each configured to output the nominally identical unit current based on the loop control voltage; and A local current rotator has the following characteristics: N rotator inputs, each rotator input is used to receive a unit current from a corresponding one of the N adjustable current sources; The first rotator output is used to output the first sinusoidal current to the first branch based on the rotational combination of NM adjustable current sources among the N adjustable current sources; and The second rotator outputs a second sinusoidal current to the second branch based on the rotational combination with M adjustable current sources out of the N adjustable current sources. These M adjustable current sources are different from the NM adjustable current sources. Where N is an integer greater than 2, and M is an integer between 1 and N-1.

8. The system according to claim 7, wherein: M is set such that the ratio between NM and M corresponds to the ratio between the i-th basic self-capacitance and the j-th basic self-capacitance.

9. The system according to claim 1, wherein, The AC mode bridge is a main AC mode bridge, and the system further includes: K slave AC-mode bridges, each slave AC-mode bridge coupled to the error amplifier, and comprising: The corresponding first branch is used to couple with the corresponding first channel of the touch panel array; The corresponding second branch is used to couple with the corresponding second channel of the touch panel array; A corresponding set of adjustable current sources is configured to output a corresponding first sinusoidal current to the corresponding first branch and a corresponding second sinusoidal current to the corresponding second branch based on the loop control voltage, such that the corresponding first branch generates a corresponding first branch voltage based on applying the corresponding first sinusoidal current to the corresponding first channel self-capacitance of the corresponding first channel, and the corresponding second branch generates a corresponding second branch voltage based on applying the corresponding second sinusoidal current to the corresponding second channel self-capacitance of the corresponding second channel; and An output amplifier is used to generate a corresponding output voltage based on the difference between the corresponding first branch voltage and the corresponding second branch voltage.

10. The system according to any one of claims 1 to 9, further comprising: The touch panel array has multiple channels including the i-th channel and the j-th channel, and each channel has a corresponding channel self-capacitance; and The AC mode bridge comprises multiple instances, each instance of which is configured to be coupled to a corresponding pair of channels among the multiple channels.

11. The system according to any one of claims 1 to 9, wherein: Multiple channels are T channels; and The AC mode bridge has fewer than T instances.

12. The system according to claim 10, further comprising: The display panel.

13. A self-capacitance sensing method for performing self-capacitance sensing in a touch panel array integrated with a display panel, the method comprising: (a) The first branch of the AC mode bridge is coupled to the i-th channel of the touch panel array having the i-th channel self-capacitance (Ci_i), and the second branch of the AC mode bridge is coupled to the j-th channel of the touch panel array having the j-th channel self-capacitance (Ci_j). (b) In response to a sinusoidal drive signal and an error feedback signal, a loop control voltage is generated, wherein the error feedback signal is a function of the first branch voltage on the first branch and the second branch voltage on the second branch; (c) Based on the loop control voltage, generate a first sinusoidal current and a second sinusoidal current; (d) Drive the first branch with the first sinusoidal current to generate the first branch voltage based on the self-capacitance of the i-th channel; (e) Drive the second branch with the second sinusoidal current to generate the second branch voltage based on the self-capacitance of the j-th channel; as well as (f) Generate the i-th output voltage (Vout_i) based on the difference between the first branch voltage and the second branch voltage.

14. The method of claim 13, wherein: The i-th channel self-capacitance includes the i-th basic self-capacitance (Cs_i) and the i-th touch capacitance (Ctouch_i). The i-th basic self-capacitance (Cs_i) corresponds to the i-th display noise coupled from the display panel capacitance to the i-th channel. The i-th touch capacitance (Ctouch_i) changes in response to the presence of a local touch event in the i-th channel. The j-th channel self-capacitance includes the j-th basic self-capacitance (Cs_j) and the j-th touch capacitance (Ctouch_j), the j-th basic self-capacitance (Cs_j) corresponding to the j-th display noise coupled from the display panel capacitance to the j-th channel, and the j-th touch capacitance (Ctouch_j) changing in response to the presence of a local touch event in the j-th channel; and At least a portion of the i-th display noise and the j-th display noise are common-mode noise on the i-th channel and the j-th channel, such that the first branch voltage and the second branch voltage include a common-mode component corresponding to the common-mode noise.

15. The method of claim 13, further comprising: The sinusoidal drive signal is generated by the AC source.

16. The method of claim 13, wherein: The first sinusoidal current is generated by a first adjustable current source based on the loop control voltage; and The second sinusoidal current is generated by a second adjustable current source based on the loop control voltage. The first adjustable current source and the second adjustable current source are nominally the same, and the first sinusoidal current and the second sinusoidal current are nominally the same.

17. The method according to claim 13, wherein, Based on the loop control voltage, the generation of the first sinusoidal current and the second sinusoidal current includes: Based on a set of chopper clock signals, the first adjustable current source and the second adjustable current source are switched between a first configuration and a second configuration, such that: In the first configuration, generating the first sinusoidal current and the second sinusoidal current includes generating the first sinusoidal current through a first CMOS portion of the first adjustable current source and a second CMOS portion of the second adjustable current source, and generating the second sinusoidal current through a second CMOS portion of the first adjustable current source and a first CMOS portion of the second adjustable current source; and In the second configuration, generating the first sinusoidal current and the second sinusoidal current includes generating the second sinusoidal current through the first CMOS portion of the first adjustable current source and the second CMOS portion of the second adjustable current source, and generating the first sinusoidal current through the second CMOS portion of the first adjustable current source and the first CMOS portion of the second adjustable current source.

18. The method of claim 13, further comprising: The touch panel array is read out by cycling between the first and second readout frames in the following manner: In each first readout frame, steps (a)-(f) are performed, wherein step (a) is performed by coupling the first branch to a first channel of the touch panel array and coupling the second branch to a second channel of the touch panel array; and In each second readout frame, steps (a)-(f) are performed, wherein step (a) is performed by coupling the first branch to the second channel and coupling the second branch to the third channel of the touch panel array.

19. The method according to any one of claims 13 to 18, wherein, Based on the loop control voltage, the generation of the first sinusoidal current and the second sinusoidal current includes: N instances of rotating unit current generated by N nominally identical adjustable current sources are generated, each instance being configured to output a corresponding instance of unit current based on the loop control voltage; The first sinusoidal current is generated by combining NM instances of a rotating unit current; and The second sinusoidal current is generated by combining M instances of N instances of rotating unit current, wherein the M instances are different from the NM instances.

20. The method of claim 19, further comprising: M is set such that the ratio between NM and M corresponds to the ratio between the i-th basic self-capacitance and the j-th basic self-capacitance, where the i-th basic self-capacitance is the basic self-capacitance portion of the i-th channel self-capacitance, which corresponds to the display noise coupled to the i-th channel by the capacitor, and the j-th basic self-capacitance is the basic self-capacitance portion of the j-th channel self-capacitance, which corresponds to the display noise coupled to the j-th channel by the capacitor.

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