Double active bridge converter extended phase-shifted under-inductor current root mean square minimum tracking control strategy

Abstract

The application provides a double active bridge converter extended phase-shift lower inductance current effective value minimum tracking control strategy, solves the optimal working point deviation problem caused by stray resistance loss and switching loss, and realizes real-time tracking of the optimal working point with minimum inductance current effective value. First, the interrelation between phase-shift ratios at the global optimal working point with minimum inductance current is obtained according to the Lagrange multiplier method, the load power is expressed as a function of a certain phase-shift ratio, if the partial derivative of the load power and the phase-shift ratio is positive, the output voltage feedback adopts positive PI to regulate the phase-shift ratio; if the partial derivative of the load power and the phase-shift ratio is negative, the output voltage feedback adopts negative PI to regulate the phase-shift ratio. When the load power is close to the critical power of two optimal modes or the load mutates, free switching between the optimal modes can be realized, so that the converter always works in the state with minimum inductance current effective value, and the output voltage tracks the reference value.

Description

Technical Field

[0001] This invention belongs to the field of power electronics technology, specifically relating to a tracking control strategy for minimizing the effective value of inductor current under extended phase shift in a dual active bridge converter. Background Technology

[0002] With the continuous reduction in the production cost of distributed generation and related policy support, distributed generation has been widely applied. Distributed generation has driven the development of bidirectional DC-DC converters. In the field of bidirectional DC-DC converters, dual-active bridge (DAB) DC-DC converters have attracted widespread attention due to their advantages such as input-output electrical isolation, high power density, bidirectional energy flow, and ease of soft switching.

[0003] When input-output voltage mismatch occurs and a two-level DAB converter uses single-phase-shift (SPS) control, several problems arise, including high current stress, high return power, and difficulty in achieving zero-voltage switching (ZVS). To address these issues, domestic and international researchers have proposed extended phase-shift (EPS), dual phase-shift (DPS), and triple phase-shift (TPS) control by increasing the phase shift angle within the primary or secondary bridge arm of the transformer. These optimizations aim to reduce current stress, lower return power, and achieve ZVS. In medium- and high-voltage applications, two-level DAB switching devices experience high voltage stress. Introducing a multi-level structure into the circuit can increase control freedom, expand the input and output voltage range, reduce voltage stress on the switching transistors, and lower device cost and switching losses by using low-voltage power devices. Furthermore, several basic phase-shift control methods of two-level DAB converters can be extended to three-level DAB converters, resulting in three-level SPS control, three-level EPS control, three-level DPS control, and three-level TPS control. Currently, in the field of three-level DAB converters, engineers mostly optimize the DAB converter based on one or more of the aforementioned control methods, selecting certain optimization targets (such as inductor current stress, RMS current, zero-return power, etc.) to improve the converter's transmission efficiency. The proposed optimization strategies for inductor current stress or RMS value typically involve obtaining the analytical relationship between the shift ratio and the power / voltage transfer ratio for each power segment. Based on the output power segment and voltage transfer ratio, the appropriate shift ratio is selected. This optimization method does not consider stray resistance losses and switching losses present in the converter, causing the actual operating point to deviate from the ideal optimum.

[0004] Therefore, in order to solve the above problems and improve the transmission efficiency of DAB converter, it is necessary to design an effective optimization control strategy to compensate for the stray resistance loss and switching loss of the converter, so that the converter always operates at the optimal operating point. Summary of the Invention

[0005] To address the shortcomings and deficiencies of existing technologies, this invention proposes optimal tracking control strategies for the effective value of inductor current under extended phase-shift control in two-level dual active bridge DC-DC converters and under extended phase-shift control in three-level dual active bridge DC-DC converters, respectively. These strategies overcome the influence of stray resistance and switching losses in the converter, ensuring that the converter always operates at its optimal operating point, achieving minimum tracking control of the effective value of inductor current, reducing the converter's conduction losses, and improving the actual operating efficiency of the converter.

[0006] In this scheme, the proposed minimum effective value tracking control strategy for inductor current under two-level and three-level extended phase-shift control of the dual active bridge DC converter solves the problem of optimal operating point deviation caused by stray resistance loss and switching loss, and achieves real-time tracking of the optimal operating point with the minimum effective value of inductor current. First, the relationship between the phase shift ratios at the globally optimal operating point with the minimum inductor current is obtained using the Lagrange multiplier method. The load power is expressed as a function of a certain phase shift ratio. If the partial derivative of the load power with respect to the phase shift ratio is positive, the output voltage feedback uses a positive PI to adjust the phase shift ratio; if the partial derivative is negative, the output voltage feedback uses a negative PI to adjust the phase shift ratio. When the load power approaches the critical power of the two optimal modes, or in the event of a sudden load change, free switching between optimal modes can be achieved, ensuring that the converter always operates in the state of minimum effective value of inductor current, and the output voltage tracks the reference value.

[0007] The specific technical solution adopted by this invention to solve its technical problem is as follows:

[0008] A dual active bridge converter with extended phase shift and minimum effective value tracking control strategy for inductor current is proposed. First, the relationship between the phase shift ratios at the globally optimal operating point where the inductor current is minimized is obtained using the Lagrange multiplier method. The load power is expressed as a function of a certain phase shift ratio. If the partial derivative of the load power with respect to the phase shift ratio is positive, the output voltage feedback uses a positive PI to adjust the phase shift ratio; if the partial derivative of the load power with respect to the phase shift ratio is negative, the output voltage feedback uses a negative PI to adjust the phase shift ratio. When the load power approaches the critical power of the two optimal modes, or in the event of a sudden load change, the optimal modes are freely switched, so that the converter always operates in the state where the effective value of the inductor current is minimized, and the output voltage tracks the reference value.

[0009] Furthermore, under two-level extended phase-shift control:

[0010] Based on the relative relationship between the inner shift ratio D1 and the outer shift ratio D0 of the two-level extended phase shift, the converter's operating modes are divided into operating mode 1: 0≤D0≤D1 and operating mode 2: D1≤D0≤1. Then, using the maximum power achievable by the converter in full-level single-phase shift control as the power reference and the square of the effective value of the inductor current as the current reference, the per-unit power P of the converter in both operating modes is obtained. 1b and P 2b And the expression for the per-unit value of the square of the effective value of the inductor current, I. 1rms 2 and I 2rms 2 When the transmission power is P* in operating mode 1 or 2, construct the Lagrangian function L. i =I irms 2 +λ(P ib -P * The extreme point where the effective value of the inductor current is minimized satisfies The extreme points where the effective value of the inductor current is minimized under operating modes i, i=1,2, are obtained. These extreme points correspond to the relationship between D0 and D1 at points S1 and S2, respectively. The range of values ​​for D1 and the achievable power range at these extreme points are as follows:

[0011]

[0012]

[0013] Where k = (U i N2 / N1) / U o U i U o Let N1 and N2 be the input voltage and output voltage, respectively, and N1 and N2 be the number of turns in the primary and secondary windings of the transformer, respectively. Then, in operating mode i, i = 1, 2, the per-unit value of the transmitted power is expressed as a function P of D1. ib (k,D1), due to That is, at the optimal solution, P 1b P 2b It is negatively correlated with D1; after subtracting the reference voltage from the output voltage, it passes through a negative PI regulator to obtain D1. Then, the operating mode of the converter is determined based on the output power. Finally, D0 is obtained based on the relationship between D0 and D1 at the optimal point in the corresponding operating mode.

[0014] Furthermore, in the two-level operating mode, when the load power approaches the critical power, the two optimal extreme points automatically switch: the boundary between the optimal extreme points S1 and S2 is: D0 = D1 = (k-1) / k; when the output power per unit value P b Below and close to the critical power point (2k-2) / k 2At that time, due to the effects of stray resistance loss and switching loss, the actual transmitted power will be greater than (2k-2) / k. 2 The state is as follows: If the converter operates at the optimal extreme point S1, D1 continuously decreases, and the transmission power continuously increases. When the controller detects that D1 has decreased to (k-1) / k, the critical point between S1 and S2 has been reached. If the output voltage has not yet reached the reference value at this time, it means that the transmission power is greater than (2k-2) / k. 2 Furthermore, it needs to be increased further, which exceeds the power range of S1. Then it will automatically switch to S2 mode, D1 will continue to decrease, the transmission power will continue to increase, and finally the output voltage will reach the reference value.

[0015] In two-level operating mode, the two optimal extreme points automatically switch between each other when the load changes abruptly: When the converter operates in S1 and the load suddenly increases, the negative PI regulator causes D1 to continuously decrease, and D0 follows the change of D1, increasing the transmission power. When the controller detects that D1 reaches the critical value (k-1) / k between the two optimal modes, if the output voltage has not yet reached the reference value, the converter automatically switches to S2 and D1 continues to decrease, and the transmission power continues to increase, eventually making the output voltage reach the reference value; When the converter operates in S2 and the load suddenly decreases, the negative PI regulator causes D1 to continuously increase, and D0 follows the change of D1, decreasing the transmission power. When the controller detects that D1 reaches the critical value (k-1) / k, if the output voltage has not yet reached the reference value, the converter will automatically switch to S1 and D1 continues to increase, and the transmission power continues to decrease, eventually making the output voltage reach the reference value.

[0016] Furthermore, under three-level extended phase-shift control:

[0017] Based on the relative relationships between the inner shift ratios D1 and D2 and the outer shift ratio D0 in the three-level extended phase shift, the converter's operating modes are divided into operating mode 1: D0≤D1 and operating mode 2: D1≤D0≤D2. Using the maximum power achievable by the converter in full-level single-phase shift control as the power reference and the square of the effective value of the inductor current as the current reference, the per-unit power P of the converter in both operating modes is obtained. 1b and P 2b And the expression for the per-unit value of the square of the effective value of the inductor current, I. 1rms 2 and I 2rms 2 ; Construct the Lagrangian function L when the transmission power is P* in operating mode i, i = 1, 2. i =I irms 2 +λ(P ib -P * The extreme point where the effective value of the inductor current is minimized satisfies When 1≤k≤2, the relationships between D0, D1 and D2 and the power ranges at the extreme points A1 and B1 where the RMS value of the inductor current is the smallest under operation mode 1 and operation mode 2 are obtained after screening respectively; when k≥2, the relationships between D0, D1 and D2 and the power ranges at the extreme points A2 and B2 where the RMS value of the inductor current is the smallest under operation mode 1 and operation mode 2 are obtained after screening respectively as follows:

[0018] When 1<k<2, the expressions of the three-level operation mode 1 solution A1 and the three-level operation mode 2 solution B1 are as follows:

[0019]

[0020]

[0021] When k>2, the expressions of the three-level operation mode one solution A2 and the three-level operation mode two solution B2 are as follows:

[0022]

[0023]

[0024] Then, under operation mode i where i=1,2, the per-unit value of the transmitted power is expressed as a function of D2, P ib (k,D2), and it is calculated that within the (k,D2) value range corresponding to the optimal solutions A1 and A2, It is indicated that under the optimal solutions A1 and A2, P b is negatively correlated with D2, and a negative PI is adopted for tracking A1 and A2; for the optimal solutions B1 and B2, within the corresponding (k,D2) value range, P b is positively correlated with D2, and a positive PI is adopted for tracking B1 and B2; after subtracting the reference voltage from the output voltage and passing through the corresponding PI regulator, D2 is obtained, then the operation mode of the converter at this time is judged according to the output power, and then D0 and D1 are obtained according to the relationship between D0, D1 and D2 at the optimal solution under this operation mode.

[0025] Further, when 1≤k≤2, the load power under three levels is close to the critical power -2(k 2 -3k+2) / k 2 and automatically switches from A1 to B1; at the critical operating point between A1 and B1, D0=D1=0, D2=(k-2) / k, and the critical power is -2(k 2 -3k+2) / k 2When the load power is below and close to the critical power, due to stray resistance losses and switching losses in the converter, the transmission power exceeds the critical power. If the converter operates at the optimal operating point A1, D2 continuously decreases, and the transmission power continuously increases. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, it means that the transmission power is greater than -2(k). 2 -3k+2) / k 2 Furthermore, it needs to be increased further, which exceeds the power range of A1. Then it automatically switches to the optimal point B1, increases D2, and the transmission power continues to increase, eventually making the output voltage reach the reference value.

[0026] When k≥2, the load power under three-level conditions approaches the critical power 2(k-2) / k at the two optimal extreme points. 2 The system automatically switches from A2 to B2; at the critical operating point between A2 and B2, D0 = D1 = D2 = (k-2) / k, and the critical power is 2(k-2) / k. 2 When the load power is below and close to the critical power, the transmission power exceeds the critical power due to stray resistance losses and switching losses in the converter. If the converter operates at the optimal operating point A2, D2 continuously decreases, and the transmission power continuously increases. When the controller detects that D0 = D1 = D2 = (k-2) / k, if the output voltage has not yet reached the reference value, it means that the transmission power is greater than 2(k-2) / k. 2 Furthermore, it needs to be increased further, which exceeds the power range of A2. Then it automatically switches to the optimal point B2, D2 increases, the transmission power continues to increase, and finally the output voltage reaches the reference value.

[0027] In three-level operating mode, the two optimal extreme points can automatically switch between each other when the load changes abruptly: When 1≤k≤2, the converter operates in A1 and the load suddenly increases. The negative PI regulator continuously decreases D2, and D0 and D1 follow D2, increasing the transmission power. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, the converter will automatically switch to B1 and increase D2, continuing to increase the transmission power until the output voltage reaches the reference value. When the converter operates in B1 and the load suddenly decreases, the positive PI regulator continuously decreases D2, and D0 and D1 follow D2, decreasing the transmission power. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, the converter will automatically switch to A1 and increase D2, continuing to increase the transmission power. When the converter operates in A2 mode with k≥2 and the load suddenly increases, the negative PI regulator causes D2 to continuously decrease, and D0 and D1 follow D2's change, increasing the transmission power. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, the converter automatically switches to B2 and D2 increases, further increasing the transmission power until the output voltage reaches the reference value. When the converter operates in B2 mode and the load suddenly decreases, the positive PI regulator causes D2 to continuously decrease, and D0 and D1 follow D2's change, decreasing the transmission power. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, the converter automatically switches to A2 and D2 increases, further decreasing the transmission power until the output voltage reaches the reference value.

[0028] Compared to existing technologies, this invention and its preferred solutions can overcome the effects of stray resistance and switching losses in the converter while minimizing the effective value of the inductor current in both two-level and three-level operating modes. This achieves optimal operating point tracking control with the minimum effective value of the inductor current, and enables optimal mode switching in the event of load changes and the need for critical power compensation. This allows the load power to be stabilized at the set value, reduces the converter's conduction losses, and improves the converter's transmission efficiency. Attached Figure Description

[0029] The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments:

[0030] Figure 1 The following is a topology diagram of the DAB converter according to an embodiment of the present invention:

[0031] Figure 2 The modulation waveform diagram used in this invention is shown in the following embodiment:

[0032] Figure 3 Comparative modulation waveform diagrams for embodiments of the present invention:

[0033] Figure 4 This is a schematic diagram illustrating the three operating modes of the three-level DAB converter according to an embodiment of the present invention:

[0034] Figure 5 This is a schematic diagram illustrating the three operating modes of the two-level DAB converter according to an embodiment of the present invention:

[0035] Figure 6 The six optimal solutions corresponding to the embodiments of the present invention image;

[0036] Figure 7 This is a block diagram of the three-level global optimal extreme point tracking optimization control according to an embodiment of the present invention;

[0037] Figure 8 This is a block diagram of the two-level global optimal extreme point tracking optimization control in an embodiment of the present invention;

[0038] Figure 9 This is an overview diagram of the construction of the embodiment of the present invention. Detailed Implementation

[0039] To make the features and advantages of this patent more apparent and understandable, specific embodiments are provided below for detailed explanation:

[0040] It should be noted that the following detailed descriptions are illustrative and intended to provide further explanation of this application. Unless otherwise specified, all technical and scientific terms used in this specification have the same meaning as commonly understood by one of ordinary skill in the art to which this application pertains.

[0041] like Figures 1-9 As shown, this invention aims to provide an optimal tracking control strategy for the effective value of inductor current under extended phase-shift control in two-level and three-level dual active bridge DC-DC converters. This strategy solves the problem of optimal operating point deviation caused by stray resistance losses and switching losses during actual converter operation, achieving real-time tracking of the optimal operating point with the minimum effective value of inductor current, and reducing converter conduction losses. When the load changes abruptly or critical power compensation is required, the converter can automatically switch to the optimal mode.

[0042] The establishment of an optimized control strategy includes the following steps:

[0043] Step 1: Determine the circuit topology, select the modulation waveform, establish the mathematical model of the converter, optimize the effective value of the inductor current of the converter using the Lagrange multiplier method, and find the relationship between the duty cycles at the global optimal operating point.

[0044] Step 2: Based on Step 1, design an optimized control model to overcome stray resistance loss and switching loss, so as to achieve real-time tracking of the optimal operating point and automatic switching of the optimal mode when the load changes suddenly or when critical power compensation is required.

[0045] In step 1, consider the converter as consisting of a primary-side diode-clamped (NPC) three-level full-bridge H1, a secondary-side two-level full-bridge H2, a high-frequency transformer T, a high-frequency inductor L, input voltage divider capacitors C1 and C2, an output filter capacitor C0, and an input DC voltage source U. i The system comprises a controller, wherein capacitors C1 and C2 have equal capacitance values, and the high-frequency inductor L includes the transformer leakage inductance; the primary-side diode-clamped three-level full-bridge H1 includes fully controlled switching devices S1 to S8 and diode D. c1 ~D c4 Among them, the fully controlled switching devices S1 to S4 and diode D c1 and D c2 S5~S8 and diode D c3 and D c4 Each arm forms a diode-clamped three-level bridge; the secondary two-level full-bridge H2 includes fully controlled switching devices S9 to S1. 12 Topology diagram as follows Figure 1 As shown. u AB u CD These are the output voltages of the primary and secondary bridge arms, u. CD 'is u CD The voltage referred to the primary side of the transformer. Define the primary-to-secondary voltage transformation ratio k = U. i / (nU o The switching frequency is f. s The switching period is T s T hs This is half a switching cycle.

[0046] This topology has a wide range of applications, and can be used for both three-level and two-level optimized control strategies.

[0047] In this embodiment, the three-level optimization control strategy adopts a left-right asymmetric three-level modulation waveform, which includes three shift ratios D0, D1, and D2, such as... Figure 2 As shown in (a), choosing this modulation waveform has two advantages:

[0048] (1) and Figure 3 Compared with the modulation waveform of the two-level EPS control shown in (a), removing the region △S1 to obtain this modulation waveform can result in a smaller current peak-to-peak value, thereby reducing the current stress on the switching transistor.

[0049] (2) and Figure 3 Compared to the left-right symmetrical three-level convex modulation waveform shown in (b), the addition of region △S2 to obtain this modulation waveform can increase the current change rate during this time period, shift the zero-crossing point of the inductor current forward, increase the forward transmission power, and reduce the return power of the primary side.

[0050] Based on the magnitude relationship of the three shift ratios, the three-level DAB converter can be divided into three operating modes: Mode 1: 0≤D0≤D1≤D2≤1; Mode 2: 0≤D1≤D0≤D2≤1; Mode 3: 0≤D1≤D2≤D0≤1. The voltage and current waveforms for each mode are shown below. Figure 4 As shown, compared to working modes 1 and 2, in working mode 3, u AB and u CD There are more regions with opposite polarities, and within these regions, the inductor current is in the range of u. AB and u CD The energy transferred to the load is limited due to the rapid increase in the inductor current under the influence of the current, and the inductor current stress is much greater than the effective value. Therefore, only operating modes 1 and 2 are considered. When the shift ratio D2 = 1, there is no U in the modulation waveform. i / 2 level, the optimized control strategy is transformed into two-level optimized control.

[0051] Mathematical models of the inductor current in the time domain of the converter under three-level operating modes 1 and 2 are established, as shown in Equations 1 and 2, respectively:

[0052]

[0053]

[0054] The effective value of the inductor current and its square have the same maximum and minimum points, and the expression for the square of the effective value of the inductor current is simpler. Therefore, the square of the effective value of the inductor current is used instead of the effective value for optimization. The maximum power transmitted by the DAB converter under full-level single-phase-shift control on the primary side is taken as the reference power value P. N Take the square of the effective value of the inductor current at this time, I N_rms 2 As a reference for the square of the effective value of the inductor current, the reference values ​​are as follows:

[0055]

[0056] Therefore, the expressions for the per-unit values ​​of the RMS square of the inductor current and the per-unit value of the transmitted power are derived for three-level operating modes 1 and 2. The per-unit value of the RMS square of the inductor current in three-level operating mode 1 is:

[0057]

[0058] In three-level operating mode 2, the per-unit value of the RMS square of the inductor current is:

[0059]

[0060] The per-unit transmission power value for three-level operating mode 1 is:

[0061]

[0062] The per-unit value of the transmission power under three-level operation mode 2 is:

[0063]

[0064] Based on the above mathematical model of the three-level operation mode, the Lagrange multiplier method is used to find the mutual relationship between the phase shift ratios at the extreme point where the RMS value of the inductor current is the minimum. According to the three-level operation mode constraints and power constraints, the global optimal extreme point is screened out, and then by using the numerical method to compare with the global boundary points, the mutual relationship between the phase shift ratios at the global optimal operating point can be obtained, and the following four global optimal solutions in the three-level operation mode can be obtained.

[0065] When 1<k<2, there are two global optimal solutions, which are the solution A1 of three-level operation mode 1 and the solution B1 of three-level operation mode 2, and the expressions are as follows:

[0066]

[0067]

[0068] When k>2, there are two global optimal solutions, which are the solution A2 of three-level operation mode 1 and the solution B2 of three-level operation mode 2, and the expressions are as follows:

[0069]

[0070]

[0071] To establish each mathematical model of the converter under the two-level operation mode, it is only necessary to set the phase shift ratio D2 in the mathematical model obtained under the three-level operation mode of the converter to 1, so as to obtain each mathematical model of the converter under the two-level operation mode, as follows:

[0072] The per-unit value of the square of the RMS inductor current under two-level operation mode 1 is:

[0073]

[0074] The per-unit value of the square of the RMS inductor current under two-level operation mode 2 is:

[0075]

[0076] The per-unit value of the transmission power under two-level operation mode 1 is:

[0077]

[0078] The per-unit value of the transmission power under two-level operation mode 2 is:

[0079]

[0080] Based on the mathematical model of the two-level operating mode described above, a calculation process similar to that of the three-level operating mode is performed. Under the mode constraints and power constraints of the two-level operating mode, two globally optimal solutions for the two-level operating mode can be obtained as shown below.

[0081]

[0082]

[0083] In step 2, based on the four globally optimal solutions for the three-level operating mode and the two globally optimal solutions for the two-level operating mode obtained in step 1, a set of minimum RMS tracking control systems for the inductor current of a three-level dual active bridge DC-DC converter and a set of minimum RMS tracking control systems for the inductor current of a two-level dual active bridge DC-DC converter can be designed, employing output voltage feedback control. In the three-level operating mode, the output voltage error u e The shift ratio to the reference D2 is obtained through the PI regulator. * Compared to D0 and D1, the shift follows D2 in the form of the optimal solution. * Changes; in two-level operating mode, the output voltage error u e The shift ratio D1 is obtained through the PI regulator. * The shift follows D1 in the form of the optimal solution compared to D0. * change.

[0084] First, the positive and negative PI values ​​of the four optimal solution modes of the three-level circuit must be determined:

[0085] Substituting the relationships between the four optimal solutions corresponding to equations 8, 9, 10, and 11 into equations 6 and 7 respectively, the transmission power P can be obtained. b The analytical expression P for D2 and k b (k, D2). Within the (k, D2) range corresponding to each of the four optimal solutions, P is obtained. b Partial derivative with respect to D2 The partial derivatives of each are expressed as follows:

[0086]

[0087]

[0088]

[0089]

[0090] Based on the analytical relationships of Equations 18 to 21, their respective partial derivative graphs are plotted, as follows: Figure 6As shown in (a) to (d), it can be seen from the figure that within the (k, D2) value range corresponding to the optimal solutions A1 and A2, Explain that under optimal solutions A1 and A2, P b Negatively correlated with D2, the optimal solution patterns corresponding to solutions A1 and A2 should be adjusted using negative PI; for optimal solutions B1 and B2, within the corresponding (k, D2) value range... P b Positively correlated with D2, the optimal solution modes corresponding to solutions B1 and B2 should be adjusted using positive PI.

[0091] Determining the positive and negative PI values ​​for the two optimal solution modes of the two-level circuit:

[0092] Find P b Partial derivatives with respect to D1 The analytical expression is as follows:

[0093]

[0094]

[0095] Based on the analytical relationships of Equations 22 to 23, their respective partial derivative graphs are plotted, as follows: Figure 6 As shown in (e) to (f). It can be seen from the figure that within the (k, D1) value range corresponding to the optimal solutions S1 and S2, Explain that under optimal solutions S1 and S2, P b It is negatively correlated with D1, and its corresponding optimal solution mode should be adjusted using negative PI.

[0096] The block diagram of the optimal tracking control of the inductor current RMS value in three-level operating mode is as follows: Figure 7 As shown, sampling U i U o and I o To obtain k and P b With error voltage u e Then, for k and P b The size of u is used to determine the optimal matching solution pattern. e The duty cycle D2 is obtained through a PI regulator, and D0 and D1 are then obtained based on the interrelationships of the current optimal solution mode, achieving tracking control of the optimal operating point. The block diagram of the optimal tracking control of the inductor current RMS value in the two-level operating mode is as follows: Figure 8 As shown, sampling U i U o and I o To obtain k and P b With error voltage u e Afterwards, regarding P b The size of u is used to determine the optimal matching solution pattern. eThe duty cycle D1 is obtained through the PI controller, and D0 is obtained based on the interrelationship of the current optimal solution mode, thus realizing the tracking control of the optimal operating point.

[0097] Typical inductor current stress or RMS value optimization control strategies, taking three-level operating mode as an example, involve obtaining the optimal shift ratio D for each power segment. i * With power P b The analytical relationship of the voltage transfer ratio k is shown in the following formula:

[0098] D i * =f i (k,P b ), where i = 0, 1, 2 (Equation 24)

[0099] The converter is based on the set output power P b Given the location of the section and the voltage transfer ratio k, select the corresponding optimal shift ratio D. i * Its PI controller typically controls the output outward shift ratio D0, while the remaining inward shift ratios are determined by Equation 24. Ideally, there is no power loss in the circuit, and the outward shift ratio signal D0 obtained from the PI controller output is equal to the optimal shift ratio signal D0 obtained from Equation 24. * When they are equal, all shift ratios are optimal, and the converter operates at its optimal operating point (D0). * D1 * D2 * In practice, due to stray resistance and switching losses in the circuit, the outward shift ratio D0 of the PI controller output will deviate from the optimal shift ratio D0. * A certain distance is used to compensate for stray resistance losses and switching losses in the circuit, as shown in the following formula:

[0100] D0 = D0 * ±Δd Equation 25

[0101] At this point, the actual operating point of the DAB converter is (D0, D1). * D2 * This deviates from the optimal operating point (D0). * D1 * D2 * ).

[0102] In this invention, when the PI controller controls the output inner shift ratio signal, it has already compensated for stray losses and switching losses in the circuit. In the three-level operating mode, the inner shift ratio signal D2 output by the PI controller is already the optimal shift ratio signal D2. * D0 and D1 follow D2 in the form of their respective optimal solutions. *change, the obtained phase-shift ratios are all the optimal phase-shift ratio D0 * , D1 * , and the operating point of the converter is the optimal operating point (D0 * , D1 * , D2 * ); in the two-level operating mode, the inner phase-shift ratio signal D1 output by the PI controller is already the optimal phase-shift ratio signal D1 * . D0 follows D1 in the form of the corresponding optimal solution * to change, the obtained phase-shift ratio is the optimal phase-shift ratio D0 * , and the operating point of the converter is the optimal operating point (D0 * , D1 * ). The converter always operates in the optimal state with the minimum RMS value of the inductor current, which realizes the minimum RMS value tracking control of the inductor current, reduces the on-state loss of the converter, and improves the actual operating efficiency of the converter.

[0103] In addition, the present invention also designs and implements the function of optimal mode switching. When the load of the converter suddenly changes or critical power compensation is required, each phase-shift ratio changes continuously. When the value of the duty cycle reaches the critical value and the operating state of the converter is still not stable, the converter automatically switches to a more matched optimal mode for continuous regulation until the converter operates stably.

[0104] In the three-level operating mode, the implementation of optimal mode switching during critical power compensation is as follows:

[0105] When 1<k<2 and k>2, the converter has two optimal modes, and there is a critical power point in both cases. When the converter operates at a position slightly lower than the critical power point, due to the existence of stray resistance loss and switching loss in the converter, the power that the power supply needs to deliver may be higher than the critical power point. At this time, critical power compensation is required, and the converter needs to switch to the optimal mode with higher transmission power to compensate for the lost power.

[0106] When 1<k<2, the boundary between the two optimal solution modes A1 and B1 is: D2=(2-k) / k, D0=D1=0; when P b is slightly lower than the critical power point -2(k 2 -3k+2) / k 2 , due to the existence of stray resistance loss and switching loss, the actual transmitted power may be greater than -2(k 2 -3k+2) / k 2, when the converter operates in mode A1, D2 decreases continuously and the transmitted power increases continuously. When the controller detects that D2 decreases to (2-k) / k and D0=D1=0, the converter reaches the critical state between the optimal mode A1 and B1. If the actual power consumed by the load side has not reached the set value at this time, the converter will automatically switch to mode B1 to operate, the transmitted power increases to compensate for stray resistance loss and switching loss, so that the actual power consumed by the load side is stabilized at the set value. Similarly, when k>2, the boundary between the two optimal solution modes A2 and B2 is: D2=(k-2) / k, D0=D1=D2; when P b is slightly lower than the critical power point (2k-4) / k 2 , due to the existence of stray resistance loss and switching loss, the actual transmitted power may be greater than (2k-4) / k 2 , when operating in mode A2, D2 decreases continuously and the transmitted power increases continuously. When the controller detects that D2 decreases to (k-2) / k and D0=D1=D2, the converter reaches the critical state between the optimal mode A2 and B2. If the actual power consumed by the load side has not reached the set value at this time, the converter will automatically switch to mode B1 to operate, so that the transmitted power increases to compensate for stray resistance loss and switching loss, thereby stabilizing the actual power consumed by the load side at the set value. In this way, the switching of the optimal mode during critical power compensation under the conditions of 1<k<2 and k>2 is realized.

[0107] The realization of optimal mode switching upon load sudden change in the three-level operating mode is as follows:

[0108] When the load of the converter suddenly increases or decreases, the PI regulator continuously adjusts D2, and D0 and D1 continuously change following D2. When the controller detects that the value of the duty cycle reaches the critical state of the two optimal modes, and the power consumed by the load side has not reached the set value, the converter will automatically switch to a more matched optimal mode for continuous adjustment, until the actual power consumed by the load side is stabilized at the set value and the converter operates stably, thus realizing the optimal mode switching upon load sudden change.

[0109] The realization of optimal mode switching during critical power compensation in the two-level operating mode is as follows:

[0110] When k>1, the converter has two optimal modes, and a critical power point exists. When the converter operates at a position slightly lower than the critical power point, due to the existence of stray resistance loss and switching loss in the converter, the power that the power supply needs to output may be higher than the critical power point. At this time, critical power compensation is required, and the converter needs to switch to an optimal mode with higher transmitted power to compensate for the loss power.

[0111] The boundary between the two optimal solution modes S1 and S2 is: D0=D1=(k-1) / k; when P bSlightly below the critical power point (2k-2) / k 2 At that time, due to stray resistance losses and switching losses, the actual transmitted power may be greater than (2k-2) / k. 2 If the converter operates in S1 mode, D1 continuously decreases, and the transmission power continuously increases. When the controller detects that D1 has decreased to (k-1) / k, the converter has reached the critical state between optimal mode S1 and S2. If the actual power consumed by the load side has not yet reached the set value at this time, the converter will automatically switch to S2 mode, increasing the transmission power to compensate for stray resistance losses and switching losses, thereby stabilizing the actual power consumed by the load side at the set value. In this way, the switching of the optimal mode during critical power compensation in two-level operating modes is realized.

[0112] In two-level operating mode, the optimal mode switching is implemented when there is a sudden change in load as follows:

[0113] When the load on the converter suddenly increases or decreases, the PI regulator continuously adjusts D1, and D0 continuously follows the change of D1. When the controller detects that the duty cycle value has reached the critical state between the two optimal modes, and the power consumed by the load side has not yet reached the set value, the converter will automatically switch to the more suitable optimal mode to continue adjusting until the actual power consumed by the load side stabilizes at the set value, and the converter works stably. In this way, the optimal mode switching is realized when the load changes suddenly.

[0114] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention in any other way. Any person skilled in the art may make changes or modifications to the above-disclosed technical content to create equivalent embodiments. However, any simple modifications, equivalent changes, and modifications made to the above embodiments based on the technical essence of the present invention without departing from the scope of the present invention shall still fall within the protection scope of the present invention.

[0115] This patent is not limited to the above-described preferred embodiments. Anyone can derive other forms of dual active bridge converter extended phase-shift inductor current minimum tracking control strategies based on the guidance of this patent. All equivalent changes and modifications made within the scope of this patent application shall fall within the scope of this patent.

Claims

1. A tracking control strategy for minimizing the effective value of inductor current under extended phase shift in a dual active bridge converter, characterized in that: First, the mutual relationship between phase-shift ratios at the globally optimal operating point with the minimum inductor current is obtained according to the Lagrange multiplier method, and the load power is expressed as a function of a certain phase-shift ratio. If the partial derivative of the load power with respect to this phase-shift ratio is positive, the output voltage feedback uses a positive PI to adjust this phase-shift ratio; if the partial derivative of the load power with respect to the phase-shift ratio is negative, the output voltage feedback uses a negative PI to adjust this phase-shift ratio; when the load power is close to the critical power of the two optimal modes, or in the case of a sudden load change, free switching between the optimal modes is performed, so that the converter always operates in a state where the RMS value of the inductor current is minimum, and the output voltage tracks the reference value; Under the two-level extended phase-shift control: Based on the relative relationship between the inner shift ratio D1 and the outer shift ratio D0 of the two-level extended phase shift, the converter's operating modes are divided into operating mode 1: 0≤D0≤D1 and operating mode 2: D1≤D0≤1. Then, using the maximum power achievable by the converter in full-level single-phase shift control as the power reference and the square of the effective value of the inductor current as the current reference, the per-unit power P of the converter in both operating modes is obtained. 1b and P 2b And the expression for the per-unit value of the square of the effective value of the inductor current, I. 1rms 2 and I 2rms 2 When the transmission power is P* in operating mode 1 or 2, construct the Lagrangian function L. i =I irms 2 +λ(P ib -P * The extreme point where the effective value of the inductor current is minimized satisfies ∂L / 2π. i / ∂D0=∂L i With ∂D1=0, the extreme points where the effective value of the inductor current is minimized in operating modes i, i=1,2 are obtained. These extreme points correspond to the relationship between D0 and D1 at points S1 and S2, respectively. The range of D1 values ​​and the achievable power range at these extreme points are as follows: Where k=(U i N2 / N1) / U o U i U o Let N1 and N2 be the input voltage and output voltage, respectively, and N1 and N2 be the number of turns in the primary and secondary windings of the transformer, respectively. Then, in operating mode i, i=1,2, the per-unit value of the transmitted power is expressed as a function P of D1. ib (k,D1), due to ∂P 1b (k,D1) / ∂D1<0、∂P 2b (k,D1) / ∂D1<0, that is, at the optimal solution, P 1b P 2b It is negatively correlated with D1; after subtracting the reference voltage from the output voltage, it passes through a negative PI regulator to obtain D1. Then, the operating mode of the converter is determined based on the output power. Finally, D0 is obtained based on the relationship between D0 and D1 at the optimal point in the corresponding operating mode. Under the three-level extended phase-shift control: Based on the relative relationships between the inner shift ratios D1 and D2 and the outer shift ratio D0 in the three-level extended phase shift, the converter's operating modes are divided into operating mode 1: D0≤D1 and operating mode 2: D1≤D0≤D2. Using the maximum power achievable by the converter in full-level single-phase shift control as the power reference and the square of the effective value of the inductor current as the current reference, the per-unit power P of the converter in both operating modes is obtained. 1b and P 2b And the expression for the per-unit value of the square of the effective value of the inductor current, I. 1rms 2 and I 2rms 2 In operating mode i, i=1,2, with a transmission power of P*, construct the Lagrangian function L. i =I irms 2 +λ(P ib -P * The extreme point where the effective value of the inductor current is minimized satisfies ∂L / 2π. i / ∂D0=∂L i / ∂D1=∂L i When ∂D2=0, and 1≤k≤2, the relationships and power ranges of D0, D1, and D2 at the extreme points A1 and B1 where the effective value of the inductor current is minimum under operating mode 1 and operating mode 2 are obtained after screening. When k≥2, the relationships and power ranges of D0, D1, and D2 at the extreme points A2 and B2 where the effective value of the inductor current is minimum under operating mode 1 and operating mode 2 are obtained as follows: When 1<k<2, the solution A1 of the three-level operating mode 1 and the solution B1 of the three-level operating mode 2 are expressed as follows: When k>2, the solution A2 of the three-level operating mode 1 and the solution B2 of the three-level operating mode 2 are expressed as follows: Next, in operating mode i, i=1,2, the per-unit value of the transmission power is expressed as a function P of D2. ib (k,D2), the calculated value of ∂P within the (k,D2) range corresponding to the optimal solutions A1 and A2 is obtained. ib (k,D2) / ∂D2<0, indicating that under optimal solutions A1 and A2, P b Negatively correlated with D2, negative PI is used to track A1 and A2; for optimal solutions B1 and B2, within the corresponding (k, D2) value range, ∂P ib (k,D2) / ∂D2>0, P b It is positively correlated with D2, and positive PI is used to track B1 and B2; after subtracting the reference voltage from the output voltage, it passes through the corresponding PI regulator to obtain D2. Then, the working mode of the converter is determined according to the output power. Then, according to the optimal solution of D0, D1 and D2 under this working mode, D0 and D1 are obtained.

2. The minimum RMS inductor current tracking control strategy under extended phase-shift of a dual active bridge converter according to claim 1, characterized in that: When 1≤k≤2, the load power under three-level conditions approaches the critical power -2(k) at the two optimal extreme points. 2 -3k+2) / k 2 Automatically switches from A1 to B1; at the critical operating point between A1 and B1, D0=D1=0, D2=(k-2) / k, and the critical power is -2(k). 2 -3k+2) / k 2 When the load power is below and close to the critical power, due to stray resistance losses and switching losses in the converter, the transmission power exceeds the critical power. If the converter operates at the optimal operating point A1, D2 continuously decreases, and the transmission power continuously increases. When the controller detects that D2 has decreased to (k-2) / k, if the output voltage has not yet reached the reference value, it means that the transmission power is greater than -2(k). 2 -3k+2) / k 2 Furthermore, it needs to be increased further, which exceeds the power range of A1. Then it automatically switches to the optimal point B1, increases D2, and the transmission power continues to increase, eventually making the output voltage reach the reference value. When k≥2, the load power under three-level conditions approaches the critical power 2(k-2) / k at the two optimal extreme points. 2 Automatically switches from A2 to B2; at the critical operating point between A2 and B2, D0=D1=D2=(k-2) / k, and the critical power is 2(k-2) / k. 2 When the load power is below and close to the critical power, due to stray resistance losses and switching losses in the converter, the transmission power exceeds the critical power. If the converter operates at the optimal operating point A2, D2 continuously decreases, and the transmission power continuously increases. When the controller detects that D0=D1=D2=(k-2) / k, if the output voltage has not yet reached the reference value, it means that the transmission power is greater than 2(k-2) / k. 2 Furthermore, it needs to be increased further, which exceeds the power range of A2. Then it automatically switches to the optimal point B2, D2 increases, the transmission power continues to increase, and finally the output voltage reaches the reference value. Under the three-level operating mode, the two optimal extreme points are allowed to automatically switch to each other when the load changes suddenly: when 1≤k≤2 and the converter operates at A1 with a sudden load increase, the negative PI regulator continuously reduces D2, D0 and D1 change following D2 to increase the transmission power; when the controller detects that D2 is reduced to (k-2) / k, if the output voltage has not reached the reference value, the converter will automatically switch to B1, D2 increases, the transmission power continues to increase, and finally the output voltage reaches the reference value; when the converter operates at B1 with a sudden load decrease, the positive PI regulator continuously reduces D2, D0 and D1 change following D2 to reduce the transmission power; when the controller detects that D2 is reduced to (k-2) / k, if the output voltage has not reached the reference value, the converter will automatically switch to A1, D2 increases, the transmission power continues to decrease, and finally the output voltage reaches the reference value; when k≥2 and the converter operates at A2 with a sudden load increase, the negative PI regulator continuously reduces D2, D0 and D1 change following D2 to increase the transmission power; when the controller detects that D2 is reduced to (k-2) / k, if the output voltage has not reached the reference value, the converter will automatically switch to B2, D2 increases, the transmission power continues to increase, and finally the output voltage reaches the reference value; when the converter operates at B2 with a sudden load decrease, the positive PI regulator continuously reduces D2, D0 and D1 change following D2 to reduce the transmission power; when the controller detects that D2 is reduced to (k-2) / k, if the output voltage has not reached the reference value, the converter will automatically switch to A2, D2 increases, the transmission power continues to decrease, and finally the output voltage reaches the reference value.

Images