A single-directional or bidirectional WPT system and a control method thereof

Abstract

The application relates to the technical field of wireless power transmission (WPT), and particularly discloses a single / dual-direction WPT system and a control method thereof, aiming at the problem that the efficiency of a traditional E-class dual-direction wireless power transmission system is easily affected by a load, an improved E-class power amplifier topology structure composed of switching tubes Q1 and Q2 or composed of switching tubes Q3 and Q4 is provided, the topology can effectively transfer input current oscillation to the ground by resonating the resonant capacitor and the resonant inductor in parallel with the switching tube, so as to reduce the ripple of the input current, and the topology has the characteristics of zero-voltage turn-on and wide load range. In view of the problem that the coupling coefficient changes due to the actual deviation of a coupling device, the switching tube enters a hard switching state, and the overall efficiency is reduced, the phase of the gate drive signal of the switching tube is controlled, the energy stored in the resonant element is released in advance or delayed, the switching tube is corrected back to a soft switching state, and the system is always operated in an optimal working efficiency state.

Description

Technical Field

[0001] This invention relates to the field of wireless power transfer (WPT) technology, and more particularly to a one-way and two-way WPT system and its control method. Background Technology

[0002] In recent years, with the accelerated global deployment and development of wireless power transfer (WPT), wireless power transfer technology has been widely used in various power industries. Among these, bidirectional WPT technology, which enables power exchange and information transfer between power sources and loads, has attracted significant industry attention. Replacing traditional plug-and-play power supply with WPT technology avoids irreversible damage caused by mechanical friction, thus preventing reduced equipment lifespan. Furthermore, improving the interaction capabilities between power sources and loads makes the entire system more efficient, safe, and stable. Therefore, continued in-depth research into bidirectional WPT technology is of great importance.

[0003] Class-E converters possess the characteristic of charging and discharging the parallel capacitor of the switching devices, achieving zero-voltage switching at the moment of conduction. Furthermore, their symmetrical characteristics enable high-frequency inversion and rectification, making them suitable for bidirectional WPT (Wide-to-Pitch) operation, which has garnered favor among industry researchers. Traditional Class-E bidirectional WPT systems suffer from low efficiency and limited load range. To effectively balance efficiency and load range, the introduction of high-frequency resonant soft-switching technology is crucial. In practical applications, it is necessary to ensure the conversion efficiency of the Class-E converter and a stable output voltage or current. Therefore, many research teams focus on design parameters and subsystem optimization to achieve higher system efficiency.

[0004] Some literature adds two Buck converters before the Class E inverter and after the Class D rectifier to achieve a stable output voltage in a WPT system; however, the addition of extra converters leads to a decrease in overall efficiency. Other literature presents a design scheme for a Class E WPT system and applies it to a robotic arm, achieving zero-voltage switching and constant-voltage output, but does not consider interference caused by the offset of the coupling coils. Some literature derives the input impedance of the Class E rectifier and improves overall efficiency through numerical design algorithms, but does not consider the hard-turn-on effect caused by the load. Some literature designs a multi-coil WPT system with multiple inputs and multiple receivers, using phase-shift control to cancel input ripple interference, reduce input current ripple, and improve efficiency. Still other literature proposes a control scheme that maintains high efficiency of the Class E WPT system at MHz operating frequencies; however, in practical applications, uncertainties such as various power requirements, coil relative positions, and load characteristics make control very complex. When the distance between the transmitter and receiver changes, some literature proposes using a novel search algorithm-controlled adaptive capacity array to adjust the efficiency of WPT and Class E inverters. However, this requires adding extra switching transistors and passive components to achieve system resonance, making control complex and uneconomical. Other literature proposes a scheme for frequency modulation and phase shift control of the Class E converter, which can control the forward and reverse operating directions by controlling the frequency and phase shift. However, the frequency modulation control method significantly impacts the ZVS safety margin. Summary of the Invention

[0005] This invention provides a single- and two-way WPT system and its control method, which solves the technical problem that traditional Class E two-way WPT systems have low efficiency and small load range.

[0006] To solve the above technical problems, the present invention provides a unidirectional WPT system, including a transmitter and a receiver, and also a control terminal;

[0007] The transmitter includes a DC power supply, an inverter circuit, and a transmitter resonant circuit connected in sequence. The inverter circuit includes a switching transistor Q1, and the transmitter resonant circuit includes a resonant inductor L. f1 Transmitting coil L t1 Resonant capacitor C f1 and resonant capacitor C t1 The source of the switching transistor Q1 is connected to the negative terminal of the DC power supply and the resonant capacitor C. f1 At one end, the drain of the switching transistor Q1 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 one end and transmitting coil L t1 One end, resonant capacitor C t1 The other end and the transmitting coil L t1 The other end is connected to the resonant inductor L.f1 One end, resonant inductor L f1 The other end is connected to the positive terminal of the DC power supply, and the gate of the switching transistor Q1 is connected to the control terminal;

[0008] The receiving end includes a receiving end resonant circuit, a rectifier circuit, and a load circuit connected in sequence. The rectifier circuit includes switching transistors Q3 and Q4. The receiving end resonant circuit includes a resonant inductor L. f2 Receiver coil L t2 Resonant capacitor C f2 and resonant capacitor C t2 The source of the switching transistor Q3 is connected to the resonant capacitor C. t2 One end, receiving coil L t2 One end and resonant capacitor C f2 At one end, the drain of switching transistor Q3 is connected to the drain of switching transistor Q4, and the source of switching transistor Q4 is connected to the resonant capacitor C. f2 The other end, one end of the load circuit, is connected to the resonant inductor L. f2 One end, resonant inductor L f2 The other end is connected to the receiving coil L t2 The other end and the resonant capacitor C f2 At the other end, the gates of switching transistors Q3 and Q4 are connected to the control terminal;

[0009] The control terminal controls the gate drive signal of switch Q1. The gate signals of switches Q3 and Q4 are obtained from the current signal of the receiving terminal and are turned on or off simultaneously. The specific operating conditions are as follows:

[0010]

[0011] I Ct1 I Lt1 These represent the resonant capacitance C. t1 and transmitting coil L t1 The current, I o I represents the load current in the load circuit. co This represents the current in the filter capacitor of the load circuit, wherein the load and the filter capacitor are connected in parallel. This indicates the initial phase of the current at the receiving end.

[0012] Specifically, the resonant capacitor C f1 C f2 C t1 C t2 Resonant inductor L f1 L f2 transmitting coil L t1 and receiving coil L t2 The parameters are designed as follows:

[0013]

[0014] Among them, f in It is the input resonant frequency, ω represents the system's operating angular frequency, Q is the quality factor, and R... L This is the load resistance.

[0015] Specifically, the control terminal controls the gate signal of switch Q1, and switches Q3 and Q4 are simultaneously turned on or off, thereby controlling the unidirectional WPT system to sequentially enter the following five modes in each cycle:

[0016] Mode I from time t0 to time t1: At time t0, the drive signal for switch Q1 is high, and the resonant inductor L... f1 Current I in transmitting coil L t1 Current I lt1 and resonant capacitor C t1 Current I ct1 The current flows through the switch Q1, and the resonant capacitor C... f1 The circuit is short-circuited by switch Q1; the drive signals for switches Q3 and Q4 are also high at this time, and the receiving coil L... t2 Current I lt2 Resonant capacitor C t2 Current I ct2 and resonant inductance L f2 Current I on s The current flows through switching transistors Q3 and Q4, and the resonant capacitor C f2 Short-circuited;

[0017] Mode II from time t1 to time t2: At time t1, switch Q1 is turned on, and the input current I... in The flow increases positively and reaches its maximum value, flowing through the transmitting coil L. t1 Current I it1 From positive to negative, the resonant capacitance C t1 Current I ct1 From negative to positive; receiving coil L t2 Current I lt2 From negative to positive, the resonant capacitance C t2 The current changes from positive to negative;

[0018] Mode III from time t2 to t3: At time t2, switch Q1 is turned off, and the drain-source voltage across switch Q1 increases. At this time, the transmitting coil L... t1 and resonant capacitor C t1 Resonance begins, I lt1 Positive increase, I ct1 Decrease, transmitting coil L t1 At this time, the receiving coil L is supplied with...t2 Energy is transferred; at time t3, the voltage across switch Q1 reaches its maximum value. When switch Q1 is off, the resonant inductor L... f1 and resonant capacitor C f1 Resonance occurs, current I in The resonant capacitance C decreases linearly. f1 Accumulated charge; transmitting coil L t1 With resonant capacitor C t1 Resonance occurs, I lt1 Decrease in the opposite direction, C t1 Accumulate charge; with switches Q3 and Q4 in the off state, the receiving coil L... t2 With resonant capacitor C t2 Resonance occurs, I lt2 Decrease in the opposite direction, C t2 Accumulated charge;

[0019] Mode IV from time t3 to time t4: resonant capacitance C f1 and resonant capacitor C t1 After the charge on the transistor Q1 is completely discharged, the drain-source voltage V of the switching transistor Q1 will decrease. ds1 The value is zero, and the switching transistor Q1 enters soft-switching mode; the resonant capacitor C... f2 and C t2 After the charge on the transistors is completely released, the drain-source voltage V across the switching transistors Q3 and Q4 will decrease. ds2 When the value is zero, switching transistors Q3 and Q4 enter soft-switching mode;

[0020] Mode V from time t4 to t5: At time t4, switches Q1, Q3, and Q4 enter the ZVS state, and the transmitting coil L... t1 Current I on lt1 and resonant inductance L f1 Current I in The diode of switch Q1 provides reverse freewheeling, at which point the voltage across switch Q1 is zero, and the drain-source voltages across switches Q3 and Q4 are also zero. The system waits for the drive signals of switches Q1, Q3, and Q4 to go high again in the next cycle, thus achieving zero-voltage turn-on.

[0021] Specifically, the present invention also provides a control method for a unidirectional WPT system, comprising: during power transmission, the control terminal is further used to sample the resonant inductor L. f2 Current I s and the drain-source voltage V across switching transistors Q3 and Q4 ds2(θ), and determine whether the switching transistors Q3 and Q4 are operating at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q3 and Q4 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q3 and Q4 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lag conduction angle δ.

[0022] Specifically, the conduction angle δ is calculated using the following formula:

[0023]

[0024]

[0025] θ = ωt represents the original conduction angle.

[0026] This invention also provides a bidirectional WPT system, the key features of which are: a transmitting and receiving end and a receiving and transmitting end, and a bidirectional charging control end; the transmitting and receiving end includes a power load circuit, an inverter rectifier circuit, and a first resonant circuit connected in sequence, and the receiving and transmitting end includes a second resonant circuit, a rectifier inverter circuit, and a load power supply circuit connected in sequence; the second resonant circuit adopts the receiving end resonant circuit in the above-mentioned WPT system; the first resonant circuit adopts the transmitting end resonant circuit in the above-mentioned WPT system, and further includes a switching transistor Q2, the drain of the switching transistor Q2 is connected to the drain of the switching transistor Q1, and the source of the switching transistor Q2 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 one end and transmitting coil L t1 At one end, the gates of switching transistors Q1, Q2, Q3, and Q4 are connected to the bidirectional charging control terminal;

[0027] The bidirectional charging control terminal is used to control the switching transistor Q2 to be in normally open mode during forward transmission, control the gate drive signal of the switching transistor Q1, and the gate signals of Q3 and Q4 are obtained from the current signal of the receiving end and are turned on or off simultaneously. The power supply load circuit, the inverter rectifier circuit, the rectifier inverter circuit, and the load power supply circuit are respectively switched to the DC power supply, inverter circuit, rectifier circuit, and load circuit in the WPT system mentioned above.

[0028] The bidirectional charging control terminal is used to control the switching transistor Q3 to be in normally open mode during reverse transmission, and to control the gate drive signal of the switching transistor Q4. The gate signals of Q1 and Q2 are obtained from the current signal of the receiving end and are turned on or off simultaneously. The power supply load circuit, the inverter rectifier circuit, the rectifier inverter circuit, and the load power supply circuit are respectively switched to the load circuit, rectifier circuit, inverter circuit, and DC power supply in the WPT system.

[0029] Specifically, during the forward power transmission process, the bidirectional charging control terminal is also used to sample the resonant inductor L. f2 Current I s and the drain-source voltage V across switching transistors Q3 and Q4 ds2 (θ), and determine whether the switching transistors Q3 and Q4 are working at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q3 and Q4 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q3 and Q4 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lag conduction angle δ.

[0030] During the reverse energy transfer process, the bidirectional charging control terminal is also used to sample the resonant inductor L. f1 Current I p and the drain-source voltage V across switching transistors Q1 and Q2 ds1 (θ), and determine whether the switching transistors Q1 and Q2 are operating at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q1 and Q2 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q1 and Q2 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q1 and Q2 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q1 and Q2 with the lag conduction angle δ.

[0031] Specifically, during the forward transmission of electrical energy, the conduction angle δ is calculated using the following formula:

[0032]

[0033]

[0034] During the reverse transmission of electrical energy, the conduction angle δ is calculated according to the following formula:

[0035]

[0036]

[0037] θ = ωt represents the original conduction angle.

[0038] This invention provides a single- and two-way WPT system and its control method. Addressing the issue that the efficiency of traditional Class E two-way wireless power transfer systems is easily affected by load, this invention proposes an improved Class E power amplifier topology (referred to as Class E in this example) composed of switches Q1 and Q2 or switches Q3 and Q4. #This topology (type) effectively transfers input current oscillations to ground by resonating the resonant capacitor and resonant inductor connected in parallel with the switching transistor, reducing input current ripple. Furthermore, this topology features zero-voltage turn-on and a wide load range. # The parallel resonant topology of this type of circuit achieves ZVS (Zero-Voltage Switching) over a wide load range, eliminating the need for additional feedback control components and maintaining a constant output voltage amplitude. Compared to traditional Class E inverters that require large choke inductors, the finite input inductance design reduces size, overall weight, and power loss, providing additional freedom in circuit design. Secondly, addressing the issue of reduced overall efficiency caused by changes in coupling coefficients due to coupling device misalignment, leading to hard-switching of the switching transistors, a phase-shifting control scheme is proposed. By controlling the phase of the gate drive signal of the switching transistors, the energy stored inside the resonant components is released either earlier or later, correcting the switching transistors back to the soft-switching (ZVS) state, ensuring the system always operates at its optimal efficiency. Attached Figure Description

[0039] Figure 1 This is a circuit diagram of a unidirectional WPT system provided in an embodiment of the present invention;

[0040] Figure 2 This is provided by the embodiments of the present invention. Figure 1 The circuit shown is in operation mode diagram;

[0041] Figure 3 This is provided by the embodiments of the present invention. Figure 1 The waveform diagram of the circuit's operation is shown below.

[0042] Figure 4 E is provided in the embodiments of the present invention. # Transmitter waveform diagram;

[0043] Figure 5 E is provided in the embodiments of the present invention. # Receiver waveform diagram;

[0044] Figure 6 E is provided in the embodiments of the present invention. # Equivalent model diagram of WPT resonant compensation network;

[0045] Figure 7 The equivalent load R provided in this embodiment of the invention eq Relationship diagram with coupling coefficient k;

[0046] Figure 8 The equivalent load R provided in this embodiment of the invention eq With E # Relationship between drain-source voltage of switching transistor in a type rectifier;

[0047] Figure 9 The bidirectional E provided in the embodiments of the present invention # Topology diagram of a WPT system;

[0048] Figure 10 The bidirectional E provided in the embodiments of the present invention # Overall structure diagram of the WPT system;

[0049] Figure 11 The R provided in the embodiments of the present invention eq >R opt Phase-shift control scheme diagram;

[0050] Figure 12 The R provided in the embodiments of the present invention eq <R opt Phase-shift control scheme diagram;

[0051] Figure 13 E is the simulation provided in the embodiment of the present invention. # Waveform diagram of the transmitter of the inverter;

[0052] Figure 14 E is the simulation provided in the embodiment of the present invention. # Waveform diagram of the receiver end of a type rectifier;

[0053] Figure 15 The bidirectional E in the simulation provided by the embodiments of the present invention # Efficiency, voltage, and load relationship diagram of WPT topology in forward and reverse operation;

[0054] Figure 16 The bidirectional E in the simulation provided by the embodiments of the present invention # A graph showing the relationship between the efficiency, phase shift angle δ, and coupling coefficient k of a WPT topology;

[0055] Figure 17 The bidirectional E in the simulation provided by the embodiments of the present invention # WPT topology efficiency, phase shift angle δ, load R L Relationship diagram;

[0056] Figure 18 E is the experiment provided in the embodiment of the present invention. # Waveforms of the transmitter switching transistors Q1 and Q2 during turn-on and turn-off;

[0057] Figure 19 This is a waveform diagram of the output current at the transmitting end in the experiment provided in the embodiment of the present invention;

[0058] Figure 20 E is the experiment provided in the embodiment of the present invention. # Soft-switching waveforms of drain-source voltages of Q3 and Q4 at the receiving end;

[0059] Figure 21 This is a waveform diagram of phase-shift control with changing coupling coefficient provided in the embodiment of the present invention;

[0060] Figure 22 This is a waveform diagram of load change phase shift control provided in the embodiment of the present invention;

[0061] Figure 23 E is the experiment provided in the embodiment of the present invention. # Input voltage, power, and efficiency relationship diagram for WPT type;

[0062] Figure 24 E is the experiment provided in the embodiment of the present invention. # A graph showing the relationship between coupling coefficient k, power, and efficiency in a WPT (Wide-type Transformer Parametric Pulse) system.

[0063] Figure 25 E is the experiment provided in the embodiment of the present invention. # A diagram showing the relationship between load, power, and efficiency in a WPT (Wastewater Transmission Platform) system. Detailed Implementation

[0064] The embodiments of the present invention are described in detail below with reference to the accompanying drawings. The embodiments are given for illustrative purposes only and should not be construed as limiting the present invention. The accompanying drawings are for reference and illustration only and do not constitute a limitation on the scope of patent protection of the present invention, because many changes can be made to the present invention without departing from the spirit and scope of the present invention.

[0065] Example 1

[0066] This invention provides a unidirectional WPT system, including a transmitter and a receiver, and also includes a control terminal.

[0067] like Figure 1 As shown in the circuit diagram, the transmitter includes a DC power supply (V) connected in sequence. in Output current I in The inverter circuit consists of an inverter circuit and a transmitter resonant circuit. The inverter circuit includes a switching transistor Q1 (whose drain-source voltage is expressed as V). ds1 The current is represented as I ds1 The transmitter resonant circuit includes a resonant inductor L. f1 (its voltage is expressed as V) Lf1 Its current is I. in ), transmitting coil L t1 (Its current is expressed as I) lt1 Voltage is expressed as V p ), resonant capacitor C f1 (Its current is expressed as I) Cf1 ) and resonant capacitor C t1 (Its current is expressed as I) ct1The source of the switching transistor Q1 is connected to the negative terminal of the DC power supply and the resonant capacitor C. f1 At one end, the drain of the switching transistor Q1 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 one end and transmitting coil L t1 One end, resonant capacitor C t1 The other end and the transmitting coil L t1 The other end is connected to the resonant inductor L. f1 One end, resonant inductor L f1 The other end is connected to the positive terminal of the DC power supply, and the gate of the switching transistor Q1 is connected to the control terminal.

[0068] like Figure 1 As shown in the circuit diagram, the receiving end includes a receiving resonant circuit, a rectifier circuit, and a load circuit (including an output filter capacitor C) connected in sequence. L and load R L Filter capacitor C L The current is I co Load R L The current and voltage are I o and V o The rectifier circuit includes switching transistors Q3 and Q4 (Q3 and Q4 together form E). # Type-type topology, the voltage of which is represented by V ds2 The current is represented as I ds2 The receiver resonant circuit includes a resonant inductor L. f2 (Current is represented as I) s Voltage is expressed as V Lf2 ), receiving coil L t2 (Its current is expressed as I) lt2 Voltage is expressed as V s ), resonant capacitor C f2 (Its current is expressed as I) Cf2 ) and resonant capacitor C t2 (Its current is expressed as I) ct2 The source of the switching transistor Q3 is connected to the resonant capacitor C. t2 One end, receiving coil L t2 One end and resonant capacitor C f2 At one end, the drain of switching transistor Q3 is connected to the drain of switching transistor Q4, and the source of switching transistor Q4 is connected to the resonant capacitor C. f2 The other end, one end of the load circuit, is connected to the resonant inductor L. f2 One end, resonant inductor L f2 The other end is connected to the receiving coil L t2 The other end and the resonant capacitor C f2At the other end, the gates of switching transistors Q3 and Q4 are connected to the control terminal.

[0069] The control terminal is used to control the gate drive signal of the switching transistor Q1 during power transmission. The gate signals of Q3 and Q4 are obtained from the current signal of the receiving terminal and are turned on or off simultaneously.

[0070] Among them, switching transistors Q3 and Q4 form E # Type-E topology. # The type circuit has positive and negative symmetry characteristics, and this E # E-type topology is used in bidirectional wireless power transmission. # The Class E inverter achieves ZVS across a wide load range due to its parallel resonant topology, eliminating the need for additional feedback control components and maintaining a constant output voltage amplitude. Compared to traditional Class E inverters that require large choke inductors, the finite input inductance design reduces size, overall weight, and power loss, providing additional freedom in circuit design.

[0071] The control terminal controls switch Q2 to be in normally open mode, controls the gate drive signal of switch Q1, and the gate signals of Q3 and Q4 are obtained from the current signal at the receiving end and are turned on or off simultaneously. Specifically, it controls the WPT system to be in the following five modes sequentially in each cycle:

[0072] Mode I[t0-t1]: such as Figure 2 (a) and Figure 3 As shown, at time t0, the drive signal for switch Q1 is high, and the resonant inductor L... f1 Current I in transmitting coil L t1 Current I lt1 and resonant capacitor C t1 Current I ct1 The current flows through the switch Q1, and the resonant capacitor C... f1 The circuit is short-circuited by switch Q1; the drive signals for switches Q3 and Q4 are also high at this time, and the receiving coil L... t2 Current I lt2 Resonant capacitor C t2 Current I ct2 and resonant inductance L f2 Current I on s The current flows through switching transistors Q3 and Q4, and the resonant capacitor C f2 It was short-circuited.

[0073] Mode II [t1-t2]: such as Figure 2 (b) and Figure 3 As shown, at time t1, switch Q1 is turned on, and the input current I... in The flow increases positively and reaches its maximum value, flowing through the transmitting coil L.t1 Current I it1 From positive to negative, the resonant capacitance C t1 Current I ct1 From negative to positive; receiving coil L t2 Current I lt2 From negative to positive, the resonant capacitance C t2 The current changes from positive to negative.

[0074] Mode III [t2-t3]: such as Figure 2 (c) and Figure 3 As shown, at time t2, switch Q1 is turned off, and the drain-source voltage across switch Q1 increases. At this time, the transmitting coil L... t1 and resonant capacitor C t1 Resonance begins, I lt1 Positive increase, I ct1 Decrease, transmitting coil L t1 At this time, the receiving coil L is supplied with... t2 Energy is transferred; at time t3, the voltage across switch Q1 reaches its maximum value. When switch Q1 is off, the resonant inductor L... f1 and resonant capacitor C f1 Resonance occurs, current I in The resonant capacitance C decreases linearly. f1 Accumulated charge; transmitting coil L t1 With resonant capacitor C t1 Resonance occurs, I lt1 Decrease in the opposite direction, C t1 Accumulate charge; with switches Q3 and Q4 in the off state, the receiving coil L... t2 With resonant capacitor C t2 Resonance occurs, I lt2 Decrease in the opposite direction, C t2 Accumulate charge.

[0075] Modal IV[t3-t4]: such as Figure 2 (d) and Figure 3 As shown, the resonant capacitor C f1 and resonant capacitor C t1 After the charge on the transistor Q1 is completely discharged, the drain-source voltage V of the switching transistor Q1 will decrease. ds1 The value is zero, and the switching transistor Q1 enters soft-switching mode; the resonant capacitor C... f2 and C t2 After the charge on the transistors is completely released, the drain-source voltage V across the switching transistors Q3 and Q4 will decrease. ds2 When the value is zero, switching transistors Q3 and Q4 enter soft-switching mode.

[0076] Modal V[t4-t5]: such as Figure 2 (e) and Figure 3As shown, at time t4, switches Q1, Q3, and Q4 enter the ZVS state, and the transmitting coil L... t1 Current I on lt1 and resonant inductance L f1 Current I in The diode of switch Q1 provides reverse freewheeling, at which point the voltage across switch Q1 is zero, and the drain-source voltages across switches Q3 and Q4 are also zero. The system waits for the drive signals of switches Q1, Q3, and Q4 to go high again in the next cycle, thus achieving zero-voltage turn-on.

[0077] For ease of analysis, the circuit analysis in this paper is based on the following assumptions:

[0078] 1) θ = ωt, ω = 2πf, f is the switching frequency, the load quality factor Q is high enough, and the output signal of the resonant coil at the transmitting end is an ideal sine wave;

[0079] 2) In practical design, the parallel resonant circuit L t1 -C t1 It is an ideal band-stop filter at the operating frequency f, and it is open-circuited for the fundamental frequency component. Therefore, it satisfies the following relationship:

[0080]

[0081] 3) Assume that the ZVS condition is satisfied when switch Q1 is open:

[0082] V ds1 (2πD)=0 (2)

[0083] D represents the duty cycle of the drive signal for the switching transistor Q1.

[0084] 4) All passive components are ideal devices.

[0085] 5) Input current I in It is continuous and independent of whether the switching transistor is on or off.

[0086] 6) Output current I o It is DC.

[0087] like Figure 1 As shown, according to Kirchhoff's voltage and current laws:

[0088] V in =V Lf1 (θ)+V p (θ)+V ds1 (θ) (3)

[0089] I in =i ds1 (θ)+i Cf1 (θ)=i Ct1(θ)+i Lt1 (θ) (4)

[0090] like Figure 4 The waveform shown for the transmitter indicates that the switching transistor has two states, on and off, within one cycle. In the range [0 ≤ θ < 2πD], the switching transistor Q1 is in the off state, and V can be obtained. ds1 and i ds1 :

[0091]

[0092] i ds1 (θ)=0 (6)

[0093] Combination Figure 1 , Figure 4 The input current I can be obtained from equations (4), (5), and (6). in :

[0094]

[0095] In formula (7) This is the initial phase. During the time interval [0 ≤ θ < 2πD], the voltage across switch Q1 is equal to the current i. Cf1 Integrals:

[0096]

[0097] Under steady-state conditions, the input inductance L f1 The average voltage across the terminals is zero. Additionally, due to the output voltage V... P Since it's a sine wave, the average value over one period is zero. Therefore, the input voltage V... in Equal to the average value of the switching voltage Q1 over one cycle:

[0098]

[0099] Assuming the ZVS condition is satisfied, when V ds1 When equal to 0,

[0100] V ds1 (π)=0 (10)

[0101] From equation (10), the ZVS operating condition of the switching transistor Q1 can be obtained:

[0102]

[0103] The receiver uses E # The topology is as follows: Figure 5 As shown, in practice, high-frequency rectification is used, and the receiving end switch also needs to be designed with ZVS (Zero-Voltage-Side Response). Figure 1 , Figure 5 According to Kirchhoff's current law:

[0104]

[0105] Within the range [0 ≤ θ < 2πD], switches Q3 and Q4 are in the off state, i ds2 When the voltage is 0, the voltage across the switching transistor is equal to:

[0106]

[0107] When V ds2 When the value is 0, the ZVS condition is satisfied:

[0108] V ds2 (π)=0 (14)

[0109] From equation (14), the ZVS operating conditions for Q3 and Q4 can be obtained:

[0110]

[0111] To maintain good soft-switching characteristics, the resonant inductor and capacitor L need to be set according to the load and switching frequency. t1 C t1 and L t2 C t2 The parameters are typically used to set the input resonant frequency f between the resonant inductor and resonant capacitor. in It is 1.3-1.5 times the switching frequency f, that is:

[0112]

[0113] Parallel capacitor C f1 C f2 Represented as:

[0114]

[0115] Resonant inductor L f1 L f2 for:

[0116]

[0117] Resonant capacitor C at the transmitter and receiver t1 C t2 for:

[0118]

[0119] Q is the quality factor, R L For load.

[0120] The resonant inductance L of the transmitting and receiving coils t1L t2 for:

[0121]

[0122] For ease of analysis, E # The WPT circuit is decoupled. Figure 6 For the proposed E # Equivalent model of WPT coupled resonant compensation network, V p V is the input voltage. s For the input voltage, R t1 and R t2 R represents the internal resistance of the transmitting coil and the internal resistance of the receiving coil, respectively. eq Let be the equivalent load resistance. In the decoupled equivalent circuit, all variables are sinusoidal quantities, as can be obtained from Kirchhoff's voltage and current laws:

[0123]

[0124] Solving equation (21) yields:

[0125]

[0126] The equivalent impedances of the receiver and transmitter can be obtained as Z. t1 and Z t2 :

[0127]

[0128]

[0129] When the system is in full resonance, Z t1 and Z t2 For purely resistive behavior, i.e., when the imaginary part is zero, the following relationship holds:

[0130]

[0131] The total impedance of the transmitter is Z. P for:

[0132]

[0133] When the system is in full resonance and its impedance is purely resistive (i.e., the imaginary part is zero), the following relationship holds:

[0134]

[0135] Im(Z P ) indicates the imaginary part.

[0136] The system input power and output power are:

[0137]

[0138] The transmission efficiency of the coupling mechanism is obtained as follows:

[0139]

[0140] Secondary load R L The equivalent reflection impedance on the primary side is expressed by equation (31):

[0141]

[0142] Equivalent load R eq Due to the mutual inductance M between the coils, which is the sum of the parasitic resistance and the secondary side reflection resistance, the equivalent load R is... eq It is proportional to the square of the coupling coefficient k, so as the coupling coefficient increases, the equivalent load R... eq It will also increase, such as Figure 7 As shown:

[0143] like Figure 8 As shown, R opt For optimal resistive load values, since a larger load distance results in a smaller coupling coefficient, increasing the load distance will decrease R when the Class E rectifier parameters strictly achieve ZVS. eq The switching transistor transitions to an underresonant state, leading to an increase in its drain-source voltage. Furthermore, the increased load distance increases the coupling coefficient k, causing R... eq Rapidly increasing the voltage causes the switching transistor to become a hard switch, which greatly increases switching losses, raises the junction temperature of the switching transistor, and affects the stable operation of the system.

[0144] Furthermore, directly changing the load size will also change the equivalent load, leading to E # The rectifier switching transistor enters a hard-turn-on state. To address the above problem, this invention also proposes a control method for a unidirectional WPT system, specifically for E... # A phase-shifting control scheme for the rectifier is used to suppress the impact of equivalent load changes caused by changes in the coupling coefficient on the ZVS of the switching transistor, thereby improving the overall efficiency. Specifically, during power transmission, the control terminal samples the receiving coil L... t2 The system measures the current and drain-source voltages across switches Q3 and Q4, and determines whether switches Q3 and Q4 are operating at their optimal switching margin based on the sampled voltage and current. If so, the phase shift angle of switches Q3 and Q4 remains unchanged. Otherwise, it determines whether the drain-source voltages of switches Q3 and Q4 are in an under-resonance or over-resonance state. If it is in an under-resonance state, the gate drive signal of switches Q3 and Q4 is turned on with a lead conduction angle δ. If it is in an over-resonance state, the gate drive signal of switches Q3 and Q4 is turned on with a lag conduction angle δ to ensure optimal efficiency output.

[0145] The conduction angle δ can be calculated using formulas (31) and (32):

[0146]

[0147]

[0148] i Cf2 (θ) represents the resonant capacitance C. f2 The current (related to parameter θ).

[0149] In practical implementation, the inverter circuit can be symmetrically arranged with the rectifier circuit at the receiving end. That is, the inverter circuit includes switching transistors Q1 and Q2, which are symmetrically designed with respect to switching transistors Q3 and Q4, such as... Figure 1 He Ru Figure 9 As shown, switching transistor Q2 may not be required. During energy transfer, the control terminal only needs to control E. # The inverter switch Q2 is in normally open mode. In normally open mode, the switch Q2 can be equivalent to a wire.

[0150] This invention provides a unidirectional WPT system and its control method. Addressing the issue that the efficiency of traditional Class E bidirectional wireless power transfer systems is easily affected by load, this invention proposes an improved Class E power amplifier topology (referred to as Class E in this example) composed of switches Q1 and Q2 or switches Q3 and Q4. # This topology (type) effectively transfers input current oscillations to ground by resonating the resonant capacitor and resonant inductor connected in parallel with the switching transistor, reducing input current ripple. Furthermore, this topology features zero-voltage turn-on and a wide load range. # The parallel resonant topology of this type of circuit achieves ZVS (Zero-Voltage Switching) over a wide load range, eliminating the need for additional feedback control components and maintaining a constant output voltage amplitude. Compared to traditional Class E inverters that require large choke inductors, the finite input inductance design reduces size, overall weight, and power loss, providing additional freedom in circuit design. Secondly, addressing the issue of reduced overall efficiency caused by changes in coupling coefficients due to coupling device misalignment, leading to hard-switching of the switching transistors, a phase-shifting control scheme is proposed. By controlling the phase of the gate drive signal of the switching transistors, the energy stored inside the resonant components is released either earlier or later, correcting the switching transistors back to the soft-switching (ZVS) state, ensuring the system always operates at its optimal efficiency.

[0151] Example 2

[0152] Based on the WPT system of Embodiment 1, this embodiment provides a bidirectional WPT system, such as... Figure 10 As shown, it includes a transmitter / receiver end and a receiver / transmitter end, as well as a bidirectional charging control end (FPGA).

[0153] The transmitting and receiving ends include a power supply load circuit, an inverter rectifier circuit, and a first resonant circuit connected in sequence. The receiving and transmitting ends include a second resonant circuit, a rectifier inverter circuit, and a load power supply circuit connected in sequence. The second resonant circuit adopts the receiving end resonant circuit shown in Embodiment 1. The first resonant circuit adopts the transmitting end resonant circuit shown in Embodiment 1, and also includes a switching transistor Q2. The drain of the switching transistor Q2 is connected to the drain of the switching transistor Q1, and the source of the switching transistor Q2 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 one end and transmitting coil L t1 At one end, the gates of switching transistors Q1, Q2, Q3, and Q4 are connected to the bidirectional charging control terminal.

[0154] The bidirectional charging control terminal is used to control the switching transistor Q2 to be in normally open mode during forward transmission, control the gate drive signal of the switching transistor Q1, and obtain the gate signals of Q3 and Q4 from the current signal of the receiving end and turn them on or off simultaneously. The power supply load circuit, inverter rectifier circuit, rectifier inverter circuit, and load power supply circuit are respectively switched to the DC power supply, inverter circuit, rectifier circuit, and load circuit as shown in Example 1.

[0155] The bidirectional charging control terminal is used to control the switching transistor Q3 to be in normally open mode during reverse transmission, and to control the gate drive signal of the switching transistor Q4. The gate signals of Q1 and Q2 are obtained from the current signal of the receiving end and are turned on or off simultaneously. The power supply load circuit, inverter rectifier circuit, rectifier inverter circuit, and load power supply circuit are respectively switched to the load circuit, rectifier circuit, inverter circuit, and DC power supply shown in Example 1.

[0156] After the system starts up, it determines whether it is operating in the forward or reverse direction. If it is operating in the forward direction, a high-level drive signal is given to switch Q2 to keep it in the normally open state, controlling switch Q2 to be in the normally open mode. The gate drive signal of switch Q1 is controlled, and the gate signals of Q3 and Q4 are obtained from the current signal at the receiving end and are turned on or off simultaneously. If it is operating in the reverse direction, switch Q3 is put in the normally open mode, and the gate drive signal of switch Q4 is controlled. The gate signals of Q1 and Q2 are obtained from the current signal at the receiving end and are turned on or off simultaneously.

[0157] This embodiment also provides a control method for a bidirectional WPT system, including: during the forward energy transfer process, the bidirectional charging control terminal is also used to sample the resonant inductor L. f2 Current I s and the drain-source voltage V across switching transistors Q3 and Q4 ds2(θ), and determine whether the switching transistors Q3 and Q4 are working at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q3 and Q4 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q3 and Q4 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lag conduction angle δ.

[0158] During the reverse energy transfer process, the bidirectional charging control terminal is also used to sample the resonant inductor L. f1 Current I p and the drain-source voltage V across switching transistors Q1 and Q2 ds1 (θ), and based on the sampled voltage and current, determine whether the switching transistors Q1 and Q2 are operating at their optimal switching margin. If so, maintain the current phase shift angle of the switching transistors Q1 and Q2 unchanged; otherwise, determine whether the drain-source voltages of the switching transistors Q1 and Q2 are in an under-resonance or over-resonance state. If under-resonance, turn on the gate drive signal of the switching transistors Q1 and Q2 with a lead conduction angle δ; if over-resonance, turn on the gate drive signal of the switching transistors Q1 and Q2 with a lag conduction angle δ. The bidirectional charging control terminal samples the secondary current signal and sends it to the comparator to obtain the secondary switching transistor drive signal. It also samples the drain-source voltage signal of the switching transistors to determine whether they are operating at their optimal switching margin. Based on different operating states and soft switching margins, it adjusts the E of the device. # The phase shift angle of the rectifier switching transistors ensures optimal efficiency output.

[0159] During the forward transmission of electrical energy, the formula for calculating the conduction angle δ is the same as that in equations (31) and (32).

[0160] Because it is a symmetrical design, the conduction angle δ during the reverse transmission of electrical energy is calculated according to the following formula:

[0161]

[0162]

[0163] The parameter design, operating modes, and phase shift control of the bidirectional WPT system are the same as those of the unidirectional WPT system. The forward and reverse transmissions are symmetrically designed, which will not be elaborated in this embodiment.

[0164] In practical design, the soft-switching margin is designed to be 0.8. If the coupling distance is reduced at this point, allowing R... eq Greater than the optimal value R opt The switching transistor enters a hard-turn-on state. For example... Figure 11The gate drive signal of the switching transistor is turned on with a lag conduction angle δ, which extends the discharge time of the capacitors at both ends of the switching transistor. This ensures that the energy stored in the parallel capacitors at both ends of the switching transistor can be released completely within the switching transistor's off time, thus enabling the switching transistor to enter the ZVS state.

[0165] like Figure 12 As shown, at this point, increasing the coupling distance allows R to... eq Less than the optimal value R opt V across the switching transistor ds The peak value will increase. At this time, the switching transistor is in an under-resonant state. The gate drive signal of the switching transistor is turned on ahead of the conduction angle δ, which reduces the charging time of the capacitors at both ends of the switching transistor. When the switching transistor releases energy, it can release it more quickly, thus reducing the peak value of the voltage across the switching transistor.

[0166] In designing E # When designing inverters and rectifiers, a reasonable ZVS voltage range can be used, and the phase of the drive gate signal of the secondary rectifier can be adjusted by combining formulas (31) and (32).

[0167] If the drain-source voltage V of the secondary rectifier-side switching transistor ds If the circuit is in an underresonant state, the gate drive signal is turned on earlier to reduce the drain-source voltage V of the switching transistor. ds This value protects the switching transistor from breakdown. If the switching transistor is in an over-resonance, i.e., hard-turn-on state, the gate drive signal is delayed to turn it on, causing the switching transistor to enter a soft-switching state, reducing switching losses and improving efficiency. When the coupling coefficient or load changes, this phase-shift control scheme can improve EL. # The ability of WPT type to resist load sensitivity or changes in coupling coefficient.

[0168] The simulation and experimental results will be verified below.

[0169] A bidirectional E-channel was built using MATLAB / Simulink software. # The WPT simulation model is shown in Table 1.

[0170] Table 1 System Simulation Parameters

[0171]

[0172] like Figure 13 As shown, the E proposed in the embodiments of the present invention # In actual operation, the inductor at the input end of the WPT transmitter will resonate with the capacitor connected in parallel across the switching transistor, transferring the current oscillation at the system input end to ground. While achieving ZVS of the switching transistor, it effectively reduces the ripple of the input current, reduces reactive power loss, and improves efficiency.

[0173] Figure 14It is the proposed bidirectional E # The waveform diagram of the WPT receiver shows that when the drive signal of the switching transistor is low, the E signal at the receiver is... # A rectifier-like circuit is used to achieve ZVS for the switching transistor.

[0174] In the simulation, under ideal conditions of input voltage 10-50V, load 5-30Ω, and coupling coefficient k of 0.3, such as Figure 15 As shown, the efficiency gradually increases with increasing input voltage, but gradually decreases with increasing load, as... Figure 15 As shown in (a), when operating in positive mode, with an input voltage of 50V and a load R L At a resistance of 10Ω, the highest efficiency is 85.3%. For example... Figure 15 As shown in (b), when operating in reverse mode, with an input voltage of 50V and a load RL of 10Ω, the highest efficiency is 84.1%. This demonstrates that the proposed bidirectional E# type WPT can achieve ZVS of the switching transistor and high-efficiency transmission when operating in both forward and reverse modes.

[0175] like Figure 16 As shown, when the load resistance is designed to be 10Ω, changing the coupling coefficient k and the distance of the simulated coupling mechanism, the initial optimal coupling coefficient is 0.3. When the coupling coefficient k increases, the distance of the coupling mechanism decreases, the equivalent load increases, and the switching transistors enter a hard-turn-on state. The delayed turn-on of switching transistors Q3 and Q4 causes the parallel capacitor C to... f2 After discharge, the switching transistor enters a soft-switching state; when the coupling coefficient k decreases, the distance of the coupling mechanism increases, the equivalent load increases or decreases, and the switching transistor enters an underresonant state, causing the switching transistors Q3 and Q4 to turn on ahead of time, thus affecting the parallel capacitor C. f2 The charging time is reduced, which reduces the drain-source voltage of the switching transistors Q3 and Q4 when they are off.

[0176] like Figure 17 As shown, with an input voltage of 20V, the bidirectional E under phase-shift control conditions is verified by changing the load size. # The wide load range characteristic of the WPT increases the load resistance R. L The equivalent load increases, and the switching transistors enter a hard-turn-on state. The delayed turn-on of switching transistors Q3 and Q4 causes the parallel capacitor C to... f2 After discharge, the switching transistor enters a soft-switching state; reduce the load resistance R. L The equivalent load decreases, the switching transistors enter an underresonant state, and the pre-turn-on of switching transistors Q3 and Q4 causes the parallel capacitor C to... f2 The charging time is reduced, which reduces the drain-source voltage of the switching transistors Q3 and Q4 when they are off. In summary, by controlling E... #The phase shift angle δ of the rectifier switch can be adjusted to allow the switch to operate in a designed low drain-source voltage and soft-switching mode, thereby reducing switching losses and improving overall efficiency.

[0177] To fully verify the feasibility of the system proposed in this example, an experimental platform was built. The driver chip was a UCC27524, and Q1-Q4 were silicon carbide MOSFETs: NTBG060N065SC1. The coupling coils were wound with 0.1*150 strands of Litz wire. Both the transmitting and receiving coils were circular coils with an inner diameter of 60mm and 10 turns. Nine coils were laid flat behind the circular coils to improve the coupling coefficient and shield leakage flux. The ferrite core dimensions of the coupling mechanism were all 300mm×300mm×10mm. The distance between the transmitting and receiving coils was 50mm. Table 2 shows the actual test parameters.

[0178] Table 2 Bidirectional E # WPT experimental parameters

[0179]

[0180] Figure 18 For bidirectional E # The soft-switching waveform of the transmitter switch of the WPT type. When the drive signal V of the switch Q1... gs Before the voltage level changes from low at time t1 to high at time t2, the drain-source voltage V across switching transistors Q1 and Q2 is... ds1 The voltage has dropped to zero, indicating that the switching transistor has achieved zero-voltage turn-on.

[0181] Figure 19 The waveforms are the soft-switching waveform of the transmitter switch Q1 and the AC current waveform of the transmitter output. The effective value of the current is 1.51A and the frequency is 500KHz. This shows that the designed inverter circuit can stably output a sinusoidal AC signal under the set value in actual operation, and there are no phenomena such as discontinuity or distortion.

[0182] Figure 20 For the receiving end E # The zero-voltage turn-on waveform of the rectifier, before the drive signals of switching transistors Q3 and Q4 change from low to high, E # The drain-source voltage V of the rectifier's switching transistor ds2 The voltage dropping to zero indicates that zero-voltage turn-on has been achieved.

[0183] Figure 21 This is a waveform diagram of phase-shift control under varying coupling coefficients. Figure 21 As shown in (a), the coupling coefficient is relatively small. In practice, this is caused by the offset of the coupling mechanism or the large distance between the coupling devices. In this case, the peak value of the drain-source voltage across the switching transistor is relatively large, and the drive signal is turned on 6° ahead of the actual switching. Figure 21As shown in (b), the peak value of the drain-source voltage across the switching transistor is reduced. Figure 21 (c) shows a large coupling coefficient, with the secondary switch drain-source voltage and drive signal. At this point, it enters a hard-switching state, delaying the drive signal by 9° to turn it on. Figure 21 As shown in (d), the drain-source voltage of the switching transistor is corrected back to the soft-switching state.

[0184] Figure 22 This is a waveform diagram of phase-shift control under load changes. (Example:) Figure 22 As shown in (a), when the load is 5Ω, the equivalent load becomes smaller. At this time, the peak value of the drain-source voltage across the switching transistor is larger, causing the drive signal to be turned on 5.4° ahead of the actual load. Figure 22 As shown in (b), the peak value of the drain-source voltage across the switching transistor is reduced by phase shifting. Figure 22 As shown in (c), when the load is 25Ω, the equivalent load increases, and the system enters a hard-switching state, delaying the drive signal by 6.8° to turn it on. Figure 22 As shown in (d), the drain-source voltage of the switching transistor is corrected back to the soft-switching state.

[0185] like Figure 23 As shown, keeping the coupling coefficient k at 0.5 and the output load constant at 10Ω, adjusting the amplitude of the input voltage yields a bidirectional Ei. # The efficiency of WPT forward and reverse transmission, such as Figure 23 As shown, the efficiency gradually increases with the input voltage. When the input voltage is 50V, the output power reaches 120W in both forward and reverse operation, with an efficiency of 84.3% in forward operation and 83.9% in reverse operation.

[0186] like Figure 24 As shown, in forward operation, the input voltage is maintained at 20V and the output load is 10Ω. The coupling coefficient k is changed by altering the distance between the receiver and transmitter. Testing showed that at a distance of 50mm, the coupling coefficient k is 0.3. Reducing the distance improves output efficiency but reduces output power. At a distance of 40mm, the coupling coefficient k is 0.4, the output power is 31.2W, and the efficiency is 82.7%. Lagging the turn-on of receiver switches Q3 and Q4 by 9° to correct the drain-source voltage results in an output power of 33.2W and an efficiency of 83.4%. Increasing the distance reduces both output efficiency and output power. At a distance of 60mm, the coupling coefficient k is 0.2, the output power is 30.3W, and the efficiency is 78.5%. Leading the turn-on of receiver switches Q3 and Q4 by 6° to correct the drain-source voltage results in an output power of 31.7W and an efficiency of 79.1%.

[0187] like Figure 25As shown, the input voltage is maintained at 30V and the coupling coefficient k is 0.3 during forward operation. With load values ​​ranging from 5 to 30Ω, the output power is 18.2W and the efficiency is 82.1% when the output load is 5Ω. By leading the turn-on of receiver switches Q3 and Q4 by 3° to correct the drain-source voltage, the output power is 22.9W and the efficiency is 83.4%. When the output load is 30Ω, the output power is 68.5W and the output efficiency is 73.4%. By lagging the turn-on of receiver switches Q3 and Q4 by 21° to correct the drain-source voltage, the output power is 83.1W and the efficiency is 78.4%.

[0188] This embodiment addresses the problem of high load sensitivity in traditional bidirectional E-class WPT systems by proposing a novel bidirectional E-class WPT system. # The WPT system features a wide load range. # The circuit can resonate the input current through the input inductor and the capacitor connected in parallel across the switching transistor, diverting the current oscillation at the system input to ground, reducing the ripple of the system input current, and improving system stability. Furthermore, to address the issues of wide load range and coupling coefficient, a method specifically for E is proposed. # A rectifier phase-shifting control scheme was proposed to enable it to operate within the optimal ZVS range, thereby improving system efficiency. Finally, simulation and experimental verification demonstrated the effectiveness and feasibility of the proposed method, leading to the following key conclusions:

[0189] 1) The proposed E # The type topology is suitable for WPT systems because its parallel resonance characteristics between resonant elements can maintain high-efficiency output over a wide load range and maintain ZVS of the transmitter and receiver switches.

[0190] 2) The proposed solution for E # The phase-shift control scheme for the rectifier can effectively correct the hard turn-on of the switching transistor caused by changes in coupling coefficient or load. This is achieved by controlling E... # The type of rectifier can turn on ahead or behind, correcting the switching transistor back to the ideal soft-switching state, reducing the losses caused by hard switching and improving efficiency.

[0191] 3) The proposed bidirectional E # The WPT system can maintain high efficiency while handling a wide range of loads. At an output power of 120W, the efficiency is 84.3% for forward operation and 83.9% for reverse operation.

[0192] The above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention shall be considered equivalent substitutions and shall be included within the protection scope of the present invention.

Claims

1. A unidirectional WPT system, characterized in that, It includes the transmitting end and the receiving end, as well as the control end; The transmitter includes a DC power supply, an inverter circuit, and a transmitter resonant circuit connected in sequence. The inverter circuit includes a switching transistor Q1, and the transmitter resonant circuit includes a resonant inductor L. f1 Transmitting coil L t1 Resonant capacitor C f1 and resonant capacitor C t1 The source of the switching transistor Q1 is connected to the negative terminal of the DC power supply and the resonant capacitor C. f1 At one end, the drain of the switching transistor Q1 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 one end and transmitting coil L t1 One end, resonant capacitor C t1 The other end and the transmitting coil L t1 The other end is connected to the resonant inductor L. f1 One end, resonant inductor L f1 The other end is connected to the positive terminal of the DC power supply, and the gate of the switching transistor Q1 is connected to the control terminal; The receiving end includes a receiving end resonant circuit, a rectifier circuit, and a load circuit connected in sequence. The rectifier circuit includes switching transistors Q3 and Q4. The receiving end resonant circuit includes a resonant inductor L. f2 Receiver coil L t2 Resonant capacitor C f2 and resonant capacitor C t2 The source of the switching transistor Q3 is connected to the resonant capacitor C. t2 One end, receiving coil L t2 One end and resonant capacitor C f2 At one end, the drain of switching transistor Q3 is connected to the drain of switching transistor Q4, and the source of switching transistor Q4 is connected to the resonant capacitor C. f2 The other end, one end of the load circuit, is connected to the resonant inductor L. f2 One end, resonant inductor L f2 The other end is connected to the receiving coil L t2 The other end and the resonant capacitor C f2 At the other end, the gates of switching transistors Q3 and Q4 are connected to the control terminal; The control terminal is used to control the gate drive signal of switch Q1 during power transmission. The gate signals of switches Q3 and Q4 are obtained from the current signal of the receiving terminal and are turned on or off simultaneously.

2. The unidirectional WPT system according to claim 1, characterized in that, The control terminal controls the gate drive signal of switch Q1. The gate signals of switches Q3 and Q4 are obtained from the current signal of the receiving terminal and are turned on or off simultaneously. The specific operating conditions are as follows: I Ct1 I Lt1 These represent the resonant capacitance C. t1 and transmitting coil L t1 The current, I o I represents the load current in the load circuit. co This represents the current in the filter capacitor of the load circuit, wherein the load and the filter capacitor are connected in parallel. This indicates the initial phase of the current at the receiving end.

3. A unidirectional WPT system according to claim 2, characterized in that, Resonant capacitor C f1 C f2 C t1 C t2 Resonant inductor L f1 L f2 transmitting coil L t1 and receiving coil L t2 The parameters are designed as follows: Among them, f in It is the input resonant frequency, ω represents the system's operating angular frequency, Q is the quality factor, and R... L This is the load resistance.

4. A unidirectional WPT system according to claim 2, characterized in that, The control terminal controls the gate signal of switch Q1, and switches Q3 and Q4 are simultaneously turned on or off. Specifically, it controls the unidirectional WPT system to sequentially enter the following five modes in each cycle: Mode I from time t0 to time t1: At time t0, the drive signal for switch Q1 is high, and the resonant inductor L... f1 Current I in transmitting coil L t1 Current I lt1 and resonant capacitor C t1 Current I ct1 The current flows through the switch Q1, and the resonant capacitor C... f1 The circuit is short-circuited by switch Q1; the drive signals for switches Q3 and Q4 are also high at this time, and the receiving coil L... t2 Current I lt2 Resonant capacitor C t2 Current I ct2 and resonant inductance L f2 Current I on s The current flows through switching transistors Q3 and Q4, and the resonant capacitor C f2 Short-circuited; Mode II from time t1 to time t2: At time t1, switch Q1 is turned on, and the input current I... in The flow increases positively and reaches its maximum value, flowing through the transmitting coil L. t1 Current I it1 From positive to negative, the resonant capacitance C t1 Current I ct1 From negative to positive; receiving coil L t2 Current I lt2 From negative to positive, the resonant capacitance C t2 The current changes from positive to negative; Mode III from time t2 to t3: At time t2, switch Q1 is turned off, and the drain-source voltage across switch Q1 increases. At this time, the transmitting coil L... t1 and resonant capacitor C t1 Resonance begins, I lt1 Positive increase, I ct1 Decrease, transmitting coil L t1 At this time, the receiving coil L is supplied with... t2 Energy is transferred; at time t3, the voltage across switch Q1 reaches its maximum value. When switch Q1 is off, the resonant inductor L... f1 and resonant capacitor C f1 Resonance occurs, current I in The resonant capacitance C decreases linearly. f1 Accumulate charge; transmitting coil L t1 With resonant capacitor C t1 Resonance occurs, I lt1 Decrease in the opposite direction, C t1 Accumulate charge; with switches Q3 and Q4 in the off state, the receiving coil L... t2 With resonant capacitor C t2 Resonance occurs, I lt2 Decrease in the opposite direction, C t2 Accumulated charge; Mode IV from time t3 to time t4: resonant capacitance C f1 and resonant capacitor C t1 After the charge on the transistor Q1 is completely discharged, the drain-source voltage V of the switching transistor Q1 will decrease. ds1 The value is zero, and the switching transistor Q1 enters soft-switching mode; the resonant capacitor C... f2 and C t2 After the charge on the transistors is completely released, the drain-source voltage V across the switching transistors Q3 and Q4 will decrease. ds2 When the value is zero, switching transistors Q3 and Q4 enter soft-switching mode; Mode V from time t4 to t5: At time t4, switches Q1, Q3, and Q4 enter the ZVS state, and the transmitting coil L... t1 Current I on lt1 and resonant inductance L f1 Current I in The diode of switch Q1 provides reverse freewheeling, at which point the voltage across switch Q1 is zero, and the drain-source voltages across switches Q3 and Q4 are also zero. The system waits for the drive signals of switches Q1, Q3, and Q4 to go high again in the next cycle, thus achieving zero-voltage turn-on.

5. The control method for a unidirectional WPT system according to claim 4, characterized in that: During power transmission, the control terminal is also used to sample the resonant inductor L. f2 Current I s and the drain-source voltage V across switching transistors Q3 and Q4 ds2 (θ), and determine whether the switching transistors Q3 and Q4 are operating at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q3 and Q4 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q3 and Q4 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lag conduction angle δ.

6. In the control method of a unidirectional WPT system according to claim 5, the conduction angle δ is calculated according to the following formula: θ=ωt represents the original conduction angle, i Cf2 (θ) represents the resonant capacitance C. f2 The current.

7. A bidirectional WPT system, characterized in that: The device includes a transmitting / receiving end and a receiving / transmitting end, as well as a bidirectional charging control end. The transmitting / receiving end includes a power load circuit, an inverter rectifier circuit, and a first resonant circuit connected in sequence. The receiving / transmitting end includes a second resonant circuit, a rectifier inverter circuit, and a load power supply circuit connected in sequence. The second resonant circuit adopts the receiving end resonant circuit according to any one of claims 1 to 6. The first resonant circuit adopts the transmitting end resonant circuit according to any one of claims 1 to 6, and further includes a switching transistor Q2. The drain of the switching transistor Q2 is connected to the drain of the switching transistor Q1, and the source of the switching transistor Q2 is connected to the resonant capacitor C. f1 The other end, resonant capacitor C t1 One end and transmitting coil L t1 At one end, the gates of switching transistors Q1, Q2, Q3, and Q4 are connected to the bidirectional charging control terminal; The bidirectional charging control terminal is used to control the switching transistor Q2 to be in normally open mode during forward transmission, control the gate drive signal of the switching transistor Q1, and the gate signals of Q3 and Q4 are obtained from the current signal of the receiving terminal and are turned on or off simultaneously. The power supply load circuit, the inverter rectifier circuit, the rectifier inverter circuit, and the load power supply circuit are respectively switched to the DC power supply, inverter circuit, rectifier circuit, and load circuit as described in any one of claims 1 to 6. The bidirectional charging control terminal is used to control the switching transistor Q3 to be in normally open mode during reverse transmission, and to control the gate drive signal of the switching transistor Q4. The gate signals of Q1 and Q2 are obtained from the current signal of the receiving end and are turned on or off simultaneously. The power supply load circuit, the inverter rectifier circuit, the rectifier inverter circuit, and the load power supply circuit are respectively switched to the load circuit, rectifier circuit, inverter circuit, and DC power supply as described in any one of claims 1 to 6. That is, the switching transistors Q4, Q2, and Q1 during reverse transmission are respectively equivalent to Q1, Q3, and Q4 during forward transmission.

8. The control method for a bidirectional WPT system according to claim 7, characterized in that: During the forward transmission of electrical energy, the bidirectional charging control terminal is also used to sample the resonant inductor L. f2 Current I s and the drain-source voltage V across switching transistors Q3 and Q4 ds2 (θ), and determine whether the switching transistors Q3 and Q4 are working at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q3 and Q4 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q3 and Q4 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q3 and Q4 with the lag conduction angle δ. During the reverse energy transfer process, the bidirectional charging control terminal is also used to sample the resonant inductor L. f1 Current I p and the drain-source voltage V across switching transistors Q1 and Q2 ds1 (θ), and determine whether the switching transistors Q1 and Q2 are operating at the optimal switching margin based on the sampled voltage and current. If so, keep the phase shift angle of the current switching transistors Q1 and Q2 unchanged. Otherwise, determine whether the drain-source voltage of the switching transistors Q1 and Q2 is in an under-resonance state or an over-resonance state. If it is in an under-resonance state, turn on the gate drive signal of the switching transistors Q1 and Q2 with the lead conduction angle δ. If it is in an over-resonance state, turn on the gate drive signal of the switching transistors Q1 and Q2 with the lag conduction angle δ.

9. The control method for a bidirectional WPT system according to claim 8, characterized in that: During the forward transmission of electrical energy, the conduction angle δ is calculated according to the following formula: During the reverse transmission of electrical energy, the conduction angle δ is calculated according to the following formula: θ = ωt represents the original conduction angle.

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