Wide-angle beam scanning phased array based on near-field coupling and port self-decoupling and design method

Abstract

The application provides a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling and a design method. The design method is different from traditional methods which need to introduce additional beam widening and decoupling structures, and is aimed at directly obtaining a wide-beam active element directional diagram by skillfully utilizing the near-field coupling effect between elements. Meanwhile, a port self-decoupling technology is introduced into the phased array to obtain a higher port isolation and improve the beam gain at a large scanning angle. Therefore, without any parasitic structure, a typical narrow-beam antenna element can be used to construct a phased array with extremely simple structure and excellent beam scanning performance.

Description

Technical Field

[0001] This invention relates to the field of millimeter-wave wireless communication technology, and in particular to a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling, and its design method. Background Technology

[0002] Due to the ever-increasing demand for high-speed wireless communication, fifth-generation and sixth-generation (5G / 6G) millimeter-wave communication technologies have received unprecedented attention. However, the propagation loss of electromagnetic waves in the millimeter-wave band severely limits the transmission distance of signals. To alleviate this problem, high-gain millimeter-wave phased arrays capable of beam scanning have become the mainstream solution. Furthermore, to achieve low cost and good spatial coverage, phased arrays with simple structures and wide beam scanning ranges are widely used.

[0003] Currently, the main method to broaden the beam scanning range of phased arrays is to design antenna elements with a wide half-power beamwidth (HPBW). Several effective beam broadening techniques have been proposed, which can be broadly classified into four categories based on their implementation schemes. The first category involves loading parasitic structures, such as metal walls or metal vias, which can generate an infinity-shaped pattern, thereby improving radiation near low elevation angles. The second category involves superimposing complementary beams provided by different resonant modes, for example, the TM of a microstrip patch antenna (MPA). 01 / TM 11 The first category includes modes and zero-order resonant modes, as well as the fundamental and higher-order modes of dielectric resonator antennas (DRAs). The third category directly utilizes the wide beam radiation characteristics of the antenna itself, such as tilted dipole antennas, conical slot antennas, magnetic dipoles parallel to electric walls, and electric dipoles parallel to magnetic walls. The last category utilizes beam reconfiguration technology to reconfigure two or more beams in different subspaces.

[0004] Another common approach is to reduce the mutual coupling effects of wide-angle impedance matching (WAIM), which can increase beam gain over large scanning angles, thereby widening the beam scanning range of the phased array. Researchers have proposed numerous decoupling techniques, which, based on their decoupling mechanisms, are mainly divided into direct suppression and indirect cancellation methods. Direct suppression methods can directly weaken the coupled field to achieve low coupling through metal via walls, electromagnetic bandgap structures, polarization conversion isolators, etc.; indirect cancellation methods achieve low coupling by establishing new coupling paths to cancel the original coupling paths, such as using symmetrical slots, periodic loops, and decoupling networks. Both decoupling techniques can improve active impedance matching, but at the cost of increased antenna complexity due to the introduction of additional decoupling structures.

[0005] In summary, traditional design methods for wide-angle beam scanning phased arrays aim to broaden the element pattern or improve active impedance matching. This typically requires additional beam stretching and decoupling structures, which inevitably increases the structural complexity, losses, and fabrication costs of the phased array. Therefore, a new solution is needed. Summary of the Invention

[0006] According to one aspect of the present invention, a design method for a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling is provided. The wide-angle beam scanning phased array includes multiple antenna elements with identical structures, and the spacing between the multiple antenna elements is equal. The design method includes the following steps:

[0007] One antenna element in the phased array is selected as the excitation element, and the remaining antenna elements are connected to a matching load as coupling elements to obtain the active element radiation pattern of the excitation element.

[0008] When the operating mode of the coupling unit is the same as that of the excitation unit, the spacing between the antenna elements of the phased array is adjusted to change the phase and amplitude of the coupling field of the coupling unit, so as to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit.

[0009] When the operating mode of the coupling unit is different from that of the excitation unit, the radiation pattern of the operating mode of the coupling unit is adjusted to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit.

[0010] In the design method provided by the present invention, when the antenna element is a microstrip patch antenna and the operating mode of the coupling element is the same as that of the excitation element, the spacing between the antenna elements of the phased array is adjusted so that the phase of the coupling field symmetrically distributed on both sides of the excitation element is equal, the phase difference between the coupling field and the excitation element is in the range of 120° to 240°, and the amplitude of the coupling field is not 0 at the same time.

[0011] In the design method provided by this invention, the microstrip patch phased array is printed on the upper layer of the PCB board and is fed by multiple coaxial probes through an insert-type microstrip line.

[0012] In the design method provided by this invention, when the antenna element is a dielectric resonator antenna, the operating mode of the coupling unit is different from that of the excitation unit; the excitation unit operates in TE mode. 113 In this mode, the coupling units symmetrically distributed on both sides of the excitation unit operate in TE mode. 112 model.

[0013] In the design method provided by this invention, each dielectric resonator antenna is fed by a microstrip coupled rectangular slot, which is etched on the upper surface of the PCB and excited by a stepped microstrip line printed on the lower surface of the PCB.

[0014] In the design method provided by this invention, the radiation pattern of the coupling mode and the radiation pattern of the excitation mode have complementary characteristics.

[0015] The present invention also provides a wide-angle beam scanning phased array designed according to the design method of wide-angle beam scanning phased array based on near-field coupling and port self-decoupling as described above.

[0016] The present invention has at least the following beneficial effects: The present invention provides a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling and its design method. It utilizes the near-field coupling effect to obtain the radiation pattern of wide-beam active elements, and also adopts self-decoupling technology to provide high port isolation to further improve the beam gain at large scanning angles. Therefore, a simple wide-angle beam scanning phased array can be realized using ordinary narrow-beam antenna elements, and no additional beamwidth or decoupling structure is required. Attached Figure Description

[0017] To more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on the provided drawings without creative effort:

[0018] Figure 1The diagram shown is a schematic of a 1 × 5 E-plane ideal linear array.

[0019] Figure 2 The image shown is in a = b , φ 1= φ Normalized radiation patterns with different amplitudes and phases at time 2. G ( ); where (a) different amplitudes a ( φ 1 = 180°, (b) different coupling phases φ 1 ( a = 0.6).

[0020] Figure 3 The following is an example of when a = b , φ 1 φ At time 2, at different phase differences = φ 1– φ 2 ( a = 0.6, φ Normalized pattern under (1 = 180°) G ( ).

[0021] Figure 4 The following is an example of when a b , φ 1= φ At 2 o'clock, different b / a Proportion( a = 0.6, φ Normalized pattern under (1 = 180°) G ( ).

[0022] Figure 5 The diagram shown is a structural diagram of the 1 × 8 millimeter-wave E-plane MPA phased array proposed in Example 1.

[0023] Figure 6 The figure shows the simulated reflection coefficients and transmission coefficients between adjacent elements of the millimeter-wave E-plane MPA phased array 1-8 proposed in Example 1.

[0024] Figure 7 The figure shows the simulated active reflection coefficients of elements 1-8 in the proposed millimeter-wave E-plane MPA phased array at different scanning angles, where (a) 0°, (b) 27°, and (c) 63°.

[0025] Figure 8 The image shows the simulated scanning performance of the millimeter-wave E-plane MPA phased array proposed in Example 1 at a frequency of 26 GHz, where (a) is the main polarization and (b) is the cross polarization.

[0026] Figure 9 The diagram shows the E-plane AEP and E-plane IEP of cells 5 and 8 at 26 GHz.

[0027] Figure 10 The figure shows the simulated current distribution on the proposed millimeter-wave E-plane MPA phased array at 26 GHz when only element 5 or element 8 is excited, where (a) is element 5 and (b) is element 8.

[0028] Figure 11 The diagram shown is a structural diagram of the 1 × 8 millimeter-wave H-plane DRA phased array proposed in Example 2.

[0029] Figure 12 The figure shows the simulated reflection coefficients of cells 1-4 and the transmission coefficients between adjacent cells in the millimeter-wave H-plane DRA phased array proposed in Example 2.

[0030] Figure 13 The figure shows the simulated active reflection coefficients of elements 1-4 in the proposed millimeter-wave H-plane DRA phased array at different scanning angles, where (a) 0°, (b) 28°, and (c) 61°.

[0031] Figure 14 The simulated scanning performance of the H-plane DRA phased array proposed in Example 2 at 26 GHz is shown, where (a) is the main polarization and (b) is the cross polarization.

[0032] Figure 15 The diagram shows the H-plane AEP of cells 1 and 4, and the H-plane IEP at 26 GHz.

[0033] Figure 16 The figure shows the simulated magnetic field distribution in the proposed millimeter-wave H-plane DRA phased array at 26 GHz when only element 4 or 1 is excited, where (a) element 4 is excited and (b) element 1 is excited.

[0034] Figure 17 The following are shown: (a) the equivalent radiation model of coupling element 3 or 5; (b) the radiation pattern of the H-plane. Detailed Implementation

[0035] To facilitate understanding of the present invention, a more complete description will be given below with reference to the accompanying drawings. Typical embodiments of the invention are shown in the drawings. However, the invention can be implemented in many different forms and is not limited to the embodiments described herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete.

[0036] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention pertains. The terminology used herein in the description of the invention is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention.

[0037] As is well known, the radiation pattern of a phased array is mainly determined by three factors, including the isotropic array factor (IAF), the isolated element pattern (IEP), and the impedance mismatch factor (IMF). For an N-element phased array with equal amplitude and equal element spacing, the IAF can be expressed as (1):

[0038] (1).

[0039] in, k It's the wavelength. d It is the unit spacing. θ It's the scanning angle. It is the phase difference between adjacent units. From (1), it can be deduced that when the phase difference is set to... At that time, AF will θ The maximum value is reached at point 0. N This means that the main beam of the phased array has scanned to θ 0. Furthermore, at any scanning angle θ Under these conditions, the maximum value of AF is always equal to the number of units. N Therefore, IAF does not affect the beam scanning performance of the phased array.

[0040] IEP refers to the radiation pattern of an isolated antenna element. Theoretically, neglecting mutual coupling effects, when the HPBW of the IEP is... θ e At that time, the 3-dB beam scanning range of the corresponding phased array can reach ± θ e / 2. However, in reality, due to antenna coupling, the beam scanning range is more directly related to the active element pattern (AEP) of all elements. Specifically, the wider the HPBW of the AEP, the wider the 3-dB beam scanning range of the phased array. Since a wide HPBW IEP usually results in a wide HPBW AEP, a wide-angle beam scanning phased array can often be achieved by using wide-beam antenna elements.

[0041] On the other hand, poor IMF can negatively impact AEP and consequently degrade the phased array's scanning performance. The effects of IMF can be described by near-field coupling and port coupling. The former can sometimes lead to AEP distortion, amplify gain fluctuations during beam scanning, and even create scanning dead zones; the latter can reduce the active reflection coefficient of all elements, especially at large scanning angles, thereby reducing the phased array's beam gain.

[0042] For the reasons mentioned above, the traditional design approach for wide-angle beam scanning phased arrays aims to broaden the IEP (for wide-beam AEP) or improve the IMF (for achieving WAIM). This typically requires additional beam stretching and decoupling structures, which inevitably increases the structural complexity, losses, and manufacturing costs of the phased array.

[0043] Therefore, existing wide-angle beam scanning phased arrays require additional beam widening and decoupling structures, increasing the structural complexity, loss, and manufacturing cost of the phased array. This invention provides a wide-angle beam scanning phased array and its design method based on near-field coupling and port self-decoupling. It utilizes the near-field coupling effect to obtain a wide beamwidth (AEP) and employs self-decoupling technology to provide high port isolation and WAIM. Therefore, a simple wide-angle beam scanning phased array can be implemented using ordinary narrow-beam antenna elements without requiring any additional beamwidth or decoupling structures.

[0044] This invention provides a design method for a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling. The wide-angle beam scanning phased array comprises multiple antenna elements with identical structures, and the spacing between these antenna elements is equal. The design method includes the following steps:

[0045] Step S1: Select one antenna element in the phased array as the excitation element, and connect the remaining antenna elements to the matching load as the coupling element to obtain the active element radiation pattern of the excitation element.

[0046] Specifically, the wide-beam AEP obtained through near-field coupling is key to achieving wide-angle beam scanning. In near-field coupling, there are typically two scenarios: either the operating mode of the coupling unit is the same as that of the excitation unit, or the operating modes of the coupling unit and the excitation unit are different. These two scenarios will be discussed separately below.

[0047] Step S2: When the working mode of the coupling unit is the same as that of the excitation unit, adjust the spacing between the antenna units to change the phase and amplitude of the coupling field of the coupling unit, so as to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit.

[0048] Specifically, for linear arrays, near-field coupling mainly occurs at a radius of approximately λ The near-field coupling outside the 0 element's central region becomes very weak due to the long coupling path. Therefore, this uses... Figure 1 The ideal 1 × 5E-plane linear array shown (cell spacing 0.5) λ 0) To study the E-plane AEP of the central element in the presence of near-field coupling effects. In this array, only the central element 3 is excited, while all other elements are connected to matched loads. Due to mutual coupling effects, the excitation current J3 on the active element 3 will couple to the adjacent passive elements 2 and 4. Generally, the coupling currents J2 and J4 are in the same direction but opposite to the direction of the excitation current J3, and the amplitudes of J2 and J4 are weaker than that of J3. On the other hand, the coupling currents J2 and J4 will act as new excitation sources, coupling to the adjacent passive elements 1 and 5 respectively, forming coupling currents J1 and J5. Similarly, the coupling currents J1 and J5 are in the same direction but opposite to that of J2 and J4, and the amplitudes of J1 and J5 are weaker than those of J2 and J4. For ease of analysis, it is assumed that the amplitude and phase of J3 are 1° and 0°, respectively, and their information along with that of the other coupling currents is listed in Table 1. Furthermore, it is assumed that the coupling from element 3 to element 2 (including coupling strength and coupling phase) is consistent with the coupling from element 2 to element 1 (including coupling phase and coupling phase). Therefore, when the amplitudes and phases of J2 and J4 are assumed to be... a , φ 1 and b , φ At time 2, the amplitudes and phases of J1 and J5 will be respectively a 2,2 φ 1 and b 2,2 φ 2. Therefore, the E-plane AEP of element 3 can be expressed as (2):

[0049] (2)

[0050] in,f e ( ) represents the E-plane IEP. G ( ) is a factor related to near-field coupling, given by equation (3):

[0051] (3)

[0052] The IEP of the antenna element has been determined. Therefore, it can be inferred that the E-plane AEP only relates to... G ( This relates to [the following study], which will examine three different coupling conditions. G ( ) to conduct research.

[0053] Table 1

[0054] Amplitude and phase of excitation current and coupling current

[0055]

[0056] The first scenario is... a = b , φ 1= φ 2. That is, the coupling strength and phase on both sides of the active unit 3 are the same. In this case, G ( ) can be represented as (4):

[0057] (4)

[0058] Therefore, this formula can be used to obtain different amplitudes. a (Assuming) φ Normalized pattern when 1 = 180° G ( ),like Figure 2 As shown in (a). It can be seen that when a = 0 (no energy is coupled to the passive unit). G ( Since is isotropic and equal to 1, AEP becomes IEP. With... a The increase, G ( The value of ) is closer to the axis of view ( θ = 0°) decreases, while in the direction closer to the equator ( θ = 90°) remains at its maximum, therefore G ( It forms an "∞" shape. When the "∞" shape... G ( Multiply by the lateral IEP of the maximum radiation located in the axial view direction.f e ( When this is done, a more uniform AEP with a wide beam will be obtained. Figure 2 (b) gives the results at different coupling phases φ 1 o'clock G ( Normalized direction plot (assuming) a = 0.6). As shown in the figure, when φ 1 = At 90° and 270°, G ( The maximum values ​​of ) appear at ±30° and ±150°. G ( It presents an "X" shape. With... φ 1. Increase from 90° to 180° G ( The shape gradually changes from an "∞" shape, thus widening the AEP. It's worth mentioning here that in a large... a and φ Within 1 range (e.g.) a [0.3, 0.9] φ 1 [120°, 240°] can all achieve an "∞" shape or a quasi-"∞" shape. G ( This provides high design flexibility for obtaining wide-beam AEP through near-field coupling effect.

[0059] The second scenario is... a = b , φ 1 φ 2. This means that the coupling strength on both sides of element 3 is equal, but the coupling phase is different. Therefore, in this case, G ( This can be represented as:

[0060] (5)

[0061] According to formula (5) Figure 3 Different phase differences are given = φ 1– φ 2 o'clock G ( Normalized direction plot (assuming) a = 0.6, φ 1 = 180°). It can be seen that... G ( )right Very sensitive. With phase difference The increase, G ( The symmetrical "∞"-shaped radiation pattern will become asymmetrical. Specifically, the maximum radiation angle on the left deviates from the end-fire direction and splits into two, while the maximum radiation angle on the right remains in the end-fire direction, but its intensity decreases rapidly. Obviously, according to formula (2), this asymmetrical radiation pattern... G ( This will severely degrade AEP. Therefore, when using near-field coupling effect to widen AEP, it should be ensured that the phase of the coupling fields on both sides of the excitation unit is consistent.

[0062] The third scenario is... a b , φ 1= φ 2. This means that the coupling strength on both sides of element 3 is not equal, but the coupling phase is equal. In this case, G ( It can be given by formula (6):

[0063] (6)

[0064] Figure 4 Different b / a Under the proportion (assuming) a = 0.6, φ Normalized radiation pattern (1 = 180°) G ( It can be seen that when the ratio increases significantly from 0 to 1.5, G ( The near-field coupling strength will change slightly, but will always exhibit an "∞" shaped pattern. This result indicates that regardless of the near-field coupling strength, as long as the coupling phase between the excitation field and the coupling field remains synchronized and around 180°, the near-field coupling will not affect the beam broadening effect of the AEP. Even if the coupling strength on one side is zero ( b / a = 0) can also achieve an "∞" shape. G ( As shown by the solid black line, this means edge units (i.e. Figure 1 The AEP of units 1 and 5 in the model may also be broadened by near-field coupling effect.

[0065] It is worth mentioning that all the above analyses are equally applicable to H-plane phased arrays. Whether in the E-plane or the H-plane, the AEP is obtained by the superposition of radiation generated by the excitation field and the coupling field, among which the coupling field, which is closely related to the near-field coupling effect, plays an indispensable role.

[0066] Therefore, in step S2, when the antenna element is a microstrip patch antenna and the operating mode of the coupling element is the same as that of the excitation element, the spacing between the antenna elements of the phased array is adjusted so that the phase of the coupling field symmetrically distributed on both sides of the excitation element is equal, the phase difference between the coupling field and the excitation element is in the range of 120° to 240°, and the amplitude of the coupling field is not 0 at the same time.

[0067] Step S3: When the working mode of the coupling unit is different from that of the excitation unit, adjust the radiation pattern of the working mode of the coupling unit to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit.

[0068] Specifically, considering that the excitation sources of the excitation unit and the coupling unit are different (specifically, the excitation source of the excitation unit is its feed network, while the excitation source of the coupling unit is the resonant near field of the excitation unit), the operating mode of the coupling unit may differ from the resonant mode of the excitation unit. For example, the excitation DRA operates in TE... 113 The pattern, but its adjacent coupling DRA exhibits TE 112 The field distribution of the mode. In this case, the near-field coupling effect can also amplify the AEP due to the complementary radiation patterns of the excitation mode and the coupling mode.

[0069] Therefore, in step S3, the operating mode of the coupling unit is different from that of the excitation unit; the excitation unit operates in TE mode. 113 In this mode, the coupling units symmetrically distributed on both sides of the excitation unit operate in TE mode. 112 This coupling mode, which cannot be excited through coupling slots, achieves port self-decoupling, improves the active impedance matching of the antenna, and increases beam gain over large scanning angles. The TE of the coupling unit... 112 The radiation pattern of the mode has zero radiation in the axial view direction, and the maximum radiation occurs at ±60° and ±120°. When this radiation pattern is superimposed on the radiation pattern of the excitation element, the gain near ±60° and ±120° is enhanced, while the gain in the axial view direction remains almost unchanged, thereby widening the active element radiation pattern of the excitation element.

[0070] Furthermore, in one embodiment of the present invention, when the antenna element is a microstrip patch antenna, the microstrip patch antenna is printed on the upper layer of the PCB board and fed by the same coaxial probe through an insert-type microstrip line. This insert-type feeding structure can generate a new coupling path to cancel the original coupling of the radiating patch, making the field at the coupling MPA very weak, thereby achieving an extremely low mutual coupling level between adjacent elements, improving the beam gain of the phased array at large scanning angles, without the need for an additional decoupling structure.

[0071] Furthermore, in one embodiment of the present invention, when the antenna element is a dielectric resonator antenna, each dielectric resonator antenna is fed by a microstrip coupled rectangular slot, the microstrip coupled rectangular slot being etched on the upper surface of the PCB and excited by a microstrip line printed on the lower surface of the PCB. The excitation unit operates in TE... 113 In this mode, the coupling units symmetrically distributed on both sides of the excitation unit operate in TE mode. 112 This coupling mode, because it cannot be excited through coupling gaps, achieves an extremely low level of mutual coupling between adjacent units without the need for additional decoupling structures.

[0072] The method of the present invention will be explained in detail below with two specific examples.

[0073] Example 1: 1 × 8 mmWave E-plane MPA Phased Array

[0074] This embodiment designs a 1 × 8 mm wave E-plane wide-angle beam scanning MPA phased array to verify the feasibility of the proposed design method.

[0075] Figure 5 The proposed millimeter-wave E-plane MPA phased array is shown in the diagram. It consists of eight identical MPA elements with a spacing of [missing information]. s e These MPA units are printed on the upper layer of a rectangular PCB with a thickness of 0.508 mm and a dielectric constant of 2.2. Furthermore, all MPA units are excited by a 50 Ω coaxial probe via an insert-type microstrip line. This insert-fed MPA can generate new coupling through the feeding structure to counteract the original coupling of the radiating patch, making the field at the coupled MPA extremely weak, thus achieving a very low mutual coupling level between adjacent units without the need for additional decoupling structures. The MPA phased array designed in this embodiment is suitable for the 26 GHz millimeter-wave band, and its detailed dimensions are shown in Table 2.

[0076] Table 2

[0077] The proposed millimeter-wave E-plane MPA phased array dimensions

[0078]

[0079] Figure 6The simulated reflection coefficients of all elements in the proposed E-plane MPA phased array and the transmission coefficients between adjacent elements are shown. As can be seen from the figure, the reflection coefficients of these eight elements are almost identical, exhibiting good impedance matching and providing a 3.1% overlap -10 dB impedance bandwidth in the 25.6–26.4 GHz range. Furthermore, the transmission coefficients between any two adjacent elements are also very close, all below -28.7 dB at 26 GHz, indicating a significant improvement in port isolation due to self-decoupling effects. Figure 7 The simulated active reflection coefficients of the proposed MPA phased array are shown at different scanning angles of 0°, 27°, and 63°. It can be seen that all antenna elements achieve good active impedance matching around 26 GHz at all three sampling angles, and similar results were observed at other scanning angles. This indicates that the MPA phased array achieves WAIM with the help of port self-decoupling technology.

[0080] Figure 8 The simulated scanning performance of the proposed MPA phased array at 26 GHz is shown. As can be seen from the figure, the main beam can scan from -63° to +63° with a very small gain change of only 1.2 dB, and the peak sidelobe level (SLL) is below -10.5 dB over a wide scanning range. Due to the implementation of WAIM, the beam gain at -63° angle reaches 13.8 dBi, and the maximum beam gain at 0° angle reaches 15.0 dBi. Furthermore, this MPA phased array also exhibits high polarization purity. Throughout the scanning range, the main polarization field is at least 44.5 dB higher than the cross-polarization field.

[0081] In summary, the proposed MPA phased array achieves a wide sweep range exceeding 126° (±63°, and ±68° for 3-dB beam scanning if SLL attenuation is ignored), as well as excellent gain and polarization performance. It is worth noting that these favorable characteristics cannot be attributed solely to the MPA element, as its E-plane HPBW is limited to 96°.

[0082] In fact, the wide beam scanning range is achieved thanks to the near-field coupling effect between elements, which provides the wide beam AEP. To explain this principle in detail, Figure 9The E-plane AEP of the central element 5 and the edge element 8 are shown, along with the E-plane IEP for comparison. It can be seen that the AEP of element 8 is very similar in the upper half-space compared to the IEP, while the AEP beamwidth of element 5 is much wider. Table III lists the specific HPBW of the AEPs of elements 5, 8, and all other elements in the MPA phased array. It can be seen that the HPBW of all AEPs is wider than that of the IEP, especially the HPBW of elements 1-6, which exceeds 140°. Although the AEP beamwidths of edge elements 7 and 8 are only 121° and 99° respectively, they have little impact on the scanning performance of the proposed phased array. Furthermore, from... Figure 9 It can be seen that the cross-polarization levels of AEP and IEP are quite similar, both below -40 dB.

[0083] Table 3

[0084] HPBW of the E-plane IEP and AEP of all elements in the proposed MPA phased array

[0085]

[0086] In addition, the near-field coupling effect of units 5 and 8 was investigated to explain the broadening mechanism of AEP. Figure 10 It shows when only unit 5 ( Figure 10 (a) or Unit 8 ( Figure 10 (b) The simulated current distribution on the proposed E-plane MPA phased array during excitation. As shown in the figure, when only element 5 is excited, the current intensity on the right-side coupled MPA is much greater than that on the left-side MPA. This is due to the inconsistent coupling strength on both sides of the excitation element. Therefore, the AEP of element 5 mainly depends on its excitation current and the coupling current on passive elements 6 and 7. In this case, based on the above analysis, G ( ) can be represented as (7):

[0087] (7)

[0088] It exhibits an "∞" shape, thus providing a wide beamwidth AEP. The difference is that when only the rightmost element 8 is excited, the coupling current on all passive elements is very weak, therefore... G ( The value is very close to 1, having little effect on broadening the AEP, resulting in a narrow-beam AEP almost identical to the IEP. These analyses demonstrate the important role of near-field coupling effects in broadening the AEP and explain why different elements have different AEPs (see...). Figure 9 ).

[0089] Example 2: 1 × 8 mmH-plane DRA phased array

[0090] Figure 11 The structure of the proposed 1 × 8 mm-wave H-plane DRA phased array is shown, which consists of eight sides with a length of... a Gao Wei h The dielectric constant is ε r1 It consists of DRA units with a square base. Each DRA consists of a unit with a size of... l s × w s Microstrip coupled rectangular slot feeding. The slot is etched at a thickness of [thickness value missing]. h s The dielectric constant is ε r2 The upper surface of the square PCB is excited by a 50 Ω microstrip line printed on the lower surface of the PCB. A stepped microstrip line is used to achieve good impedance matching. Furthermore, the proposed DRA phased array design operates in the 26 GHz millimeter-wave band, and its detailed dimensions are shown in Table 4. Notably, in this design, the proposed port self-decoupling technique is achieved by appropriately designing the operating modes (dimensions) of the excitation and coupling DRA units. Specifically, the excitation DRA operates in the TE... 113 The pattern, while coupled DRA exhibits TE 112 The field distribution of the mode has a relatively weak magnetic field strength at the feed slot location, thus achieving extremely low mutual coupling.

[0091] Table 4

[0092] The proposed dimensions of the millimeter-wave H-plane DRA phased array

[0093]

[0094] Figure 12 Simulated reflection coefficients for elements 1-4 and transmission coefficients between adjacent elements in the proposed DRA phased array are presented. Here, due to the array and... xoz The plane (H-plane) is completely symmetrical; for clarity, only the results for elements 1-4 are presented. It can be seen that all four elements are well-matched, with a coincident -10 dB impedance bandwidth of 14.6% over a frequency range of 24.2-28.0 GHz. Furthermore, due to the use of port self-decoupling technology, the isolation between any two adjacent elements exceeds 25.7 dB at 26 GHz. Figure 13 As shown, the DRA unit also achieved good active impedance matching at different scanning angles of 0°, 28° and 61°.

[0095] Figure 14The scanning performance of the proposed DRA phased array at 26 GHz is presented. As shown in the figure, the main beam can scan from -61° to +61°. Over the wide scanning range, the gain variation is only 1.6 dB, with a peak SLL of -9.1 dB. Similarly, due to the implementation of WAIM using self-decoupling technology, the maximum beam gain at 0° reaches 15.8 dBi, and the beam gain at the large angle of 61° reaches 14.2 dBi. Detailed information on the beam scanning performance at 25, 26, and 27 GHz is provided in Table 5. It is worth noting that we also investigated other frequencies, but for brevity, these are not detailed here. The results show that the proposed DRA phased array achieves a wide ±60° beam scanning bandwidth of up to 12.7% (with a 3-dB beam scanning range exceeding ±60° across the entire band) from 24.2 to 27.5 GHz.

[0096] Table 5

[0097] The proposed millimeter-wave H-plane DRA phased array scanning performance at 25, 26, and 27 GHz

[0098]

[0099] Similar to the E-plane MPA phased array proposed in Example 1, the wide-angle beam scanning capability of the H-plane DRA phased array under near-field coupling also benefits from the broadened AEP. Figure 15 The H-plane AEP and H-plane IEP of elements 1 and 4 at 26 GHz are shown. It can be seen that the AEP beamwidth of elements 1 and 4 is significantly wider than the IEP beamwidth. For a clear comparison, Table 6 lists specific information on the HPBW of the H-plane IEP and AEP. It can be observed that at 26 GHz, the HPBW of the AEP of elements 1-4 is more than 30° wider than the IEP. Similar results were observed at 25 and 27 GHz. Notably, at 27 GHz, although the AEP of element 1 is only 95°, it is still 24° wider than the IEP. Furthermore, the relatively narrow AEP of the edge elements has no substantial impact on the beam scanning performance of the DRA phased array, as other elements with wide-beam AEP can compensate for this deficiency.

[0100] Table 6

[0101] At 25, 26, and 27 GHz, the proposed H-plane IEP and AEP HPBW of 1-4 elements in the H-plane DRA phased array

[0102]

[0103] In the DRA phased array, from Figure 15As can be seen from Table 6, the AEP beamwidth of the central element (element 4) is wider than that of the edge element (element 1). Similar to the MPA array, this is also due to the difference in their near-field coupling environment (position). Figure 16 Only excitation unit 4 is shown. Figure 16 (a) or Unit 1 ( Figure 16 (b) Simulated magnetic field distribution in the DRA array. As shown in the figure, the operating mode of the excitation unit (unit 4 or unit 1) is different from that of the coupling unit, namely TE. 113 ( x ) and TE 112 ( x ( ) mode. According to mirror theory, the radiation pattern of passive element 3 or 5 can be considered as being generated by a radiation pattern located 0.25 Å above the ground plane. λ 0 and below 0.25 λ The radiation pattern generated by two magnetic currents in the same direction at 0. Therefore, its H-plane radiation pattern can be calculated from (8).

[0104] |cos |(8)

[0105] Figure 17 (a) and (b) show the equivalent radiation models and H-plane radiation patterns of coupling elements 3 and 5, respectively. Figure 17 As can be seen in (b), a radiation null point appears in the axial view direction, and the maximum radiation occurs at ±60° and ±120°. When this radiation pattern is superimposed on the IEP of excitation element 4, the gain near ±60° and ±120° is enhanced, while the gain in the axial view direction remains almost unchanged, thus making the AEP more uniform. This also explains why the AEP of element 4 has a relatively high gain near ±120° (see...). Figure 15 ).like Figure 16 As shown in (b), when only cell 1 is excited, a + will also form in passive cell 2. x Similar magnetic currents in directions M Therefore, the AEP of element 1 will also be widened. However, unlike element 4, since there are no passive elements on the left side of element 1, the AEP widening effect of element 1 is slightly lower, resulting in a relatively narrower HPBW. These results show that even if the coupling mode and the excitation mode are different, a wide-beam AEP can still be achieved by superimposing their complementary radiation patterns.

[0106] This invention proposes a novel wide-angle beam scanning phased array design method, which cleverly utilizes the near-field coupling effect between antenna elements to achieve wide-beam AEP (Aspect-Effect Coefficient), while reducing port coupling through self-decoupling technology to obtain WAIM (Wide-in-the-moment). Based on this, an E-plane MPA (Multi-Purpose Aspect-of-Earth) phased array and an H-plane DRA (Digital-Range Aspect-of-Earth) phased array were designed. Results show that both phased arrays achieve excellent beam scanning performance without any parasitic structures or active control circuits. The MPA phased array can scan from -64° to +64° with a gain ripple as low as 1.0 dB, while the DRA phased array can scan from -60° to +60° with a gain ripple of less than 3 dB over a 2.7% wideband. Furthermore, the MPA and DRA phased arrays also achieve high beam gains of 14.5 dBi and 15.8 dBi, respectively, and low SLL (Short-Range Gain) of -8.3 dB and -9.1 dB, respectively. This novel design method, in addition to its significant advantage of structural simplicity, also possesses good versatility. This technology can be used not only to realize one-dimensional wide-angle scanning arrays, but also potentially two-dimensional and broadband wide-angle scanning arrays. Finally, it is important to emphasize that in our design, field coupling and port coupling between antenna elements are analyzed and handled separately, utilizing the former while suppressing the latter. This opens a new window for future research into self-decoupling techniques and wide-angle scanning phased arrays.

[0107] The above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention shall be considered equivalent substitutions and shall be included within the protection scope of the present invention.

Claims

1. A design method for a wide-angle beam scanning phased array based on near-field coupling and port self-decoupling, characterized in that, The wide-angle beam scanning phased array includes multiple antenna elements with identical structures, and the spacing between the multiple antenna elements is equal. The design method includes the following steps: One antenna element in the phased array is selected as the excitation element, and the remaining antenna elements are connected to a matching load as coupling elements to obtain the active element radiation pattern of the excitation element. When the operating mode of the coupling unit is the same as that of the excitation unit, the spacing between the antenna elements of the phased array is adjusted to change the phase and amplitude of the coupling field of the coupling unit, so as to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit. When the operating mode of the coupling unit is different from that of the excitation unit, the radiation pattern of the operating mode of the coupling unit is adjusted to broaden the active element radiation pattern of the excitation unit based on the near-field coupling effect between the coupling unit and the excitation unit.

2. The design method as described in claim 1, characterized in that, When the antenna element is a microstrip patch antenna and the operating mode of the coupling element is the same as that of the excitation element, the spacing between the antenna elements of the phased array is adjusted so that the phase of the coupling field symmetrically distributed on both sides of the excitation element is equal, the phase difference between the coupling field and the excitation element is in the range of 120° to 240°, and the amplitude of the coupling field is not 0 at the same time.

3. The design method as described in claim 2, characterized in that, The microstrip patch antenna is printed on the upper layer of the PCB board and is fed by multiple coaxial probes through an insertable microstrip line.

4. The design method as described in claim 1, characterized in that, When the antenna element is a dielectric resonator antenna, and the operating mode of the coupling element is different from that of the excitation element, the excitation element operates in TE mode. 113 In this mode, the coupling units symmetrically distributed on both sides of the excitation unit operate in TE mode. 112 model.

5. The design method as described in claim 4, characterized in that, Each dielectric resonator antenna is fed by a microstrip-coupled rectangular slot etched on the upper surface of the PCB and excited by stepped microstrip lines printed on the lower surface of the PCB.

6. The design method as described in claim 4, characterized in that, The radiation patterns of coupled modes and excited modes are complementary.

7. A wide-angle beam scanning phased array designed according to the design method of wide-angle beam scanning phased array based on near-field coupling and port self-decoupling as described in any one of claims 1 to 6.

Images