Brushless doubly-fed machine full-order terminal sliding mode control method under unbalanced load

Abstract

The application belongs to the field of motor control, and particularly relates to a full-order terminal sliding mode control method for a brushless doubly-fed motor. The application aims to solve the problem that when the existing brushless doubly-fed motor is connected with an unbalanced load, negative sequence unbalanced voltage and current appear on the power winding side, torque pulsation is caused, and the motor is internally heated and even catches fire. The process is as follows: obtaining a non-matching power winding output voltage amplitude outer loop subsystem and a matching control winding output current inner loop subsystem in the positive sequence and negative sequence dq coordinate system considering motor parameter perturbation; for the non-matching power winding output voltage amplitude outer loop subsystem and the matching control winding output current inner loop subsystem considering motor parameter perturbation, a full-order terminal sliding mode controller is designed, so that the power winding side positive sequence or negative sequence voltage is stably tracked to a given value, and the control winding positive sequence or negative sequence current is stably tracked to a given value.

Description

Technical Field

[0001] This invention belongs to the field of motor control, specifically relating to a full-order terminal sliding mode control method for brushless doubly fed motors. Background Technology

[0002] With the rapid development of my country's economy, energy demand has grown rapidly, while fossil fuels are gradually being depleted. Therefore, the government has introduced a series of policies to encourage the development of wind power, photovoltaic power, and hydropower. Brushless doubly-fed motors are used in wind power and marine power generation. Compared to traditional doubly-fed motors, they eliminate the need for brushes and slip rings, saving costs while improving operational reliability.

[0003] Brushless doubly-fed induction generators (DFIGs) can be connected to the grid for power generation or operate as an independent power generation system connected to a load. When a DFIG is connected to an unbalanced load, negative sequence voltage and current will appear in the system, causing voltage imbalance on the power winding side. This leads to large current and torque pulsation in the motor stator winding, potentially causing high temperatures or even fires inside the motor. To address these issues, some researchers have proposed using negative sequence voltage compensators to reduce the impact of unbalanced loads on the power winding voltage. However, this adds an additional loop, increasing system cost and complexity. Other researchers have proposed using filters to extract 5th and 7th harmonics, but this increases the proportional or proportional-integral (PI) stage of the system, further increasing system complexity.

[0004] Sliding mode control has advantages such as simple structure, strong robustness, and fast response speed, and is widely used in electric vehicles, robotics, and aerospace. However, due to the switching function in the control law, sliding mode control can lead to chattering, which reduces the control effect of the system, increases the performance requirements of the controller, and reduces the robustness of the system. Summary of the Invention

[0005] The purpose of this invention is to solve the problem that when an existing brushless doubly fed motor is connected to an unbalanced load, negative sequence unbalanced voltage and current appears on the power winding side, causing torque pulsation, internal heating of the motor, and even fire. Therefore, a full-order terminal sliding mode control method for brushless doubly fed motors under unbalanced loads is proposed.

[0006] The specific process of the full-order terminal sliding mode control method for brushless doubly-fed motors under unbalanced load is as follows:

[0007] Step 1: Collect the voltage and current on the two stator winding sides of the brushless doubly fed motor to obtain the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system. Perform coordinate transformation on the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to obtain the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system.

[0008] Step 2: Decompose the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system into positive and negative order to obtain the dynamic mathematical model in the positive order dq coordinate system and the dynamic mathematical model in the negative order dq coordinate system.

[0009] Consider motor parameter perturbations;

[0010] Based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation.

[0011] Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

[0012] Step 3: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the positive sequence dq coordinate system considering the perturbation of motor parameters, design a full-order terminal sliding mode controller to make the positive sequence voltage amplitude on the power winding side stably track the given value;

[0013] For the inner loop subsystem of the output current of the matched control winding considering motor parameter perturbation in the positive sequence dq coordinate system, a full-order terminal sliding mode controller is designed to make the positive sequence current of the control winding stably track the given value.

[0014] For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the negative sequence dq coordinate system considering motor parameter perturbation, a full-order terminal sliding mode controller is designed to make the negative sequence voltage on the power winding side stably track the given value 0.

[0015] For the inner loop subsystem of the matching control winding output current considering motor parameter perturbation in the negative sequence dq coordinate system, a full-order terminal sliding mode controller is designed to ensure that the negative sequence current of the control winding stably tracks the given value 0.

[0016] The beneficial effects of this invention are as follows:

[0017] This invention proposes a full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load. First, a mathematical model of the brushless doubly-fed motor in a three-phase stationary coordinate system is established. After coordinate transformation and positive-negative sequence separation, a mathematical model of the brushless doubly-fed motor in a positive-negative sequence dq coordinate system is obtained. Feedforward decoupling compensation is performed, and the unmatched uncertainty caused by system parameter perturbation is fully considered. An outer loop subsystem of the unmatched power winding output voltage amplitude and an inner loop subsystem of the matched control winding output current are established in the positive-negative sequence coordinate system. Then, full-order terminal sliding mode controllers are designed for the dual closed-loop subsystems in the positive-negative sequence coordinate system, and the positive and negative sequence components are independently adjusted to suppress the negative sequence component. Finally, the three-phase imbalance of output current and power is reduced, current ripple is reduced, and system robustness is improved.

[0018] This invention discloses a full-order terminal sliding mode control method for a brushless doubly fed motor under unbalanced load. First, a dynamic model of the brushless doubly fed motor in the dq coordinate system is established, fully considering the uncertainty caused by the perturbation of motor parameters due to temperature and frequency changes. Then, an outer loop subsystem for the output voltage amplitude of the unmatched power winding and an inner loop subsystem for the decoupling of the matching control winding current are constructed in the positive and negative sequence coordinate systems.

[0019] This invention designs full-order terminal sliding mode controllers for the inner and outer loop subsystems respectively, and independently adjusts the positive-sequence and negative-sequence components in the system to suppress the negative-sequence component in the output voltage.

[0020] The method proposed in this invention, based on the compensation of matching and unmatching uncertainties in the system, can drive the tracking error to converge quickly and has a smooth control signal, thereby improving the dynamic performance, control accuracy and robustness to parameter perturbations of the BDFIG independent power generation system under unbalanced load, and optimizing the three-phase balance of the output voltage. Attached Figure Description

[0021] Figure 1 This is a flowchart of the method of the present invention;

[0022] Figure 2 This is a topology diagram of BDBFIG, where BDBFIG represents a brushless doubly-fed generator, ω r denoted as the rotor angular velocity in positive sequence coordinates, and p1 and p2 are the number of pole pairs of the power winding and the control winding, respectively. Detailed Implementation

[0023] Specific Implementation Method 1: The specific process of the full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load in this implementation method is as follows:

[0024] Step 1: Collect the voltage and current on the two stator winding sides of the brushless doubly fed motor to obtain the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system. Perform coordinate transformation on the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to obtain the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system.

[0025] Step 2: Decompose the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system into positive and negative order to obtain the dynamic mathematical model in the positive order dq coordinate system and the dynamic mathematical model in the negative order dq coordinate system.

[0026] Consider the perturbation of motor parameters caused by changes in operating conditions during actual motor operation;

[0027] Based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation.

[0028] Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

[0029] Step 3: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the positive sequence dq coordinate system considering the perturbation of motor parameters, design a full-order terminal sliding mode controller so that the positive sequence voltage amplitude on the power winding side can quickly and stably track the given value.

[0030] For the inner loop subsystem of the matching control winding output current considering motor parameter perturbation in the positive sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the positive sequence current of the control winding to quickly and stably track the given value.

[0031] For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the negative sequence dq coordinate system considering motor parameter perturbation, a full-order terminal sliding mode controller is designed to enable the negative sequence voltage on the power winding side to quickly and stably track the given value of 0.

[0032] For the inner loop subsystem of the output current of the matching control winding considering motor parameter perturbation in the negative sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the negative sequence current of the control winding to quickly and stably track the given value of 0.

[0033] Specific Implementation Method Two: This implementation method differs from Specific Implementation Method One in that: in step one, the voltage and current on the two stator winding sides of the brushless doubly fed motor are collected to obtain the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system. The dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system is then transformed to obtain the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system.

[0034] The specific process is as follows:

[0035] Step 1: Establish a dynamic mathematical model of the brushless doubly fed motor in a three-phase stationary coordinate system;

[0036] Step 1 and Step 2: Using Clark and Park coordinate transformation, transform the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to the dynamic mathematical model in the two-phase rotating dq coordinate system.

[0037] In step one by one, the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system includes voltage equation, flux linkage equation and electromagnetic torque equation.

[0038] Voltage equation:

[0039]

[0040]

[0041]

[0042] In the formula: u1 is the voltage vector on the power winding side, u2 is the voltage vector on the control winding side, u r The rotor-side voltage vector;

[0043] i1 is the power winding side current vector, i2 is the control winding side current vector, i r The rotor current vector;

[0044] ψ1 is the flux linkage on the power winding side, ψ2 is the flux linkage on the control winding side, ψ r For rotor flux linkage;

[0045] R1 is the resistance on the power winding side, R2 is the resistance on the control winding side, R r The resistance of the rotor winding;

[0046] t represents time, and d / dt is the derivative operation with respect to time t;

[0047] Magnetic flux linkage equation:

[0048] ψ1=M1i1+M 1r i r

[0049] ψ2=M2i2+M 2r i r

[0050]

[0051] Where: M1, M2 and M r These are the self-inductance matrices for the power winding, control winding, and flux linkage winding, respectively.

[0052] M 1r This represents the mutual inductance matrix of the power winding and the rotor winding;

[0053] M 2r To control the mutual inductance matrix of the rotor winding and the control winding;

[0054] For M 1r The transpose of a matrix. For matrix M 2r The transpose of the matrix;

[0055]

[0056]

[0057]

[0058]

[0059]

[0060] In the formula: L σ1 For the single-phase leakage inductance on the power winding side, L σ2 To control single-phase leakage inductance on the winding side;

[0061] L m1 L is the single-phase magnetizing inductor on the power winding side. m2 To control the single-phase magnetizing inductance on the winding side; L mr L is the single-phase magnetizing inductance of the rotor winding. σr For single-phase leakage inductance of the rotor winding;

[0062] L pr L is the mutual inductance amplitude between the power winding side and the rotor winding. cr To control the mutual inductance amplitude between the rotor winding and the rotor winding;

[0063] θ0 is the initial position difference between the A-phase axis on the power winding side and the A-phase axis on the control winding side;

[0064] θ r The rotor position angle;

[0065] p1 and p2 are the number of pole pairs of the power winding and the number of pole pairs of the control winding, respectively;

[0066] Electromagnetic torque equation:

[0067]

[0068] In the formula: T e Electromagnetic torque;

[0069] The superscript T indicates transpose;

[0070] This is the transpose of the power winding side current vector matrix;

[0071] d / dθ r For the rotor position angle θ r Differentiation operation.

[0072] The other steps and parameters are the same as in Specific Implementation Method 1.

[0073] Specific Implementation Method Three: This implementation method differs from Specific Implementation Method One or Two in that, in steps one and two, Clark and Park coordinate transformations (Clark coordinate transformation first, then Park coordinate transformation) are used to transform the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to the dynamic mathematical model in the two-phase rotating dq coordinate system; the specific process is as follows:

[0074] The dynamic mathematical model of a brushless doubly fed motor in a two-phase rotating dq coordinate system includes voltage equations, flux linkage equations, and electromagnetic torque equations.

[0075] Voltage equation:

[0076]

[0077]

[0078]

[0079] In the formula: u 1d and u 1q These are the d-axis and q-axis voltages of the power winding, respectively.

[0080] u 2d and u 2q These are the d-axis and q-axis voltages of the control winding, respectively;

[0081] u rd and u rq These are the d-axis and q-axis voltages of the rotor winding, respectively.

[0082] i 1d and i 1q These are the d-axis and q-axis currents of the power winding, respectively.

[0083] i 2d and i 2q These are the d-axis and q-axis currents of the control winding, respectively;

[0084] i rd and i rq These are the d-axis and q-axis currents of the rotor winding, respectively.

[0085] ψ 1d and ψ 1q These are the d-axis and q-axis flux linkages of the power winding, respectively.

[0086] ψ 2d and ψ 2q These are the d-axis and q-axis flux linkages of the control winding, respectively;

[0087] ψ rd and ψ rq These are the d-axis and q-axis flux linkages of the rotor winding, respectively.

[0088] ω and ω r These represent the rotational velocity and rotor angular velocity in the dq coordinate system, respectively.

[0089] p1 and p2 are the number of pole pairs of the power winding and the number of pole pairs of the control winding, respectively;

[0090] Magnetic flux linkage equation:

[0091]

[0092]

[0093]

[0094] In the formula: L1, L2 and L r These are the self-inductances of the power winding, control winding, and rotor winding, respectively; L 1r For the mutual inductance between the power winding and the rotor winding, L 2r To control the mutual inductance between the rotor and rotor windings; electromagnetic torque equation:

[0095]

[0096] In the formula:

[0097] L1 = 1.5L m1 +L σ1

[0098] L2 = 1.5L m2 +L σ2

[0099] L r =1.5L mr +Lσr

[0100] L 1r =1.5L pr

[0101] L 2r =1.5L cr .

[0102] Other steps and parameters are the same as in specific implementation method one or two.

[0103] Specific Implementation Method Four: This implementation method differs from Specific Implementation Methods One to Three in that, in step two...

[0104] The dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system is decomposed into positive and negative order to obtain the dynamic mathematical model in the positive order dq coordinate system and the dynamic mathematical model in the negative order dq coordinate system.

[0105] Consider the perturbation of motor parameters caused by changes in operating conditions during actual motor operation;

[0106] Based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation.

[0107] Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

[0108] The specific process is as follows:

[0109] Step 2.1: Based on the improved synchronous coordinate system phase-locked loop with dual second-order generalized integrators, the dynamic mathematical model of the brushless doubly-fed motor in the two-phase rotating dq coordinate system is separated into positive and negative sequences to obtain the dynamic mathematical model in the positive-sequence dq coordinate system and the dynamic mathematical model in the negative-sequence dq coordinate system; the specific process is as follows:

[0110] The dynamic mathematical model in the positive-sequence dq coordinate system is as follows (the dynamic model of the brushless doubly fed motor in the positive-sequence coordinate system is taken as an example; the dynamic mathematical model in the negative-sequence dq coordinate system is simply a matter of changing "+" to "-"):

[0111]

[0112]

[0113]

[0114]

[0115]

[0116]

[0117] In the formula: ω r The rotor angular velocity is in the positive sequence coordinates.

[0118] This represents the positive-sequence component of the d-axis voltage of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis voltage of the power winding in positive-sequence coordinates.

[0119] This represents the positive-sequence component of the d-axis current of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis current of the power winding in positive-sequence coordinates.

[0120] This represents the positive-sequence component of the d-axis flux linkage of the power winding in the positive-sequence coordinate system. This represents the positive-sequence component of the q-axis flux linkage of the power winding in the positive-sequence coordinate system.

[0121] ω is the angular frequency of the control winding current in the positive sequence coordinate system; s is the differential operator, which represents the derivative with respect to the adjacent variable;

[0122] This represents the positive-sequence component of the d-axis voltage of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis voltage of the control winding in positive-sequence coordinates.

[0123] This represents the positive-sequence component of the d-axis current of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis current of the control winding in positive-sequence coordinates.

[0124] This represents the positive-sequence component of the d-axis flux linkage of the power winding in the positive-sequence coordinate system. This refers to the positive-sequence component of the q-axis flux linkage of the control winding in the positive-sequence coordinate system.

[0125] This represents the positive-sequence component of the rotor winding d-axis voltage in positive-sequence coordinates. This represents the positive-sequence component of the rotor winding q-axis voltage in positive-sequence coordinates.

[0126] This represents the positive-sequence component of the rotor winding d-axis current in positive-sequence coordinates. This represents the positive-sequence component of the rotor winding q-axis current in positive-sequence coordinates.

[0127] This represents the positive-sequence component of the d-axis flux linkage of the rotor winding in the positive-sequence coordinate system. This represents the positive-sequence component of the q-axis flux linkage of the rotor winding in the positive-sequence coordinate system.

[0128] Step 22: Consider the perturbations of motor parameters (inductance, resistance) caused by changes in operating conditions (working conditions, temperature and frequency) during the actual operation of the motor;

[0129] Steps 2 and 3: Based on the perturbation of motor parameters, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering the perturbation of motor parameters.

[0130] Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward decoupling compensation to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

[0131] The other steps and parameters are the same as those in one of the specific implementation methods one to three.

[0132] Specific Implementation Method 5: This implementation method differs from one of the specific implementation methods one to four in that step two-two considers the perturbation of motor parameters (resistance and inductance) caused by changes in operating conditions during the actual operation of the motor.

[0133] It is expressed as follows:

[0134]

[0135] In the formula: R i0 and L j0 These represent the initial values ​​of the stator and rotor resistances and inductances, respectively, while ΔR i and ΔL j These represent parameter perturbations of the stator and rotor resistances and inductances, respectively.

[0136] R 10 R is the initial resistance of the power winding. 20 To control the initial resistance value of the winding, R r0 The initial resistance value of the rotor winding;

[0137] ΔR1 is the resistance perturbation value of the power winding, ΔR2 is the resistance perturbation value of the control winding, ΔR r This represents the resistance perturbation value of the rotor winding;

[0138] L 10 L is the initial value of the inductance of the power winding. 20 To control the initial inductance of the winding, L r0L is the initial value of the rotor winding inductance. 1r0 L is the initial value of the mutual inductance between the power winding and the rotor winding. 2r0 To control the initial value of the mutual inductance between the rotor winding and the rotor winding;

[0139] ΔL1 is the inductance perturbation value of the power winding, and ΔL2 is the inductance perturbation value of the control winding. r ΔL is the inductance perturbation value of the rotor winding. 1r ΔL represents the mutual inductance perturbation value between the power winding and the rotor winding. 2r To control the mutual inductance perturbation value between the winding and the rotor winding;

[0140] Considering that the parameter changes are limited under actual operating conditions, the parameter perturbations of rotor resistance and inductance satisfy the following boundary conditions:

[0141]

[0142] Where: M R1 M is the upper bound of the absolute value of the resistance perturbation of the power winding. R2 To control the upper bound of the absolute value of the winding resistance perturbation, M Rr This is the upper bound of the absolute value of the resistance perturbation of the rotor winding;

[0143] M L1 M is the upper bound of the absolute value of the inductance perturbation of the power winding. L2 To control the upper bound of the absolute value of the inductance perturbation of the winding, M Lr M is the upper bound of the absolute value of the inductance perturbation of the rotor winding. L1r M is the upper bound of the absolute value of the mutual inductance perturbation between the power winding and the rotor winding. L2r This is used to control the upper bound of the absolute value of the mutual inductance perturbation between the winding and the rotor winding.

[0144] The other steps and parameters are the same as those in one of the specific implementation methods one to four.

[0145] Specific Implementation Method Six: This implementation method differs from Specific Implementation Methods One to Five in that, in steps two and three, based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward decoupling compensation to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation.

[0146] Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

[0147] In step 22, changes in operating conditions can lead to matching and non-matching uncertainties in the system, i.e., perturbations in the resistance and inductance parameters. These perturbations are reflected in the tracking error dynamic equation of the positive-sequence voltage amplitude of the power winding in step 23. Include Includes the perturbation parameters ΔR and ΔL;

[0148] The specific process is as follows:

[0149] Step 231: In the positive sequence coordinate system, establish the outer loop subsystem of the output voltage amplitude of the unmatched power winding; the specific process is as follows:

[0150] Step 2311: The input of the sliding mode controller for the outer loop subsystem of the unmatched power winding output voltage amplitude is... The output of the sliding mode controller of the outer loop subsystem of the unmatched power winding output voltage amplitude is

[0151] Step 2312: Define the error between the reference value and the actual value of the power winding output voltage amplitude as [missing value]. The outer loop subsystem of the unmatched power winding output voltage amplitude is as follows:

[0152]

[0153] In the formula: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding, for The derivative is also the outer loop subsystem of the output voltage amplitude of the unmatched power winding;

[0154] for arrive The linearized open-loop transfer function;

[0155] This represents the magnitude increment of the positive-sequence power winding voltage at the equilibrium point;

[0156] This is a reference value for the positive sequence voltage output by the power winding. Reference value for the positive sequence voltage of the power winding output The derivative;

[0157] This represents the actual value of the positive sequence output voltage of the power winding. The actual value of the positive sequence output voltage of the power winding. The derivative;

[0158] This represents the magnitude increment of the positive sequence control winding current at the equilibrium point. To control the increment of the positive sequence current output of the winding The derivative;

[0159] Step 2313: Considering that changes in operating conditions will cause perturbations in motor parameters, It becomes:

[0160]

[0161] In the formula: for arrive Linearized open-loop transfer function The initial value;

[0162] for arrive Linearized open-loop transfer function The perturbation value;

[0163] The current angular frequency of the control winding in the positive sequence coordinate system;

[0164] It is the magnitude of the positive sequence control winding current at the equilibrium point, and also the feedforward compensation amount of the outer loop subsystem of the output voltage magnitude of the unmatched power winding.

[0165] B 10 The initial value of the custom motor parameter 1 is ΔB1, and the perturbation value of the custom motor parameter 1 is ΔB1.

[0166] B 20 The initial value of the second custom motor parameter is ΔB2, and the perturbation value of the second custom motor parameter is ΔB2.

[0167] B 30 The initial value of custom motor parameter 3 is ΔB3, which is the perturbation value of custom motor parameter 3.

[0168] in,

[0169] B1 = (L 1r0 +ΔL 1r (L) 2r0 +ΔL 2r ) = B 10 +ΔB1

[0170] B2=(L 1r0 +ΔL 1r ) 2 =B 20 +ΔB2

[0171] B3=(R 10 +ΔR1)(L r0 +ΔL r ) = B 30+ΔB3

[0172] In the formula: B1, B2, and B3 are intermediate variables;

[0173]

[0174]

[0175] Steps 2314: Based on steps 2311, 2312, and 2313, obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding; the specific process is as follows:

[0176] Define the error between the actual value and the reference value of the d-axis positive sequence current of the control winding.

[0177] In the formula: To control the reference value of the d-axis positive sequence current of the winding, This represents the positive-sequence component of the d-axis current of the control winding in positive-sequence coordinates (the actual value of the positive-sequence current of the d-axis of the control winding).

[0178] according to The outer loop subsystem of the unmatched power winding output voltage amplitude considering parameter perturbation is expressed as follows:

[0179]

[0180] In the formula: To control the perturbation of the d-axis positive sequence current reference value of the winding. To control the perturbation of the d-axis positive sequence current reference value of the winding The derivative;

[0181] To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding, To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding The derivative;

[0182] Define virtual control variables Then, considering the parameter perturbation, the outer loop subsystem of the unmatched power winding output voltage amplitude is re-expressed as follows:

[0183]

[0184] Where: gain uncertainty

[0185] This is a virtual control variable;

[0186] To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding The derivative;

[0187] Let the gain be uncertain Satisfy boundary conditions The specific process is as follows:

[0188] when as well as At that time, gain uncertainty Satisfy boundary conditions

[0189]

[0190] In the formula: i 1dmax This represents the maximum value of the d-axis current of the power winding.

[0191] i 1qmax This represents the maximum value of the q-axis current of the power winding.

[0192] i 2dmax To control the maximum value of the d-axis current of the winding;

[0193] Step 232: For the positive sequence, establish the inner loop subsystem of the output current of the matching control winding;

[0194] Similarly, the process of establishing the inner and outer loop subsystems of a brushless doubly fed motor in the negative-sequence rotating coordinate system is the same as that in the positive-sequence rotating coordinate system, except that all superscripts and subscripts are "-".

[0195] During dynamic processes, coupling still exists between the positive sequence currents of the d-axis and q-axis of the control winding, meaning that the dynamic performance of the currents will affect each other. Therefore, it is necessary to decouple the positive sequence currents of the d-axis and q-axis to ensure their independent dynamic characteristics.

[0196] The other steps and parameters are the same as those in one of the specific implementation methods one to five.

[0197] Specific Implementation Method Seven: This implementation method differs from Specific Implementation Methods One through Six in that: in steps two, three, and two, for the positive sequence, a matching control winding output current inner loop subsystem is established; the specific process is as follows:

[0198] Steps two, three, two, one,

[0199] The inner loop subsystem of the matching control winding output current in the positive sequence coordinate system is as follows:

[0200]

[0201] In the formula: To control the positive sequence current vector on the winding side;

[0202] T 10 It is a diagonal matrix;

[0203] This is the feedforward compensation amount for the dq-axis current of the control winding in the positive sequence coordinate system.

[0204] The gain uncertainty is in the positive-order coordinate system;

[0205] The matching lumped uncertainty of the negative sequence component of the control winding current dq axis in the positive sequence coordinate;

[0206] In a brushless doubly fed motor independent power generation system, the inner loop subsystem with matching uncertainty is the matching control winding output current inner loop subsystem.

[0207] Taking the positive sequence as an example, the positive sequence current controller on the winding side is used to control the dq-axis current setpoint on the winding side. and For input, where the given input value To control the positive sequence voltage of the dq axis on the winding side and As a control output, it generates the modulation signal of SVPWM; (the control output is a voltage signal, which is the switching signal of the inverter switching transistors acting on the control winding side of the three-phase inverter. This kind of transformation where the voltage signal determines the on and off of the inverter switching transistors is called SVPWM).

[0208] Considering that the cross perturbation between the positive d-axis component and the positive q-axis component contains unknowns that cannot be handled by feedforward compensation, it is regarded as a matching uncertainty.

[0209] Step 2.2.2. Introduce feedforward compensation term The inner loop subsystem of the matched control winding output current in the positive-sequence coordinate system considering parameter perturbations is written as:

[0210]

[0211] In the formula:

[0212] T1 and T2 are diagonal matrices;

[0213] T1 = diag(T1′, T1′);

[0214] T2 = diag(T2′, T2′);

[0215] d / dt is the derivative of the differential;

[0216] To control the positive sequence voltage vector of the winding side;

[0217] To control the positive sequence current vector on the winding side;

[0218] This represents the cross-perturbation between the positive-sequence components of the d-axis and q-axis in the positive-sequence coordinate system; (known quantity)

[0219] The matching uncertainty caused by external disturbances (changes in external temperature);

[0220] The inner loop subsystem of the matched control winding output current in the positive sequence coordinate system considering parameter perturbations is transformed into:

[0221]

[0222] In the formula:

[0223] T 10 Let T1 be the initial value of the diagonal matrix;

[0224] T 20 Let T2 be the initial value of the diagonal matrix;

[0225] T 10 =diag(T1′0,T1′0);

[0226] T 20 =diag(T2′0,T2′0);

[0227] The initial value (known) of the cross disturbance between the positive d-axis component and the positive q-axis component in the positive sequence coordinate system;

[0228] This is the feedforward compensation term for the positive-sequence component of the control winding current dq axis in positive-sequence coordinates.

[0229] Gain uncertainty And satisfy

[0230] ΔT2 is the parametric perturbation of the diagonal matrix T2.

[0231] The other steps and parameters are the same as those in one of the specific implementation methods one to six.

[0232] Specific Implementation Method Eight: This implementation method differs from Specific Implementation Methods One through Seven in that: in step three, for the outer loop subsystem of the unmatched power winding output voltage amplitude considering motor parameter perturbations in the positive-sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the positive-sequence voltage on the power winding side to quickly and stably track the given value (set); and its stability is proven using the Lyapunov stability criterion (in the positive-sequence coordinate system, under the action of the designed sliding surface and the new approach law, the voltage error definition of the outer loop subsystem of the unmatched power winding output voltage amplitude is given by...). It will approach 0, meaning the positive sequence component of the actual output voltage amplitude of the power winding will stably track the given value.

[0233] For the inner-loop subsystem of the matched control winding output current considering motor parameter perturbations in the positive-sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the positive-sequence current of the control winding to quickly and stably track the given value (set). Its stability is proven using the Lyapunov stability criterion (in the positive-sequence coordinate system, under the action of the designed sliding surface and the novel reaching law, the current error definition of the inner-loop subsystem of the matched control winding output current is given). It will approach 0, meaning the positive sequence component of the actual output current of the control winding will stably track the given value.

[0234] For the outer loop subsystem of the output voltage amplitude of the unmatched power winding in the negative-sequence dq coordinate system considering motor parameter perturbations, a full-order terminal sliding mode controller is designed to enable the negative-sequence voltage on the power winding side to quickly and stably track the given value of 0; and its stability is proved using the Lyapunov stability criterion.

[0235] For the inner loop subsystem of the output current of the matched control winding considering motor parameter perturbation in the negative sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the negative sequence current of the control winding to quickly and stably track the given value 0; and its stability is proved using the Lyapunov stability criterion.

[0236] The specific process is as follows:

[0237] The above text gives and For the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding, a full-order terminal sliding mode controller is designed, which can make the voltage error and current error approach 0.

[0238] Step 31

[0239] The outer loop subsystem of the unmatched power winding output voltage amplitude considering motor parameter perturbations in the positive-sequence dq coordinate system is as follows:

[0240]

[0241] The inner loop subsystem of the output current of the matched control winding considering motor parameter perturbations in the positive sequence dq coordinate system is as follows:

[0242]

[0243] In the formula,

[0244] This is a virtual control variable;

[0245] For gain uncertainty;

[0246] For ΔI2+ To ΔU1 + Linearized open-loop transfer function The initial value;

[0247] For ΔI2 + To ΔU1 + Linearized open-loop transfer function The perturbation value;

[0248] Error of the d-axis output current of the control winding in the positive sequence coordinate system The derivative;

[0249] This represents the dq-axis current value of the control winding in the positive sequence coordinate system.

[0250] This is the feedforward compensation amount for the dq-axis current of the control winding in the positive sequence coordinate system.

[0251] The gain uncertainty is in the positive-order coordinate system;

[0252] The matching lumped uncertainty (known quantity) of the negative sequence component of the control winding current dq axis under positive sequence coordinates;

[0253] T 10 It is a diagonal matrix;

[0254] The outer loop subsystem of the output voltage amplitude of the unmatched power winding considering motor parameter perturbations in the negative-sequence dq coordinate system is as follows:

[0255]

[0256] The inner loop subsystem of the matching control winding output current considering motor parameter perturbations in the negative-sequence dq coordinate system is as follows:

[0257]

[0258] In the formula: This refers to the outer loop subsystem of the output voltage amplitude of the unmatched power winding in the negative sequence coordinate system. This is the reference value for the d-axis current of the control winding in the negative sequence coordinate system, and also the control signal output from the outer loop subsystem to the inner loop subsystem; For the gain uncertainty in the negative order coordinate system, satisfying for arrive Linearized open-loop transfer function initial value (and) Similarly, only the signs are different (the text only provides examples of the definition in ascending order). for arrive Linearized open-loop transfer function perturbation value (and) Similarly, only the signs are different (the text only provides examples of the definition in ascending order). The power winding d-axis output voltage error in the negative sequence coordinate system The derivative;

[0259] This represents the dq-axis current value of the control winding in the negative sequence coordinate system. This is the feedforward compensation amount for the dq-axis current of the control winding in the negative sequence coordinate system. For the gain uncertainty in the positive-sequence coordinate system, satisfying The matching lumped uncertainty (known quantity) of the negative sequence component of the control winding current dq axis in negative sequence coordinates;

[0260] Step 3.2: For the outer loop subsystem of the unmatched power winding output voltage amplitude considering motor parameter perturbations in the positive-sequence dq coordinate system, design a full-order terminal sliding mode controller to enable the positive-sequence voltage on the power winding side to quickly and stably track the given value (set); and use the Lyapunov stability criterion to prove its stability (in the positive-sequence coordinate system, under the action of the designed sliding surface and the new reaching law, the voltage error definition of the outer loop subsystem of the unmatched power winding output voltage amplitude is given by...). It will approach 0, meaning the positive sequence component of the actual output voltage amplitude of the power winding will stably track the given value.

[0261] According to the Lyapunov stability criterion It can be proven that the state trajectory of the outer loop subsystem of the non-matched power winding output positive sequence voltage amplitude will, under the action of the designed full-order terminal sliding mode control law, start from any initial state. To reach the ideal sliding surface within a limited time, along the sliding surface Maintain sliding mode dynamics. The system output tracking error will asymptotically converge to the equilibrium point.

[0262] Similarly, a full-order terminal sliding mode controller is designed for the outer loop subsystem of the negative-sequence voltage amplitude output of the unmatched power winding in the negative-sequence coordinate system, and its stability is proven using the Lyapunov stability criterion. The controller design and proof in the negative-sequence coordinate system are identical to those in the positive-sequence coordinate system, except for the use of "-" subscripts and superscripts. When the system's unmatching uncertainties are fully considered, the designed controller can achieve stable tracking of the power winding output voltage amplitude to the given voltage value.

[0263] Step 3: For the inner loop subsystem of the matching control winding output current considering motor parameter perturbations in the positive-sequence dq coordinate system. Design a full-order terminal sliding mode controller to enable the positive-sequence current of the control winding to quickly and stably track the given value (set); and prove its stability using the Lyapunov stability criterion (in the positive-sequence coordinate system, under the action of the designed sliding surface and the novel reaching law, the current error definition of the inner loop subsystem of the matching control winding output current). It will approach 0, meaning the positive sequence component of the actual current output value on the control winding side will stably track the given value.

[0264] According to the Lyapunov stability criterion It can be proven that the state trajectory of the positive sequence current tracking error subsystem on the control winding side will be within the integral actual control law. Driven by any initial state To reach the ideal full-order end sliding surface within a limited time. The sliding mode is then maintained on the sliding surface and converges to the equilibrium point within a finite time.

[0265] In the formula: For Lyapunov functions in the positive-order coordinate system, For the designed full-order terminal sliding surface s i + The derivative;

[0266] Similarly, in the negative-sequence coordinate system (in step two-one, the mathematical model of the brushless doubly-fed motor in the two-phase rotating dq coordinate system is decomposed into mathematical models in the positive-sequence coordinate system and the negative-sequence coordinate system), for the inner loop subsystem of the output negative-sequence current amplitude of the matching control winding (similar to the inner loop subsystem of the output positive-sequence current amplitude, the definition is as follows): Design a full-order terminal sliding mode controller (similar to the mathematical model under positive sequence coordinates, only with the signs reversed, and the full-order terminal sliding surface is...). There is also a full-order terminal sliding mode control law. The design of the full-order terminal sliding mode controller consists of two parts: the design of the sliding surface and the sliding mode control law. Its stability is proved using the Lyapunov stability criterion.

[0267] Similarly, a full-order terminal sliding mode controller is designed for the inner loop subsystem of the matched control winding output negative-sequence current amplitude in the negative-sequence coordinate system, and its stability is proven using the Lyapunov stability criterion. The controller design and proof in the negative-sequence coordinate system are identical to those in the positive-sequence coordinate system, except for the use of "-" subscripts and superscripts. When the unmatched uncertainties of the system are fully considered, the designed controller can achieve stable tracking of the control winding output current amplitude to the current setpoint.

[0268] The proposed full-order terminal sliding mode control method for brushless doubly-fed generators under unbalanced loads, based on compensating for the matching and unmatching uncertainties in the system, can drive the tracking error to converge quickly and provide a smooth control signal. This enables the output voltage amplitude of the power winding and the output current of the control winding to stably track the voltage amplitude setpoint and current setpoint, thereby improving the dynamic performance, control accuracy, and robustness to parameter perturbations of the BDFIG independent power generation system under unbalanced loads, and optimizing the three-phase balance of the output voltage.

[0269] The other steps and parameters are the same as those in any of the specific implementation methods one to seven.

[0270] Specific Implementation Method Nine: This implementation method differs from Specific Implementation Methods One through Eight in that, in step three-two, for the outer loop subsystem of the unmatched power winding output voltage amplitude considering motor parameter perturbations in the positive-sequence dq coordinate system, a full-order terminal sliding mode controller is designed to enable the positive-sequence voltage amplitude on the power winding side to quickly and stably track the given value (set); and its stability is proven using the Lyapunov stability criterion (in the positive-sequence coordinate system, under the action of the designed sliding surface and the novel reaching law, the voltage error definition of the outer loop subsystem of the unmatched power winding output voltage amplitude is given by...). It will approach 0, meaning the positive sequence component of the actual output voltage amplitude of the power winding will stably track the given value.

[0271] The specific process is as follows:

[0272] Step 321: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding Design the full-order terminal sliding surface as follows:

[0273]

[0274] In the formula:

[0275] The full-order terminal sliding surface of the outer loop subsystem for the output voltage amplitude of the unmatched power winding;

[0276] Sliding surface parameters is a constant and It is a constant with a value of 600;

[0277] q′ and p′ are positive odd numbers and satisfy 0<q′ / p′<1, q′ / p′ is a terminal attractor, q′ takes the value of 9, and p′ takes the value of 13;

[0278] Step 3.2.2. The composite reaching law of the outer loop subsystem for the output voltage amplitude of the unmatched power winding is as follows:

[0279]

[0280] In the formula:

[0281] Full-order terminal sliding surface designed for the outer loop subsystem of the power winding The derivative of is defined as the outer ring composite reaching law. for saturation function;

[0282] c1′, c2′, a, η, k and All are constants; c1′>0, c2′>0, 0<a<1, 0<η<1, k>0,

[0283]

[0284] Where Δ represents the boundary layer and Δ is a constant; switching control is used outside the boundary layer; and linearization control is used inside the boundary layer.

[0285] Step 323: Apply the full-order terminal sliding surface With the law of compound convergence By performing simultaneous integration of the integrals, we can obtain the solution. based on Seeking according to Seek based on Seeking

[0286] Based on equivalent control law and actual control law The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem for obtaining the output voltage amplitude of the unmatched power winding

[0287] The specific process is as follows:

[0288] The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem of the unmatched power winding output voltage amplitude The expression is:

[0289]

[0290] Equivalent control law The expression is:

[0291]

[0292] Actual control law The expression is:

[0293]

[0294] In the formula:

[0295] The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem for the output voltage amplitude of the unmatched power winding;

[0296] This is an equivalent control law;

[0297] This is the actual control law;

[0298] t is the upper bound of the integral, and τ is the sliding surface parameter;

[0299] Specifically, c1′=800, a=2 / 3, eta=3 / 5, k=30, c2′=800,

[0300] The other steps and parameters are the same as those in one of the specific implementation methods one to eight.

[0301] Specific Implementation Method Ten: This implementation method differs from Specific Implementation Methods One through Nine in that step three-three concerns the inner loop subsystem of the matching control winding output current considering motor parameter perturbations in the positive-sequence dq coordinate system. Design a full-order terminal sliding mode controller to enable the positive-sequence current of the control winding to quickly and stably track the given value (set); and prove its stability using the Lyapunov stability criterion (in the positive-sequence coordinate system, under the action of the designed sliding surface and the novel reaching law, the current error definition of the inner loop subsystem of the matching control winding output current). It will approach 0, meaning the positive sequence component of the actual current output value on the control winding side will stably track the given value.

[0302] The specific process is as follows:

[0303] Step 331: To ensure tracking error and and and (definition is) As given above, convergence occurs within a finite time for the inner loop subsystem of the matched control winding output current considering motor parameter perturbations in the positive-sequence dq coordinate system (Equation [1]). Design the full-order terminal sliding surface:

[0304]

[0305] In the formula: To match the full-order terminal sliding surface of the inner loop subsystem of the control winding output current;

[0306] It is a positive definite diagonal matrix.

[0307] To control the error between the given and actual values ​​of the positive sequence output current of the winding, the following formula is defined:

[0308]

[0309] q′ and p′ are positive odd numbers and satisfy 0<q′ / p′<1, q′ / p′ is a terminal attractor, q′ takes the value of 9, and p′ takes the value of 13;

[0310] To control the error between the given value and the actual value of the positive sequence current output of the winding. The derivative;

[0311] This is a custom parameter with a value of 6000. This is a custom parameter with a value of 6000.

[0312] Step 332: The composite reaching law of the inner loop subsystem of the output current of the matching control winding is as follows:

[0313]

[0314] In the formula: Full-order terminal sliding surface designed for the outer loop subsystem of the power winding The derivative of is defined as the outer ring composite reaching law;

[0315] for saturation function;

[0316] c1′, c2′, a, η, k and All are constants;

[0317] c1′>0, c2′>0, 0<a<1, 0<η<1, k>0,

[0318] Step 333: Adjust the full-order terminal sliding surface With the law of compound convergence By performing simultaneous integration of the integrals, we can obtain the solution. based on and Find the equivalent control law Based on Find the actual control law

[0319] Based on equivalent control law and actual control law Design the full-order terminal sliding mode control law for the inner loop subsystem;

[0320] The specific process is as follows:

[0321]

[0322] Equivalent control law The expression is:

[0323]

[0324] Actual control law The expression is:

[0325]

[0326] In the formula: This is an equivalent control law; This is the actual control law;

[0327] This is the full-order terminal sliding mode control law for the inner-loop subsystem;

[0328] This is a custom parameter with a value of 2500.

[0329] τ is the integral operator.

[0330] The other steps and parameters are the same as those in any of the specific implementation methods one to nine.

[0331] This invention may have other embodiments. Without departing from the spirit and essence of this invention, those skilled in the art can make various corresponding changes and modifications according to this invention, but these corresponding changes and modifications should all fall within the protection scope of the appended claims.

Claims

1. A method for full-order terminal sliding mode control of a brushless doubly-fed motor under unbalanced load, characterized in that: The specific process of the method is as follows: Step 1: Collect the voltage and current on the two stator winding sides of the brushless doubly fed motor to obtain the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system. Perform coordinate transformation on the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to obtain the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system. Step 2: Decompose the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system into positive and negative order to obtain the dynamic mathematical model in the positive order dq coordinate system and the dynamic mathematical model in the negative order dq coordinate system. Consider motor parameter perturbations; Based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation. Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation. Step 3: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the positive sequence dq coordinate system considering the perturbation of motor parameters, design a full-order terminal sliding mode controller to make the positive sequence voltage amplitude on the power winding side stably track the given value; For the inner loop subsystem of the output current of the matched control winding considering motor parameter perturbation in the positive sequence dq coordinate system, a full-order terminal sliding mode controller is designed to make the positive sequence current of the control winding stably track the given value. For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the negative sequence dq coordinate system considering motor parameter perturbation, a full-order terminal sliding mode controller is designed to make the negative sequence voltage on the power winding side stably track the given value 0. For the inner loop subsystem of the output current of the matching control winding considering motor parameter perturbation in the negative sequence dq coordinate system, a full-order terminal sliding mode controller is designed to make the negative sequence current of the control winding stably track the given value 0. Step three also includes: Step 321: Design the full-order terminal sliding surface for the outer loop subsystem of the output voltage amplitude of the unmatched power winding. In the formula: The sliding surface of the outer loop subsystem for the output voltage amplitude of the unmatched power winding is the full-order terminal sliding surface, and the sliding surface parameters are as follows: is a constant and , and It is a positive odd number and satisfies , For terminal attractors; Step 3.2.

2. The composite reaching law of the outer loop subsystem for the output voltage amplitude of the unmatched power winding is as follows: In the formula: Full-order terminal sliding surface designed for the outer loop subsystem of the power winding The derivative of is defined as the outer ring composite reaching law. for saturation function; , , , , and All are constants; , , , , , ; in, For the boundary layer, Δ is a constant; Step 323: Apply the full-order terminal sliding surface With the law of compound convergence By performing simultaneous integration of the integrals, we can obtain the solution. ,based on Seeking ,according to Seek ,based on Seeking ; Based on equivalent control law and actual control law The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem for obtaining the output voltage amplitude of the unmatched power winding .

2. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load as described in claim 1, characterized in that: In step one, the voltage and current on the two stator winding sides of the brushless doubly fed motor are collected to obtain the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system. The dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system is transformed to obtain the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system. The specific process is as follows: Step 1: Establish a dynamic mathematical model of the brushless doubly fed motor in a three-phase stationary coordinate system; Step 1 and Step 2: Using Clark and Park coordinate transformation, transform the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system to the dynamic mathematical model in the two-phase rotating dq coordinate system. In step one by one, the dynamic mathematical model of the brushless doubly fed motor in the three-phase stationary coordinate system includes voltage equation, flux linkage equation and electromagnetic torque equation. Voltage equation: In the formula: The voltage vector on the power winding side. To control the voltage vector on the winding side, The rotor-side voltage vector; This is the current vector on the power winding side. To control the current vector on the winding side, The rotor current vector; For the flux linkage on the power winding side, To control the flux linkage on the winding side, For rotor flux linkage; The resistance is on the power winding side. To control the resistance on the winding side, The resistance of the rotor winding; For time, For time Differentiation operation; Magnetic flux linkage equation: In the formula: , and These are the self-inductance matrices for the power winding, control winding, and flux linkage winding, respectively. This represents the mutual inductance matrix of the power winding and the rotor winding; To control the mutual inductance matrix of the control winding and the rotor winding; for The transpose of a matrix. For matrix The transpose of the matrix; In the formula: For the single-phase leakage inductance on the power winding side, To control single-phase leakage inductance on the winding side; This is a single-phase magnetizing inductor on the power winding side. To control the single-phase magnetizing inductance on the winding side; This is the single-phase magnetizing inductance of the rotor winding. For single-phase leakage inductance of the rotor winding; This refers to the mutual inductance amplitude between the power winding side and the rotor winding. To control the mutual inductance amplitude between the rotor winding and the rotor winding; This is the initial position difference between the A-phase axis on the power winding side and the A-phase axis on the control winding side; The rotor position angle; and These are the number of pole pairs of the power winding and the number of pole pairs of the control winding, respectively. Electromagnetic torque equation: In the formula: The value is the electromagnetic torque; the superscript T indicates the transpose. This is the transpose of the power winding side current vector matrix. For rotor position angle Differentiation operation.

3. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load according to claim 2, characterized in that: In steps one and two, the Clark and Park coordinate transformations are used to transform the dynamic mathematical model of the brushless doubly-fed motor in the three-phase stationary coordinate system to the dynamic mathematical model in the two-phase rotating dq coordinate system; the specific process is as follows: The dynamic mathematical model of a brushless doubly fed motor in a two-phase rotating dq coordinate system includes voltage equations, flux linkage equations, and electromagnetic torque equations. Voltage equation: In the formula: and These are the d-axis and q-axis voltages of the power winding, respectively. and These are the d-axis and q-axis voltages of the control winding, respectively; and These are the d-axis and q-axis voltages of the rotor winding, respectively. and These are the d-axis and q-axis currents of the power winding, respectively. and These are the d-axis and q-axis currents of the control winding, respectively; and These are the d-axis and q-axis currents of the rotor winding, respectively. and These are the d-axis and q-axis flux linkages of the power winding, respectively. and These are the d-axis and q-axis flux linkages of the control winding, respectively. and These are the d-axis and q-axis flux linkages of the rotor winding, respectively. and These represent the rotational velocity and rotor angular velocity in the dq coordinate system, respectively. and These are the number of pole pairs of the power winding and the number of pole pairs of the control winding, respectively. Magnetic flux linkage equation: In the formula: , and These are the self-inductances of the power winding, control winding, and rotor winding, respectively. For the mutual inductance between the power winding and the rotor winding, To control the mutual inductance between the rotor winding and the rotor winding; Electromagnetic torque equation: In the formula: , , , , .

4. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load as described in claim 3, characterized in that: In step two, the dynamic mathematical model of the brushless doubly fed motor in the two-phase rotating dq coordinate system is decomposed into positive and negative order to obtain the dynamic mathematical model in the positive order dq coordinate system and the dynamic mathematical model in the negative order dq coordinate system. Consider motor parameter perturbations; Based on motor parameter perturbation, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering motor parameter perturbation. Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation. The specific process is as follows: Step 2.1: Based on the improved synchronous coordinate system phase-locked loop with dual second-order generalized integrators, the dynamic mathematical model of the brushless doubly-fed motor in the two-phase rotating dq coordinate system is separated into positive and negative sequences to obtain the dynamic mathematical model in the positive-sequence dq coordinate system and the dynamic mathematical model in the negative-sequence dq coordinate system; the specific process is as follows: The dynamic mathematical model in the orthogonal dq coordinate system is as follows: In the formula: The rotor angular velocity is in the positive sequence coordinates. It is a differential operator; This represents the positive-sequence component of the d-axis voltage of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis voltage of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the d-axis current of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis current of the power winding in positive-sequence coordinates. This represents the positive-sequence component of the d-axis flux linkage of the power winding in the positive-sequence coordinate system. This represents the positive-sequence component of the q-axis flux linkage of the power winding in the positive-sequence coordinate system. The current angular frequency of the control winding in the positive sequence coordinate system; This represents the positive-sequence component of the d-axis voltage of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis voltage of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the d-axis current of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the q-axis current of the control winding in positive-sequence coordinates. This represents the positive-sequence component of the d-axis flux linkage of the power winding in the positive-sequence coordinate system. This refers to the positive-sequence component of the q-axis flux linkage of the control winding in the positive-sequence coordinate system. This represents the positive-sequence component of the rotor winding d-axis voltage in positive-sequence coordinates. This represents the positive-sequence component of the rotor winding q-axis voltage in positive-sequence coordinates. This represents the positive-sequence component of the rotor winding d-axis current in positive-sequence coordinates. This represents the positive-sequence component of the rotor winding q-axis current in positive-sequence coordinates. This represents the positive-sequence component of the d-axis flux linkage of the rotor winding in the positive-sequence coordinate system. This represents the positive-sequence component of the q-axis flux linkage of the rotor winding in the positive-sequence coordinate system. Step 22: Consider the perturbation of motor parameters caused by changes in operating conditions during actual motor operation; Steps 2 and 3: Based on the perturbation of motor parameters, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering the perturbation of motor parameters. Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward decoupling compensation to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation.

5. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load as described in claim 4, characterized in that: The motor parameter perturbation caused by the change in operating conditions in step two is as follows: In the formula: and These represent the initial values ​​of the stator and rotor resistances and inductances, respectively. and These represent parameter perturbations of the stator and rotor resistances and inductances, respectively. The initial resistance value of the power winding. To control the initial resistance value of the winding, The initial resistance value of the rotor winding; This is the resistance perturbation value of the power winding. To control the resistance perturbation value of the winding, This represents the resistance perturbation value of the rotor winding; The initial value of the inductance of the power winding. To control the initial value of the winding inductance, The initial value of the rotor winding inductance. This represents the initial value of the mutual inductance between the power winding and the rotor winding. To control the initial value of the mutual inductance between the rotor winding and the rotor winding; This is the inductance perturbation value of the power winding. To control the inductance perturbation value of the winding, This represents the inductance perturbation value of the rotor winding. This represents the mutual inductance perturbation value between the power winding and the rotor winding. To control the mutual inductance perturbation value between the winding and the rotor winding; The parameter perturbations of rotor resistance and inductance satisfy the following boundary conditions: In the formula: This is the upper bound of the absolute value of the resistance perturbation of the power winding. To control the upper bound of the absolute value of the resistance perturbation of the winding, This is the upper bound of the absolute value of the resistance perturbation of the rotor winding; This is the upper bound of the absolute value of the inductance perturbation of the power winding. To control the upper bound of the absolute value of the inductance perturbation of the winding, This is the upper bound of the absolute value of the inductance perturbation of the rotor winding. This is the upper bound of the absolute value of the mutual inductance perturbation between the power winding and the rotor winding. This is used to control the upper bound of the absolute value of the mutual inductance perturbation between the winding and the rotor winding.

6. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load according to claim 5, characterized in that: In steps two and three, based on the perturbation of motor parameters, the dynamic mathematical model in the positive sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the positive sequence dq coordinate system considering the perturbation of motor parameters. Based on motor parameter perturbation, the dynamic mathematical model in the negative sequence dq coordinate system is fed forward and decoupled to obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding and the inner loop subsystem of the output current of the matched control winding in the negative sequence dq coordinate system considering motor parameter perturbation. The specific process is as follows: Step 231: In the positive sequence coordinate system, establish the outer loop subsystem of the output voltage amplitude of the unmatched power winding; the specific process is as follows: Step 2311: The input of the sliding mode controller for the outer loop subsystem of the unmatched power winding output voltage amplitude is... The output of the sliding mode controller of the outer loop subsystem of the unmatched power winding output voltage amplitude is ; Step 2312: Define the error between the reference value and the actual value of the power winding output voltage amplitude as [missing value]. The outer loop subsystem of the output voltage amplitude of the unmatched power winding is obtained as follows: In the formula: For the outer loop subsystem of the output voltage amplitude of the unmatched power winding, for The derivative; for arrive The linearized open-loop transfer function; This represents the magnitude increment of the positive-sequence power winding voltage at the equilibrium point; This is a reference value for the positive sequence voltage output by the power winding. Reference value for the positive sequence voltage of the power winding output The derivative; This represents the actual value of the positive sequence output voltage of the power winding. The actual value of the positive sequence voltage U1 of the power winding output + The derivative; This represents the magnitude increment of the positive sequence control winding current at the equilibrium point. To control the increment of the positive sequence current output of the winding The derivative; Step 2313: Considering that changes in operating conditions will cause perturbations in motor parameters, It becomes: In the formula: for arrive Linearized open-loop transfer function initial value, for arrive Linearized open-loop transfer function The perturbation value; The current angular frequency of the control winding in the positive sequence coordinate system; The positive sequence control winding current amplitude at the equilibrium point; To set the initial values ​​for the first custom motor parameter, For the perturbation value of custom motor parameter one; The initial values ​​for the second custom motor parameter are... To customize the perturbation value of motor parameter two, The initial values ​​for custom motor parameter three, For the perturbation value of custom motor parameter three; in, In the formula: , , As an intermediate variable; Steps 2314: Based on steps 2311, 2312, and 2313, obtain the outer loop subsystem of the output voltage amplitude of the unmatched power winding; the specific process is as follows: Define the error between the actual value and the reference value of the d-axis positive sequence current of the control winding. ; In the formula: To control the reference value of the d-axis positive sequence current of the winding, The positive-sequence component of the d-axis current of the control winding in positive-sequence coordinates; according to and Then, considering the parameter perturbation, the outer loop subsystem of the unmatched power winding output voltage amplitude is expressed as: In the formula: To control the perturbation of the d-axis positive sequence current reference value of the winding. To control the perturbation of the d-axis positive sequence current reference value of the winding The derivative; To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding, To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding The derivative; Define virtual control variables Then, considering the parameter perturbation, the outer loop subsystem of the unmatched power winding output voltage amplitude is re-expressed as: Where: gain uncertainty ; This is a virtual control variable; To control the error between the actual value and the reference value of the d-axis positive sequence current of the winding The derivative; Let the gain be uncertain Satisfy boundary conditions The specific process is as follows: when , as well as At that time, gain uncertainty Satisfy boundary conditions ; In the formula: This represents the maximum value of the d-axis current of the power winding. This represents the maximum value of the q-axis current of the power winding; To control the maximum value of the d-axis current of the winding; Step 232: For the positive sequence, establish the inner loop subsystem of the output current of the matching control winding.

7. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load as described in claim 6, characterized in that: In steps two and three, for the positive sequence, an inner loop subsystem for the output current of the matching control winding is established; the specific process is as follows: Step Two Three Two One The inner loop subsystem of the matching control winding output current in the positive sequence coordinate system is as follows: In the formula: To control the positive sequence current vector on the winding side; It is a diagonal matrix. This is the feedforward compensation amount for the dq-axis current of the control winding in the positive sequence coordinate system. The gain uncertainty is in the positive-order coordinate system. The matching lumped uncertainty of the negative sequence component of the control winding current dq axis in the positive sequence coordinate; Step 2.2.

2. Introduce feedforward compensation term The inner loop subsystem of the matched control winding output current in the positive sequence coordinate system considering parameter perturbations is written as: In the formula: , It is a diagonal matrix; d / dt is the derivative of the differential. diagonal matrix The initial value of ; To control the positive sequence voltage vector of the winding side; To control the positive sequence current vector on the winding side; This refers to the cross-perturbation between the positive-sequence components of the d-axis and the positive-sequence components of the q-axis in the positive-sequence coordinate system. Matching uncertainty caused by external disturbances; This represents the initial value of the cross-perturbation between the positive-sequence components of the d-axis and the positive-sequence components of the q-axis in the positive-sequence coordinate system. The inner loop subsystem of the matched control winding output current in the positive sequence coordinate system considering parameter perturbations is transformed into: In the formula: diagonal matrix The initial value; This is the feedforward compensation term for the positive-sequence component of the control winding current dq axis in positive-sequence coordinates. Gain uncertainty And satisfy ; diagonal matrix The parameter perturbation.

8. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load according to claim 7, characterized in that: In step three, for the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the positive sequence dq coordinate system considering the perturbation of motor parameters, a full-order terminal sliding mode controller is designed so that the positive sequence voltage amplitude on the power winding side can stably track the given value. For the inner loop subsystem of the output current of the matched control winding considering motor parameter perturbation in the positive sequence dq coordinate system, a full-order terminal sliding mode controller is designed to make the positive sequence current of the control winding stably track the given value. For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the negative sequence dq coordinate system considering motor parameter perturbation, a full-order terminal sliding mode controller is designed to make the negative sequence voltage on the power winding side stably track the given value 0. For the inner loop subsystem of the output current of the matching control winding considering motor parameter perturbation in the negative sequence dq coordinate system, a full-order terminal sliding mode controller is designed to make the negative sequence current of the control winding stably track the given value 0. The specific process is as follows: Step 3.1 The outer loop subsystem of the unmatched power winding output voltage amplitude considering motor parameter perturbations in the positive sequence dq coordinate system is as follows: The inner loop subsystem of the output current of the matched control winding considering motor parameter perturbations in the positive sequence dq coordinate system is as follows: In the formula, For virtual control variables, For gain uncertainty, For ΔI2 + To ΔU1 + Linearized open-loop transfer function initial value, For ΔI2 + To ΔU1 + Linearized open-loop transfer function The perturbation value; Error of the d-axis output current of the control winding in the positive sequence coordinate system The derivative; This represents the dq-axis current value of the control winding in the positive sequence coordinate system. This is the feedforward compensation amount for the dq-axis current of the control winding in the positive sequence coordinate system. The gain uncertainty is in the positive-order coordinate system. The matching lumped uncertainty of the negative-sequence component of the control winding current dq-axis in the positive-sequence coordinate system is given. It is a diagonal matrix; The outer loop subsystem of the output voltage amplitude of the unmatched power winding considering motor parameter perturbations in the negative-sequence dq coordinate system is as follows: The inner loop subsystem of the matching control winding output current considering motor parameter perturbations in the negative-sequence dq coordinate system is as follows: In the formula: This refers to the outer loop subsystem of the output voltage amplitude of the unmatched power winding in the negative sequence coordinate system. This is the reference value for the d-axis current of the control winding in the negative sequence coordinate system, and also the control signal output from the outer loop subsystem to the inner loop subsystem; For the gain uncertainty in the negative order coordinate system, satisfying ; for arrive Linearized open-loop transfer function initial value, for arrive Linearized open-loop transfer function The perturbation value, The power winding d-axis output voltage error in the negative sequence coordinate system The derivative; This represents the dq-axis current value of the control winding in the negative sequence coordinate system. This is the feedforward compensation amount for the dq-axis current of the control winding in the negative sequence coordinate system. For the gain uncertainty in the positive-sequence coordinate system, satisfying ; The matching lumped uncertainty of the negative-sequence component of the control winding current dq axis under negative-sequence coordinates; Step 3.

2. For the outer loop subsystem of the output voltage amplitude of the unmatched power winding under the positive sequence dq coordinate system considering the perturbation of motor parameters, design a full-order terminal sliding mode controller so that the positive sequence voltage amplitude on the power winding side can stably track the given value. Step 3: For the inner loop subsystem of the matching control winding output current considering motor parameter perturbation in the positive sequence dq coordinate system, design a full-order terminal sliding mode controller so that the positive sequence current of the control winding stably tracks the given value.

9. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load as described in claim 8, characterized in that: Step three, two, three: the full-order terminal sliding surface With the law of compound convergence By performing simultaneous integration of the integrals, we can obtain the solution. ,based on Seeking ,according to Seek ,based on Seeking ; Based on equivalent control law and actual control law The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem for obtaining the output voltage amplitude of the unmatched power winding ; The specific process is as follows: The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem of the unmatched power winding output voltage amplitude The expression is: Equivalent control law The expression is: Actual control law The expression is: In the formula: The overall control law of the full-order terminal sliding mode controller of the outer loop subsystem for the output voltage amplitude of the unmatched power winding; This is an equivalent control law; This is the actual control law; This is the upper bound of the integral. These are the parameters of the sliding surface.

10. The full-order terminal sliding mode control method for a brushless doubly-fed motor under unbalanced load according to claim 9, characterized in that: In step three, for the inner loop subsystem of the matching control winding output current considering motor parameter perturbation in the positive sequence dq coordinate system, a full-order terminal sliding mode controller is designed so that the positive sequence current of the control winding can stably track the given value. The specific process is as follows: Step 331: For the inner loop subsystem of the matching control winding output current considering motor parameter perturbations in the positive sequence dq coordinate system, design the full-order terminal sliding surface: In the formula: To match the full-order terminal sliding surface of the inner loop subsystem of the control winding output current; It is a positive definite diagonal matrix; To control the error between the given and actual values ​​of the positive sequence output current of the winding, the following formula is defined: ; and It is a positive odd number and satisfies , For terminal attractors; To control the error between the given value and the actual value of the positive sequence current output of the winding. The derivative; Step 332: The composite reaching law of the inner loop subsystem of the output current of the matching control winding is as follows: In the formula: Full-order terminal sliding surface designed for the outer loop subsystem of the power winding The derivative of is defined as the outer ring composite reaching law. for saturation function; , , , , and All are constants; , , , , , ; Step 333: Adjust the full-order terminal sliding surface With the law of compound convergence By performing simultaneous integration of the integrals, we can obtain the solution. ,based on and Find the equivalent control law Based on Find the actual control law ; Based on equivalent control law and actual control law Design the full-order terminal sliding mode control law for the inner loop subsystem; The specific process is as follows: Equivalent control law The expression is: Actual control law The expression is: In the formula: This is an equivalent control law; This is the actual control law; This is the full-order terminal sliding mode control law for the inner-loop subsystem; These are the motor parameters; It is an integral operator.

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