Medium antenna for telesensory integration

Abstract

A dielectric antenna comprising a substrate layer, a ground layer on top of the substrate layer, and a dielectric body on top of the ground layer. Wherein the dielectric body has an elongated shape. The dielectric antenna further comprises a perturbation structure on top of the dielectric body for cutting the link between high order modes in the resonant part of the dielectric antenna, maintaining the longitudinal resonant mode of the dielectric antenna and leaking the transversal transmission mode of the dielectric antenna. The invention aims to investigate novel antenna solutions for integrated sensing and communication (ISAC) applications.

Description

Technical Field

[0001] This invention relates to dielectric antennas, filling a gap in antennas designed for integrated communication and sensing. Background Technology

[0002] The biggest difference between 6G wireless communication and its predecessors will be its ability to intelligently connect people, vehicles, robots, and objects, integrating humanity into the digital society and smart cities. Experts in industry and academia have already begun preparing and researching 6G networks with integrated sensing and communication infrastructure. Sensing capabilities are key to intelligent systems or smart cities, enabling high-precision positioning, highly resilient network reconfiguration, and ultra-fast and secure communication. This unique characteristic of sensor-assisted communication can deliver higher quality and more reliable link performance, enabling existing precise beamforming technologies to rebuild beams faster without human intervention for efficient tracking communication and high network capacity. Future cellular networks will utilize dense sectors for sensing functions, effectively utilizing spectrum, channel capacity, high-precision positioning, and efficient power management. In fact, new and advanced antenna technologies will be indispensable components supporting the realization of 6G networks.

[0003] Existing broadband, MIMO (Multiple-Input Multiple-Output), millimeter-wave (mmW), beamforming, and reconfigurable antenna technologies have become core technologies in the fields of communication and sensing, respectively. However, simply combining these two types of systems makes it difficult to unleash the full potential of 6G intelligent networks. Therefore, an improved antenna is needed that can simultaneously support the sensing and communication functions of future communication networks. Summary of the Invention

[0004] This invention provides a dielectric antenna comprising a substrate, a ground layer above the substrate, and a dielectric body above the ground layer. The dielectric body has an elongated shape. The dielectric antenna further includes a perturbation structure located on top of the dielectric body for cutting connections between higher-order modes in the longitudinal resonant operating mode of the dielectric antenna.

[0005] Preferably, the perturbation structure includes a metal wire disposed on top of the dielectric body.

[0006] More preferably, the metal wire is perpendicular to the longitudinal direction of the medium body.

[0007] In one embodiment, the perturbation structure further includes a plurality of air holes on the dielectric body, which are adjacent to the metal wires.

[0008] In another specific embodiment, multiple air vents are symmetrically located on both sides of the metal wire.

[0009] In another specific embodiment, the perturbation structure includes multiple metal lines that are parallel to each other and perpendicular to the longitudinal direction of the dielectric bulk.

[0010] In some exemplary embodiments, multiple metal lines are distributed at equal intervals.

[0011] In one specific embodiment, the dielectric antenna can be further decomposed into a plurality of longitudinally arranged antenna elements. Each antenna element corresponds to two adjacent elements and an intermediate portion of a plurality of perturbation structures. The plurality of antenna elements are adjacent to each other at one of a plurality of metal lines.

[0012] In another embodiment, the substrate layer includes a first microstrip feed and a second microstrip feed. The ground layer includes a first coupling trench that intersects the first microstrip feed in its longitudinal direction, and a second coupling trench that intersects the second microstrip feed in its longitudinal direction.

[0013] Preferably, the first microstrip feed line and the second microstrip feed line correspond to a first antenna element and a second antenna element defined in the dielectric body, respectively. The first antenna element and the second antenna element are adjacent to each other at the metal line.

[0014] In one specific embodiment, a first coplanar waveguide feed port and a second coplanar waveguide feed port are formed at both ends of the dielectric body.

[0015] Preferably, the first coplanar waveguide feed port and the second coplanar waveguide feed port correspond to the two side ports of the transverse traveling wave antenna defined in the dielectric body, respectively.

[0016] Therefore, embodiments of the present invention provide a novel antenna solution for Integrated Sensing and Communication (ISAC) applications, addressing the bottlenecks of existing sensing and communication networks by introducing a compact, highly integrated, cost-effective, and high-gain antenna solution to realize ISAC functionality. Thanks to its fusion design, this laterally fully interconnected antenna can simultaneously provide directional and frequency-scanning radiation in the millimeter wave, thus achieving both phased array-based communication performance and frequency-scanning-based sensing performance.

[0017] The above description of the invention is neither intended to define the invention claimed in this application (as described in the claims) nor to limit the scope of the invention in any way. Attached Figure Description

[0018] The above and further features of the present invention will readily become apparent from the following description of embodiments. These embodiments are described by way of example only, in conjunction with the accompanying drawings, wherein:

[0019] Figure 1a shows the structure of a conventional DRA (dielectric resonator antenna) with a single dielectric block.

[0020] Figure 1b shows another conventional 1×2 DRA structure with two unconnected media blocks.

[0021] Figure 1c The structure of a 1×2 dielectric antenna having a connection structure for two antenna elements is shown according to an embodiment of the present invention.

[0022] Figure 1d The structure of a 1×2 dielectric antenna, physically connected and separated by a perturbation structure, is shown according to another embodiment of the present invention.

[0023] Figure 2 It is displayed Figure 1d Cross-sectional views of different layers in a dielectric antenna.

[0024] Figure 3 A 1×8 dielectric antenna according to another embodiment of the present invention is shown, the element design of which is similar to... Figure 1d The antennas are the same.

[0025] Figure 4a Showing Figure 3 Simulation S-parameter results for one port of the antenna.

[0026] Figure 4b Showing Figure 3 Simulated gain results (frequency response) of the antenna.

[0027] Figure 5a , 5b 5c and 5c respectively showed Figure 3 Simulated radiation patterns of the antenna at 25 GHz, 28 GHz, and 30 GHz.

[0028] Figure 6 It is a display Figure 3 A schematic diagram illustrating the working principle of an antenna.

[0029] Figure 7a , 7b 7c and 7c respectively showed Figure 3 The resonant mode, transmission mode, and superposition mode of the antenna.

[0030] Figure 8 Showing with Figure 1c The simulation results and gain results are similar to those of the antennas in the model, but the antenna lengths are different.

[0031] Figure 9 The simulated gain results for two 1×8 antennas are shown, with the element designs of these two antennas being respectively compared to... Figure 1c and 1d The antennas are the same.

[0032] Figure 10a The simulated gain results for a 1×2 antenna are shown, with the antenna's element design and... Figure 1d The antennas are the same, but the dielectric bodies have different lengths.

[0033] Figure 10b Showing Figure 10a Simulation of 1×2 antennas |S11| and |S21|.

[0034] Figure 11a The simulated gain results for a 1×2 antenna are shown, with the antenna's element design and... Figure 1d The antennas are the same, but the metal strips have different widths.

[0035] Figure 11b Showing Figure 11a Simulation of 1×2 antennas |S11| and |S21|.

[0036] Figure 12a The simulated gain results for a 1×2 antenna are shown, with the antenna's element design and... Figure 1d The antennas are the same, but the metal strips are of different lengths.

[0037] Figure 12b Showing Figure 12a Simulation of 1×2 antennas |S11| and |S21|.

[0038] Figure 13 A 1×8 dielectric antenna according to another embodiment of the present invention is shown, its element design being similar to... Figure 1d The antennas are the same, but it has a SIW-DIL port.

[0039] Figure 14 A 1×16 dielectric antenna according to another embodiment of the present invention is shown, its element design being similar to... Figure 9 The antennas are the same.

[0040] Figure 15a , 15b 15c, 15d, 15e and 15f respectively show different perturbation structures suitable for dielectric antennas according to embodiments of the present invention. Figure 16 The structure of a 1×8 dielectric antenna according to another embodiment of the present invention is shown, wherein the perturbation structure of the antenna contains air holes.

[0041] Figure 17a Showing Figure 16 The leakage radiation performance of the antenna in the image.

[0042] Figure 17b Showing Figure 16 The standing wave radiation performance of the antenna in the image.

[0043] Figure 18 The structure of a 1×16 dielectric antenna according to another embodiment of the present invention is shown, which contains an air via in the perturbation structure.

[0044] Figure 19a Showing Figure 18 The S-parameter performance of the antenna in the image.

[0045] Figure 19b Showing Figure 18 The standing wave radiation performance of the antenna in the image.

[0046] Figure 19c Showing Figure 18 The leakage radiation performance of the antenna in the image.

[0047] Figure 20 The sensor beam manipulation of a leaky mode of a dielectric antenna according to another embodiment of the present invention is shown.

[0048] Figure 21a Showing Figure 20 Communication beam manipulation of the antenna standing wave mode.

[0049] Figure 21b Showing Figure 21a The radiation patterns of an antenna at different beam angles when phase reversal is applied to different numbers of antenna elements in the antenna.

[0050] Figure 21c Showing Figure 21a The antenna exhibits radiation patterns at different beam angles when different numbers of antenna elements are excited.

[0051] In the specification and drawings, similar numbers denote similar portions in several of the described embodiments. Detailed Implementation

[0052] Different embodiments of the present invention provide solutions for generating millimeter-wave leaky-wave and standing-wave radiation on the same antenna, thus providing two antenna functions. As a basic introduction, leaky-wave radiation is a type of traveling-wave radiation. Traveling-wave antennas use traveling waves on a guiding structure as the primary radiation mechanism, while standing-wave antennas radiate using standing waves in a resonator structure. The biggest difference between standing-wave and leaky-wave antennas lies in their radiation performance at different frequencies. For standing-wave antennas, all radiation patterns are oriented in the same direction; while for leaky-wave antennas, the direction of the radiation pattern varies with frequency due to the phase difference of each radiating element. This method can also be used for standing-wave antennas, a technique known as phased array technology. The advantage of leaky-wave antennas is that patterns in all directions are radiated simultaneously, which improves the system's time efficiency. Conversely, standing-wave antennas using phased array networks can sequentially scan all directions at a single frequency, saving system spectrum resources.

[0053] Figures 1a and 1b illustrate two conventional DRAs. The antenna in Figure 1a is a simple DRA with a single dielectric block 6, excited by aperture 4 and slot 2. This antenna is suitable for covering the frequency range of 24 GHz to 28 GHz, providing higher communication capacity in the millimeter-wave region. The dielectric block 6 is used instead of metal because dielectric materials do not have metal losses. For the antenna in Figure 1a, the electric field distribution at three different frequencies within the aforementioned frequency range follows the basic pattern, TE111.

[0054] The antenna in Figure 1b is a small 1×2 subarray containing two independent dielectric blocks 16. Each dielectric block 16, along with its corresponding aperture 14 and corresponding slot 12, defines a separate antenna element. Due to the coupling between the two antenna elements, the electric field distribution will differ slightly at different frequencies, while the radiation performance can remain stable. However, a drawback associated with the DRA in Figures 1a and 1b is a manufacturing challenge.

[0055] Let's look again. Figure 1c This shows a 1×2 dielectric antenna similar to Figure 1b, but with two dielectric blocks connected together (i.e., the dielectric blocks become a continuous single block, indicated by arrow 15). The single dielectric block 15 has an elongated shape, and since the dielectric block 15 is a single dielectric resonator, the antenna modes tend to be distributed in the middle of the elongated dielectric block 15. In fact, in Figure 1c Several modes at different frequencies can be seen in the antenna. For example... Figure 8 As shown, Figure 1c An ideal effect of this antenna is that the exiting multimodes extend the impedance bandwidth, but at the same time, they also lead to unstable radiation performance. Furthermore, Figure 8 Showing Figure 1c The simulation results for |S11| and gain of the antenna in the horizontal plane with different sizes of the dielectric block 15 are presented. It can be seen that when the size of the dielectric block 15 is 6×2 mm, the antenna has the best |S11| compared to the other two listed sizes. On the other hand, since each size has good and bad gain in different frequency regions, the gains achieved by the antenna in the three dimensions are close to each other. Nevertheless, it was found that the width of the dielectric block 15 cannot be used to adjust the resonant frequency of the antenna because its length is twice or more than the width (e.g., ...). Figure 8 As shown), therefore, the length has a much smaller effect on the resonant frequency than the width and height. Therefore, when Figure 1c As the proportion of antennas in a system increases, the situation becomes more complex, making it difficult to optimize the antennas and resulting in highly unstable radiation performance.

[0056] exist Figure 1d and Figure 2 The image shows a 1×2 dielectric antenna 20 according to another embodiment of the present invention. The antenna 20 comprises two... Figure 1c This is a similar antenna element to the one described above. Antenna 20 also comprises three layers: (from bottom to top) a substrate layer 22, a ground layer 24, and a dielectric body 26 serving as a DRA layer. The ground layer 24 is interposed between the dielectric body 26 and the substrate layer 22.

[0057] The substrate 22 contains two microstrip feed lines 32, each for one of the two antenna elements. In this embodiment, the two microstrip feed lines 32 are parallel to each other, and their input ports are located at the same edge of the substrate 22. In particular, as can be seen from Figure 1, each microstrip feed line 32 has an elongated shape and extends from the same edge of the substrate 22 toward the center (not shown) of the substrate 22 in a horizontal plane (i.e., a virtual plane not shown, and parallel to each of the three PCB layers). Furthermore, the two microstrip feed lines 32 are symmetrically arranged in the substrate 22 around a virtual center line (not shown) that extends in the width direction of the substrate 22. The microstrip feed lines 32 are disposed on the bottom surface of the substrate 22 and are planar metal feed lines.

[0058] Ground layer 24 accommodates two coupling slots 34, each corresponding to a respective microstrip feed 32. A pair 30, including the coupling hole / slot 34 and its corresponding microstrip feed 32, is part of the corresponding antenna element. Therefore, due to the... Figure 1d and Figure 2 The antenna system 20 has two pairs of coupling slots 34 and microstrip feed lines 32, thus having two antenna elements. Each of the two coupling slots 34 has a longitudinal direction intersecting its corresponding microstrip feed line 32, although in Figure 2For clarity, the ground layer 24 and the base layer 22 are offset from each other; therefore, the coupling slot 34 is not shown as directly intersecting its corresponding microstrip feed line 32. Figure 1d As best shown in the diagram, each coupling slot 34 has a projection on the horizontal plane that intersects with its corresponding microstrip feed line 32, specifically, the longitudinal direction of the coupling slot 34 forms a right angle with the longitudinal direction of its corresponding microstrip feed line 32. Figure 2 As shown, in terms of physical dimensions, the length of each microstrip feed 32 exceeds half the width of the substrate 22.

[0059] from Figure 1d and Figure 2 As can be seen, the dielectric body 26 has an elongated shape, with its longitudinal direction perpendicular to the direction of the microstrip feed 32. The dielectric body 26 also has a thickness (in the vertical direction in Figure 1), which is greater than the thickness of the ground layer 24 and the substrate layer 22. In one exemplary embodiment, the dielectric body 26 is made of Rogers... TM RO3006 TM It is made of laminate with a thickness of 1.28 mm, while the base layer 22 is made of Rogers. TM RO4003C M It is made of laminate with a thickness of 0.4 mm. In other words, antenna 20 can be manufactured using standard PCB (printed circuit board) manufacturing technology.

[0060] Antenna 20 further includes a perturbation structure located on top of dielectric body 26 for severing connections between higher-order modes during longitudinal resonant operation of antenna 20. Specifically, the perturbation structure includes three metal lines 28 disposed on top of dielectric body 26. These metal lines 28 are parallel to each other and orthogonal to the longitudinal direction of dielectric body 26. In other words, the metal lines 28 are parallel to the microstrip feed line 32 in substrate layer 22. The metal lines 28 extend along most of the width of dielectric body 26 and are equidistant from each other. The metal lines 28 can be viewed as boundaries of antenna elements, or as “segmenting” dielectric body 26 to form segmented dielectric resonator antennas. As described above, the three metal lines 28 define the boundaries of two antenna elements, which border each other at the middle of the three metal lines 28. It can be seen that the projection of the region of dielectric body 26 between any two adjacent metal lines 28 in the horizontal plane overlaps with the projections of the corresponding coupling slot 34 and the corresponding microstrip feed line 32.

[0061] Now let's look at the working principle of antenna 20. As you can see, it is similar to... Figure 1c Compared to the antenna in the previous model, the main difference in antenna 20 lies in the metal wire 28. The addition of the metal wire 28 is to improve... Figure 1cThe radiation performance of the dielectric resonator in the antenna. The use of interconnected structures helps reduce the size of the antenna. When the antenna frequency increases to millimeter waves, the size of the resonator antenna needs to be reduced, which is difficult to fabricate as a three-dimensional structure. Therefore, interconnected structures are necessary for forming leakage wave antennas.

[0062] However, the interconnected structure leads to the appearance of many higher-order modes in the resonant region (note, not the traveling wave / leakage wave region), which cannot provide stable radiation performance. This is why a metal line 28 is added to the top of the interconnected structure to form antenna 20. Since the profile of the DRA is small enough, the printed metal line 28 can be considered an electric wall, thus disrupting the connection between higher-order modes and generating a standing wave distribution of the electric field at the designed frequency, much like a classic split antenna structure. On the other hand, the elongated shape of the rectangular dielectric body 26 provides the feasibility of exciting a traveling wave distribution within the dielectric body 26. The metal line 28 acts as a perturbation structure, allowing the traveling wave to function as a leaky wave radiation in the dielectric rod antenna. Therefore, in antenna 20, the designs of both the traveling wave antenna and the standing wave antenna are implemented within the same interconnected structure.

[0063] Antenna 20 enables the use of multiple frequencies with a wide scanning angle range for checking fast sensing based on leakage radiation. This significantly improves the time efficiency of ISAC systems using antenna 20, as communication speed in ISAC systems is improved when fast sensing methods are available. Antenna 20 outperforms conventional antenna networks in terms of frequency scanning and beam manipulation, including directivity, steering, and segmentation. For example, in newer antenna networks, operators can use the same MIMO antenna array at base stations for both sensing and communication. However, in these networks, the signal for communication beamforming needs to wait for a period of time from the discretely checked sensing signal. A longer check time can provide more accurate information for communication beamforming, but it introduces more latency into the system. Simply put, in conventional technologies, no antenna can simultaneously satisfy the performance requirements of both communication and sensing signals because the coding, hardware requirements, etc., for sensing and communication functions are completely different. It is not possible to change the function of a specific antenna by changing the beam.

[0064] Let's take a look Figure 3 It shows a 1×8 dielectric antenna 120 according to another embodiment of the present invention, whose element design is similar to... Figure 1d and 2 The antennas in this example are the same. Here, unit design refers to the structure of a single antenna element, which is... Figure 3 Antenna 120 and Figure 1d and 2Antenna 20 is the same. The only difference between the two antennas 20 and 120 is the number of antenna elements, which also determines the difference in their lengths. In fact, for Figure 3 For antenna 120 in the middle, the traveling wave needs a much longer length than the resonant wavelength to obtain higher directivity or gain performance, which can achieve higher sensing resolution. So for the traveling wave, it actually has only one single element with two ports (consisting of eight separate antenna elements), while for the resonant part, it has eight antenna elements.

[0065] Antenna 120 also has a three-layer structure, including a substrate 122, a ground layer 124, and a dielectric body 126. For each of the eight antenna elements, two printed metal lines 128 are disposed on top of the dielectric body 126, a coupling slot 134 is present in the ground layer 124, and a microstrip feed line 132 is present in the substrate 122. In one embodiment, the dielectric body 126 also uses Rogers... TM RO3006 TM The antenna 120 is made of laminate, with a thickness of 1.28 mm and a width of 6 mm. The length of the dielectric block 126 is not a sensitive parameter. Instead, the spacing between the perturbation structures, i.e., the spacing between the metal lines 128 (7.5 mm in the specific implementation), is a critical parameter that can affect the resonant frequency. In the specific embodiment described above, the width of the metal lines 128 is 0.15 mm. Since the antenna 120 is entirely based on PCB technology, there are virtually no manufacturing errors. At the bottom, the substrate is made of Rogers 4003C with a thickness of 0.305 mm.

[0066] like Figure 4a As shown, antenna 120 provides an impedance bandwidth of 25.3 GHz to 30 GHz, while offering better than -18 dB isolation from adjacent and next-to-adjacent structural ports. Radiation performance is as follows... Figure 4b As shown, the peak gain is 14.9 dBi, and the 3-dB gain bandwidth includes the impedance bandwidth. However, the radiation mode is symmetrical, as... Figures 5a-5c As shown in the full description. Overall, the performance of antenna 120 is acceptable.

[0067] Figure 6 , 7a 7b and 7c conceptually illustrate the working principle of antenna 120, which also applies to... Figure 1d and Figure 2 Antenna 20. The dielectric body 126 has two ports 138 at both ends along its longitudinal direction, such as... Figure 3 As shown, they correspond to Figure 6 Port 1 and port 2 in the example. Figure 6As shown, the sensing signal can utilize port 1, port 2, or both together for a difference mode along a longer dimension of the leaky portion of the antenna. On the other hand, antenna 120 has a communication port, namely... Figure 6 Port 3, shown, is formed by multiple microstrip feed lines 132. Port 3 is used for communication signals that move along a shorter dimension of antenna 120. It should be noted that... Figure 6 The image shows four subports of port 3, but this is for illustrative purposes only, as the number of subports can be any number, such as two (by...). Figure 1d and Figure 2 The two microstrip feeds (represented by 32) or eight (by...) Figure 3 (represented by the eight microstrip feed lines 132 in the diagram). Preferably, the number of sub-ports of port 3 is at least eight, because if the number is too small, the peaks in the sensed signal will be difficult to distinguish, which increases the difficulty of sensing. Figure 6 The E-plane of the LWA (Leaky Wave Antenna) used for sensing signals and the SWA (Standing Wave Antenna) used for communication signals are shown. The transmission directions of the two signals are orthogonal to each other, so the same antenna structure can be used for both functions simultaneously.

[0068] like Figure 6 The concept shown was developed through two processes, such as Figures 7a-7c As shown. The initial purpose was to solve manufacturing problems associated with millimeter-wave DRAs. When the frequency rises to millimeter waves, the size of antenna elements, the distance between each element, and the relative positions of the feed structures between antenna elements become extremely small. Traditional manufacturing techniques cannot guarantee the absence of processing errors. Therefore, interconnected structures were introduced to reduce the size of the dielectric antenna, resulting in… Figure 1c and Figure 1d The interconnected structure shown replaces the basic antenna mode with a higher-order mode through multi-stage feeding. Due to the use of higher-order modes, the overall size of the dielectric antenna is much larger than that of conventional antenna elements. The distance between the two antenna elements and the relative positions of the feeding structures can also be addressed. The biggest remaining challenge is maintaining the radiation mode: how to control the electric field distribution of the higher-order modes. Figures 7a-7c As shown, by adding a row of equidistant perturbation structures (e.g., metal lines), all E-field distributions are kept in the same direction, providing a good radiation mode. After roughly achieving the above objective, it was found that the antenna structure is identical to a leaky antenna based on DIL (dielectric image lines). Therefore, the perturbation structure was optimized to realize a frequency-beam-scanning leaky antenna. Since the electric field distributions of the LWA and SWA sections are orthogonal, it is possible to use resonant modes to realize a phased array for communication and to use a frequency-beam-scanning leaky antenna for sensing.

[0069] Figure 9Antenna 120 is shown (in) Figure 9 The antenna is represented as D in the diagram. Figure 1c The antenna in the array was expanded into a 1×8 antenna array (in Figure 9 The simulated radiation comparison is shown in Figure C. It can be seen that the gain of antenna 120 is relatively uniform in the frequency range of 22GHz to 34GHz, but from... Figure 1c The gain of the antenna extended from the substrate shows a significant drop between 26 GHz and 31 GHz. Therefore, it is demonstrated that adding perturbation structures (such as metal wires) to the substrate contributes to the stability of the antenna radiation performance.

[0070] To further illustrate the perturbation structure, similar to... Figure 1c The fundamental cause of gain instability in antennas is the back electric field of higher-order modes. The presence of this back electric field inhibits radiation across the entire antenna, which is a normal wide-side radiation mode. The key to solving the instability problem is how to reduce and lower the back electric field between different modes. Therefore, a series of metal strips are used to solve this problem, such as... Figure 1d , 2 As shown in Figure 3. For Figure 1d , 2 Compared to the antennas in section 3, all electric fields maintain the same direction throughout the entire operating band, which means they can provide very stable radiation performance. Figure 10a and 10b Showing Figure 1d and 2 The simulation results for the gain and S-parameters of antenna 20 at different lengths of the dielectric body 26 are presented. All simulation results are shifted simultaneously as the distance between the metal lines 28 increases. Due to the increased spacing between the metal lines 28, the size of the dielectric body 26 increases, and the resonator frequency decreases. The isolation effect improves due to the increased distance between the two ports of the leakage section, while the improved gain performance is due to the larger radiating aperture. Figure 1d and 2 The antenna structure in the design allows for easy, rapid, and convenient scaling up of the antenna.

[0071] Some parameter studies were also conducted on the metal wire 28 in the antenna 20. Figure 11a and 11bSimulated S-parameters and gain results for antenna 20 at different widths of each metal line 28 are shown. Among the three widths of 0.15 mm, 0.3 mm, and 0.5 mm, the 0.15 mm width provides the best performance. As the width increases, |S11| changes little, while |S21| becomes lower. Furthermore, the gain also decreases slightly during this increase. Therefore, it can be seen that increasing the width of the metal line 28 is neither very useful nor necessary. Additionally, the length of each metal line 28 is another important parameter. Figure 12a and 12b As shown, the longer the metal wire 28, the better the performance, including higher isolation and gain. Among the three lengths of 3 mm, 4 mm, and 5 mm, the 5 mm length provides the best performance. That is to say, when the length of the metal wire 28 is too small, all the performance changes due to the high coupling.

[0072] Let's take a look Figure 13 It shows antenna 220, which is based on Figure 3 Antenna 120 has two SIW-to-DIL (Substrate Integrated Waveguide-Dielectric Image Line) ports 238, configured at the ends of the rod-shaped dielectric body 226. The SIW-to-DIL ports 238 are added because antenna 120 functions as a DIL-based leaky-wave antenna. The SIW-to-DIL ports 238 provide a traveling-wave mode in the DIL that is orthogonal to the resonant mode. Therefore, SIW-to-DIL ports 238 are equivalent to the two side ports of the transverse traveling-wave antenna defined in antenna 220. SIW-to-DIL ports 238 provide an alternative radiation mode without affecting the resonator's performance, and SIW-to-DIL ports 238 are optional. However, it should be noted that the radiation performance of the leaky portion in antenna 220 is not entirely satisfactory, one possible reason being that the perturbation is too small.

[0073] For the DIL of antenna 220, the transmission performance is acceptable because the electric field and slot are in the same direction. It is well known that radiation efficiency can increase as the dielectric layer height decreases. However, due to the limitation of the dielectric layer by the PCB material, leakage exists from the top surface of the DIL. Therefore, the insertion loss is high. To improve radiation performance, one attempt is to increase the number and size of the perturbation structures. A potential difficulty with this type of antenna is the stop band that appears when the beam is scanned to the wide side. This is a narrow region around the wide side, often called the open-stopband region, where undesirable scanning performance occurs. In practical antennas, this means that the amount of radiated power drops significantly within this narrow angular region, and a large VSWR is encountered.

[0074] Let's take a look Figure 14 It shows antenna 320, which is related to Figure 13 Antenna 220 is similar, but has 16 antenna elements. Antenna 320 is a dual-function antenna, comprising a leaky antenna and a 1×16 wide-side antenna array. As those skilled in the art will understand, an arrangement in which the principal direction of radiation is perpendicular to the array axis and also perpendicular to the plane containing the array elements is called a wide-side antenna. For the wide-side portion, all performance is good except for the presence of a high backleaf. This drawback can be improved by redesigning the feed network. On the other hand, the leaky portion is predicted to have an open stopband phenomenon. The peak gain near 26.5 GHz will decrease by approximately 4 dBi.

[0075] In order to overcome the above-mentioned shortcomings of antenna 220, and Figure 1d , Figure 2 , Figure 3 The drawbacks of the antenna in the model have been considered, and simulations have been performed for different types of disturbance structures. Figures 15a-15f Some of these perturbation structures are shown, in particular Figure 15a Showing the widened metal line 428 (with) Figure 1d , 2 (Compared to the metal wires in 3 and 13). Figure 15b The metal line 528b on the side of the rectangular patch 528a is shown. Figure 15c Multiple parallel metal lines 628 are shown, closely adjacent to each other. Figure 15d It is shown that there are two wider metal lines 728b on each side of a narrower metal line 728a in the center. Figure 15e and Figure 15f The metal lines 1028b and 1128b overlap with the rectangular patches 1028a and 1128a. However, Figures 15a-15d The perturbation structures in the model do not provide a good balance between the leakage and standing wave components. This is because the short distance between the perturbation structures and the resonator frequency increases, while the radiation bandwidth decreases. According to the radiation mode, this decrease is due to the coupling between the DRA and the metal strip, resulting in an undesirable electric field distribution that leads to high sag and low directivity.

[0076] Let's take a look Figure 16 Antenna 820 is generally similar to Figure 13 Similar to antenna 220 in the previous section, for the sake of simplicity, similar structures and components will not be described again here. Instead, only the components in antenna 820 will be described. Figure 13 Different characteristics. Antenna 820 also has a three-layer structure, including a base layer 822, a ground layer 824, and a dielectric body 826, but the perturbation structure in antenna system 820 is different from... Figure 13 The differences.

[0077] Specifically, several rows of air vias 836 are employed near each metal line 828 on the top of the dielectric body 826 to achieve good leakage performance. In each group, the air vias 836 are symmetrically located on both sides of the metal line 828. For example, an enlarged view showing two metal lines 828 and their adjacent air vias 836 is provided. Figure 2 The illustration shows two rows of air vents 836 on each side of a metal wire 828, for a total of nine vents. The row closer to the metal wire 828 has five vents 836, while the row farther from the metal wire 828 has four vents 836. Within each row, the air vents 836 are equidistant from each other. All the air vents 836 have approximately the same diameter.

[0078] and Figure 13 Compared to antenna 220, Figure 16 The perturbation structure in the model achieves a better perturbation effect. Improvement Figure 13 Another method to improve the perturbation effect of the antenna system is to increase the width of the metal line at the top of the dielectric (i.e., the length of the metal line along the longitudinal direction of the dielectric). However, increasing the width of the metal line degrades the performance of the standing wave antenna section. Therefore, as a preferred approach, rows of air vias 836 are used to achieve good leakage performance, and the symmetrical non-metallic structure has little impact on the standing wave section. The air vias also provide perturbation without creating coupling between DRAs. Replacing large air regions with grouped vias / holes makes it easier to manufacture and is also beneficial for open stopband phenomena. Figures 19a-19c As shown, the performance of both LWA and SWA is acceptable. For the leaky section, the open stopband issue has been resolved, making it suitable for providing continuous back-to-front scanning. For the resonant section, it behaves like a conventional DRA array, with >20dB of isolation, and the effects of disturbances appear to extend beyond the operating bandwidth.

[0079] By carefully designing the control parameters of antenna system 820, both leaky and standing wave radiation can be obtained within the same dielectric body 826. A wide-area directional radiation pattern can be generated from the sum of the radiated electric fields of the segmented DRA elements. This indicative radiation pattern can be used for good communication between the base station and the target user. The leaky wave radiation pattern in this antenna provides frequency scanning characteristics, with a wide angular scan over its operating bandwidth. This frequency-scanning radiation pattern can detect or sense moving objects within the network. Therefore, it is possible to achieve wide-angle scanning and a wide frequency range for the sensing signal of an ISAC network. Thus, antenna 820 is suitable for simultaneously generating communication and sensing beams.

[0080] Experimental simulations were conducted by studying different antenna systems with 2 to 8 antenna elements. Figure 17a and 17bThe standing wave radiation performance and leakage wave radiation performance of antenna 820 are shown respectively. From Figure 17a It can be seen that antenna 820 can generate a directional radiation pattern from its leaky mode. Figure 17b It can be seen that the antenna 820 can also generate a nice frequency scanning radiation pattern from the standing wave mode.

[0081] Besides radiation performance, isolation between components is also crucial for antenna systems. When designing an ISAC antenna array, eliminating interference between the two signals of the sensing and communication (S&C) subsystems must be considered. The time-grating method is one of the more recently used solutions in ISACs, providing sufficient isolation for the S&C signals. However, it introduces significant drawbacks to ISAC systems, especially when the number of sensing frames in the system increases to achieve better resolution and accuracy in distinguishing and defining objects. To achieve this, it is recommended to construct a high-isolation antenna array with both leaky and standing-wave modes. It is well known that a good antenna array with dual polarization or dual operating modes must address the mutual coupling between the array's radiating elements. The radiating mode excited along the x-axis originates from its leaky mode, while the standing-wave mode originates from the y-axis excitation. Since these two modes are orthogonal, the mutual coupling between the radiating elements is sufficiently small.

[0082] In another embodiment of the invention, a configuration is thus proposed, comprising establishing two separate power dividers to connect the subarray elements for the S&C port accordingly. Figure 18 Antenna 920 with this configuration is shown, wherein port 1 and port 2 (i.e., two ports 938) are located at both ends of the dielectric rod for leakage wave radiation, becoming the two side ports of the transverse traveling-wave antenna defined in antenna 920, and... Figure 16 Similar to what is shown. However, with Figure 16 Compared to an antenna with eight antenna elements, antenna 920 has sixteen antenna elements, thus forming a 1-16 power divider to excite standing wave radiation. Preliminary results from antenna 920 confirm that port-to-port isolation can produce more than 40 dB over the array's wide operating bandwidth (see...). Figure 19a This high level of isolation provides low-interference S&C signals in ISAC antenna systems between sensing and communication ports. The array operates over a bandwidth from 24 GHz to 32 GHz. Furthermore, the array exhibits very high purity in its radiation modes. Both modes achieve low cross-polarization in the radiation mode (see...). Figure 19b and Figure 19c ).

[0083] Based on promising preliminary results, the following projects were identified to improve the performance of the antenna array, including (1) a study on bandwidth enhancement of the array's antenna elements, (2) mode enhancement to suppress side spheres and back spheres, (3) a feasibility study of a circularly polarized antenna array for indoor ISAC applications, and (4) a study of a MIMO array based on the proposed subarray design. It is proposed to introduce two reconfigurable networks connected to the antenna array to independently control the sensing and communication reconfigurable radiation beams. Preliminary results show that wide-angle scanning range control of the frequency scanning beam can be achieved at the sensing port of this antenna array. The frequency scanning range of a traditional leaky antenna is determined by the fixed structure of the leaky body of the antenna. Additional phase information is applied to the reconfigurable network to achieve dynamic angle scanning control of the leaky radiation modes. Simulation results confirm this effective method for beam manipulation of the leaky modes of the antenna array (see [link to simulation results]). Figure 20 The radiation pattern can be effectively controlled on both horizontal and vertical cut surfaces. For the communication port of this antenna array, another reconfigurable network with amplitude and phase control can be applied to manipulate the beam steering / strapping / tilting of the communication radiation pattern (see...). Figure 21a Beam splitting (see) Figure 21b ), and beam splitting with different beamwidths (see Figure 21c ).exist Figure 21b In the diagram, different curves represent the gain results when different numbers of antenna elements in the antenna are out of phase. For example, "every two flips" means that in the example of 16 antenna elements, the first and second elements have the original phase, but the third and fourth elements have the opposite phase, then the fifth and sixth elements have the original phase again, and so on. Figure 21c In the diagram, different curves represent the gain results when different numbers of antenna elements are excited. For example, "4 elements" means that in an example with 16 antenna elements, only 4 elements are excited, while the other elements are not excited.

[0084] Therefore, exemplary embodiments have been fully described. Although specific embodiments are mentioned in the description, it will be clear to those skilled in the art that the invention can be practiced with variations in these specific details. Therefore, the invention should not be construed as being limited to the embodiments described herein.

[0085] While embodiments of the invention have been detailed and described in the accompanying drawings and foregoing description, they should be considered illustrative rather than restrictive. It is to be understood that only exemplary embodiments are shown and described, and the scope of the invention is not limited in any way. It is understood that any feature described herein can be used in any embodiment. The illustrative embodiments do not exclude each other or other embodiments not mentioned herein. Therefore, the invention also provides embodiments comprising combinations of one or more of the foregoing illustrative embodiments. Modifications and variations can be made to the invention without departing from its spirit and scope; therefore, only the limitations set forth in the appended claims should be applied.

[0086] For example, microstrip feedlines are displayed in various shapes, such as Figure 2 and Figures 15a-15d The present invention is not limited to these feed shapes, as will be understood by those skilled in the art. Conversely, other shapes of microstrip feeds can also be configured in the antenna. The invention is also not limited to the excitation method of the DRA, i.e., slotted feed, conformal feed, probe feed, or any other suitable method can be used.

[0087] In some of the embodiments described above, air vias help improve the traveling wave portion of the antenna with minimal impact on the resonator portion. However, air vias are not the only perturbation structure that can be used. Essentially, air windows can enhance perturbation, so they don't necessarily have to be porous structures. In other variations, it can be a single large rectangular air window.

Claims

1. A dielectric antenna, comprising: basal layer; A ground layer located above the substrate layer; as well as The dielectric body located above the grounding layer; The dielectric body has an elongated shape; the dielectric antenna further includes an electric wall structure located on top of the dielectric body for cutting the connection between higher-order modes in the resonant portion of the dielectric antenna; wherein the electric wall structure includes a metal wire disposed on the top of the dielectric body; and The dielectric body further includes a first coplanar waveguide feed port located at a first end of the dielectric body and a second coplanar waveguide feed port located at a second end of the dielectric body; the first coplanar waveguide feed port and the second coplanar waveguide feed port are adapted to feed together a plurality of antenna elements defined in the dielectric body; the plurality of antenna elements constitute a leaky wave antenna.

2. The dielectric antenna according to claim 1, wherein the metal wire is substantially perpendicular to the longitudinal direction of the dielectric body.

3. The dielectric antenna according to claim 1, wherein the electric wall structure further includes a plurality of air holes on the dielectric body, the air holes being adjacent to the metal wire.

4. The dielectric antenna according to claim 3, wherein the plurality of air holes are symmetrically located on both sides of the metal wire.

5. The dielectric antenna according to claim 1, wherein the electric wall structure comprises a plurality of said metal lines, which are parallel to each other and perpendicular to the longitudinal direction of the dielectric body.

6. The dielectric antenna according to claim 5, wherein the plurality of metal lines are equidistant from each other.

7. The dielectric antenna of claim 6, wherein each of the plurality of antenna elements corresponds to one of the plurality of metal lines; the plurality of antenna elements are adjacent to each other at one of the plurality of metal lines.

8. The dielectric antenna according to claim 1, wherein the substrate layer includes a first microstrip feed line and a second microstrip feed line; the ground layer includes a first coupling slot whose longitudinal direction intersects the first microstrip feed line, and a second coupling slot whose longitudinal direction intersects the second microstrip feed line.

9. The dielectric antenna according to claim 8, wherein the first microstrip feed line and the second microstrip feed line correspond to the first antenna element and the second antenna element among the plurality of antenna elements, respectively; the first antenna element and the second antenna element are adjacent to each other at the metal wire.

10. The dielectric antenna according to claim 1, wherein the plurality of antenna elements further constitute a standing wave antenna.

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