A harmonic suppression and consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine

Abstract

This invention relates to a harmonic suppression and power consumption reduction control method for low-wind-speed permanent magnet direct-drive wind turbines, belonging to the field of wind power. It employs a Vienna rectifier as the turbine-side converter and uses a sensorless algorithm based on a sliding mode observer to accurately acquire position data. d The shaft position angle employs a control strategy that combines proportional-integral resonant controller, carrier pulse width modulation technology based on zero-sequence current injection and midpoint potential balance, and active damping control with capacitor current feedback and quasi-proportional resonant control to suppress harmonics. This achieves optimal control for multiple objectives, including reducing stator current and grid current harmonic content, minimizing converter losses, and suppressing midpoint potential oscillations. Simultaneously, it reduces the size and weight of the filter inductor, thus ensuring effective control and efficient operation of low-wind-speed permanent magnet direct-drive wind turbine units. This invention achieves the goals of low power consumption, lightweight design, miniaturization, low cost, and high efficiency for low-wind-speed permanent magnet direct-drive wind turbine units, making it particularly suitable for such units.

Description

Technical Field

[0001] This invention relates to a control method, and more particularly to a harmonic suppression and energy consumption reduction control method for low-wind-speed permanent magnet direct-drive wind turbine generators, belonging to the field of wind power technology. Background Technology

[0002] Permanent magnet synchronous wind turbines (PMSGs) have been widely used in vertical axis and horizontal axis wind power generation systems. Their converter system consists of a machine-side converter (MSC) and a grid-side converter (GSC).

[0003] Low-wind-speed wind power has become one of the key wind power development areas in my country. For low-wind-speed power generation, low losses in the entire wind turbine are essential; therefore, reducing the losses of the wind turbine generator and its converter is particularly necessary and also a challenge. On the other hand, low-wind-speed wind farms are characterized by strong turbulence, variable wind direction, and large wind speed fluctuations, often causing uncertainty in external interference to the wind turbine generator. Furthermore, long-term operation of the wind turbine generator can lead to changes in its own parameters. All of these factors pose significant challenges to the control of low-wind-speed permanent magnet direct-drive wind turbine generators. Therefore, achieving stable control of low-wind-speed wind turbine generators and effectively reducing PMSG stator current harmonics and converter losses are crucial measures for achieving efficient operation of low-wind-speed wind turbine generators.

[0004] Harmonics in low-wind-speed permanent magnet direct-drive wind turbines mainly originate from the PMSG (Generator-Modulated Generator Switch) and the converter. Harmonics in the PMSG are caused by the generator's own structure and nonlinear devices in the generator-side converter. The magnetic field generated by the harmonic current inside the generator interacts with the fundamental current's magnetic field, producing pulsating torque that affects generator rotation, increasing generator losses and reducing output power. The permanent magnets heat up due to eddy currents, and in severe cases, may permanently lose their magnetism. Furthermore, due to grid voltage imbalances and the influence of nonlinear devices in the grid-side converter, a large number of harmonics also exist in the grid-connected current.

[0005] For machine-side converters, when using rotor field-oriented vector control, a coordinate transformation of the stator current is required to obtain the dq-axis current. Generally, a zero d-axis current (ZDC) control strategy is used to control the reactive current, making i d =0. However, when the permanent magnet adopts an asymmetrical shape and non-traditional radial magnetization, the direction of the strongest magnetic field of the permanent magnet is not necessarily the N pole direction. Considering the permanent magnet installation process, the d-axis position angle cannot be simply represented by multiplying the rotor permanent magnet position angle obtained by the position sensor by the number of pole pairs. Therefore, if a position sensor is used, it will cause the reactive current i d If the current is not equal to 0, the stator current and voltage cannot operate in phase, resulting in more harmonics and more losses.

[0006] Currently, the MSC topology of permanent magnet direct-drive wind turbines is divided into VSC and Vienna rectifiers. Vienna rectifiers are gaining increasing attention in the wind power field due to their advantages such as low current harmonics, low voltage stress on power devices, no dead time, and low losses. However, Vienna rectifiers require balanced neutral point voltage, and as wind speed increases, the neutral point voltage imbalance becomes increasingly severe. If not properly controlled, this imbalance will eventually damage the Vienna rectifier.

[0007] For permanent magnet direct-drive wind turbines (PGVRs) using VIENNA rectifiers as the MSC, commonly used control methods include PID control and model predictive control (MPC). PID control offers high reliability and ease of programming, but due to its fixed parameters, it is difficult to achieve optimal control performance under different wind speeds. MPC optimizes the performance of PGVR units by designing appropriate cost functions. However, MPC selects the next action of the bidirectional switch by comparing the cost of different switching states; therefore, MPC requires modulation capabilities, necessitating hardware with high computational speed.

[0008] For grid-side converters (GSCs), conventional LCL filters are large, bulky, and have high losses. Traditional active damping control uses control algorithms to increase system damping without using additional components, thus eliminating additional losses. However, it requires sensors to assist in realizing the function of the damping resistor and may also cause problems such as reduced phase margin and steady-state error. Therefore, it is necessary to study improved active damping control strategies to effectively solve the above problems and achieve lightweight, miniaturized, and low-power GSCs. Summary of the Invention

[0009] The main objective of this invention is to address the shortcomings of existing technologies by providing a harmonic suppression and power consumption reduction control method for low-wind-speed permanent magnet direct-drive wind turbines. This method employs a Vienna rectifier as the turbine-side converter, uses a sensorless algorithm based on a sliding mode observer to accurately obtain the d-axis position angle, and utilizes a proportional-integral resonant controller, carrier modulation technology based on zero-sequence current injection and midpoint potential balance, and an improved active damping control strategy to suppress harmonics. This achieves optimal control across multiple objectives, including reducing stator and grid-connected current harmonic content, minimizing converter losses, and suppressing midpoint potential oscillations. Ultimately, this ensures effective control and efficient operation of the low-wind-speed permanent magnet direct-drive wind turbine, minimizing its power consumption.

[0010] To achieve the above objectives, the low-wind-speed permanent magnet direct-drive wind turbine generator of the present invention includes: a permanent magnet synchronous wind turbine generator, a machine-side converter, a grid-side converter, etc.; one end of the machine-side converter is connected to the stator of the permanent magnet synchronous wind turbine generator, and the other end is connected to the grid-side converter; the grid-side converter is connected to the power grid through a power frequency transformer.

[0011] The generator-side converter uses a Vienna rectifier, which includes three filter inductors, three bidirectional switching units, and two output capacitors. One end of each of the three filter inductors is connected to the three-phase stator windings of the permanent magnet synchronous wind turbine. The two output capacitors are connected in series. The output capacitor connected to the positive terminal of the Vienna rectifier's DC output is called the upper capacitor, and the output capacitor connected to the negative terminal of the Vienna rectifier's DC output is called the lower capacitor. The grid-side converter is a voltage source converter, which includes an LCL filter.

[0012] This invention discloses a harmonic suppression and power consumption reduction control method for low-wind-speed permanent magnet direct-drive wind turbine generators, comprising the following steps:

[0013] Step 1: Obtain the d-axis position angle θ using a sensorless algorithm based on a sliding mode observer. se The specific process is as follows:

[0014] 11) The stator output of the permanent magnet synchronous wind turbine generator is a three-phase alternating current with variable frequency and amplitude, the voltage of which is denoted as u. a u b u c Their currents are denoted as i a i b i c ; will u a u b u c u is obtained through abc / αβ coordinate transformation α and u β ;change i a i b i c i is obtained through abc / αβ coordinate transformation α and i β ;

[0015] 12) The voltage equation of the permanent magnet synchronous wind turbine in the αβ coordinate system is:

[0016]

[0017] In the formula, L s For stator inductance; R s ω is the stator resistance; e ψ is the electric angular velocity of the permanent magnet synchronous wind turbine. f θ is the rotor permanent magnet flux linkage of the permanent magnet synchronous wind turbine generator; se Let be the d-axis phase angle, and have

[0018]

[0019] In the formula, e sα e sβ These are the α-axis and β-axis components of the back electromotive force, respectively;

[0020] 13) To facilitate the use of a sliding mode observer to estimate the back electromotive force, the voltage equation in equation (1) is rewritten in the form of a current state equation:

[0021]

[0022] 14) To obtain the back electromotive force e sα e sβ The estimated value e sα ,e、e sβ,e The sliding mode observer is designed as follows:

[0023]

[0024] In the formula, i α,e i β,e The stator current i output by the sliding mode observer is respectively α i β The estimated value; u α u β is the input value of the sliding mode observer.

[0025] 15) Subtracting equations (3) and (4), we obtain the stator current error equation as follows:

[0026]

[0027] 16) The sliding mode control rate is designed as follows:

[0028]

[0029] In the formula, sgn() is the sign function; k is the maximum amplitude of the generator's no-load back electromotive force, and satisfies:

[0030]

[0031] 17) The sliding surface is designed as (i α -i α,e ), (i β -sβ,e), by rapidly switching between positive and negative sliding surfaces, reduces the back electromotive force error (e) sα -e sα,e ), (e sβ -e sβ,e It quickly converges to 0, at which point we can obtain:

[0032]

[0033] In actual operation e sα,e esβ,e It is a discontinuous high-frequency switching signal. To obtain a continuous waveform, an external low-pass filter is required, which is expressed as:

[0034]

[0035] In the formula, τ0 is the time constant of the low-pass filter;

[0036] 18) In obtaining e sα e sβ Then, θ is obtained using the arctangent function method. se That is, it can be expressed as:

[0037]

[0038] Filtering will cause phase delay. Angle compensation is added to equation (10), which is expressed as:

[0039]

[0040] In the formula, ω c Let ω be the cutoff frequency of the low-pass filter. c =1 / τ0;

[0041] Step 2: For the machine-side converter, a proportional-integral resonant controller with a variable resonant angular frequency is used to suppress the specified harmonics; the specific process is as follows:

[0042] 21) Based on the electric angular velocity ω of the permanent magnet synchronous wind turbine generator e The fundamental frequency of the stator current is obtained, and then the stator current i is... s Perform Fast Fourier Decomposition and calculate the current distortion rate (THD) according to Equation (12):

[0043]

[0044] In the formula, I1 is the amplitude of the fundamental current, I k Let be the amplitude of the kth harmonic current.

[0045] 22) The stator voltage and current of the permanent magnet synchronous wind turbine generator are obtained by abc / dq coordinate transformation to obtain u respectively. d u q and i d i q ;

[0046] 23) Set a reference value for THD. * =0, the difference between this reference value and its actual value is input to the first PI controller, and the output of the first PI controller is equal to T. e * / (1.5pψ fAdd them together to get i q Reference value i q * T e * The electromagnetic torque T of the permanent magnet synchronous wind turbine generator e Reference value;

[0047] 24) Set the reference value i of the stator current. q * Its actual value i q The difference is input to the first proportional-integral resonant controller, and the output of the first proportional-integral resonant controller is decoupled from the q-axis feedforward decoupling compensation term -ω. e L s i d +ω e ψ f By adding them together, we obtain the q-axis component u of the stator voltage of the permanent magnet synchronous wind turbine. q Reference value u q * ;

[0048] The transfer function of the first proportional-integral resonant controller is:

[0049]

[0050] In the formula, K p K is the proportionality coefficient. i ω is the integral coefficient; cs The controller bandwidth is given by h; the harmonic order is given by K. h is the h-th harmonic resonant gain coefficient.

[0051] 25) Let the d-axis component i of the stator current of the permanent magnet synchronous wind turbine be... d Reference value i d * =0, and compare it with its actual value i d The second proportional-integral resonant controller is used as the input, and the output of the second proportional-integral resonant controller is decoupled from the d-axis feedforward decoupling compensation term ω. e L s i q By adding them together, we obtain the d-axis component u of the stator voltage of the permanent magnet synchronous wind turbine. d Reference value u d * ;

[0052] The transfer function of the second proportional-integral resonant controller is the same as that of the first proportional-integral resonant controller.

[0053] Step 3: The Vienna rectifier is controlled using carrier pulse width modulation technology based on zero-sequence component injection and midpoint potential balance; the specific process is as follows:

[0054] 31) The stator voltage reference value u obtained in steps 24) and 25) q * and u d * The voltage u is obtained after dq / abc transformation. a u b u c Reference value u a * u b * u c * Let the DC side voltage U of the Vienna rectifier be... dc Half of it is used as the baseline value, i.e., U dc / 2, for u a * u b * u c * Standardize to obtain the per-unit value u. ap * u bp * u cp * Find u ap * u bp * u cp * The maximum and minimum values ​​in the range are denoted as u. pmax u pmin ;

[0055] 32) Define the zero-order component as u0:

[0056] u0 = -0.5(u pmax +u pmin (14)

[0057] 33) Inject the zero-sequence components into u respectively. ap * u bp * u cp * , to obtain m ap * m bp * m cp * Find m ap* m bp * m cp * The maximum and minimum values ​​in the range are denoted as m. pmax m pmin ;

[0058] 34) Measure the voltage U1 across the upper capacitor and the voltage U2 across the lower capacitor of the Vienna rectifier; the difference between the two is input to the second PI controller, and the output of the second PI controller is recorded as the suppression factor z;

[0059] 35) The inhibition factor z is limited by a limiter, with an upper limit of 1-m. pmax Its lower limit is -1-m pmin ;

[0060] 36) The suppression factor z after the limiter is compared with the m obtained in step 33). ap * m bp * m cp * Add them together to obtain the corrected voltage reference value u. am * u bm * u cm * ;

[0061] 37) The corrected voltage reference value u am * u bm * u cm * The drive signals for the three bidirectional switching units of the Vienna rectifier are obtained by carrier pulse width modulation, and the three bidirectional switching units are driven to work.

[0062] Step 4: The grid-side converter employs a control strategy combining active damping control with capacitor current feedback and quasi-proportional resonant control to filter the grid-side current; the specific process is as follows:

[0063] 41) Quasi-resonant control is adopted in the outer loop: the three-phase current value that meets the grid connection requirements is used as the reference value i of the grid-side three-phase current. * gabc The control objective is to make the filter output current in phase with the grid voltage, that is:

[0064] The reference current i * gabc i is obtained through abc / αβ coordinate transformation gα * and igβ * , respectively with their actual current i gα and i gβ After the difference is calculated, the results are input to the first quasi-proportional resonant controller and the second quasi-proportional resonant controller, respectively.

[0065] The transfer function of the quasi-proportional resonant (QPR) controller is

[0066]

[0067] In the formula, K p K is the proportional gain coefficient; r ω is the resonant gain coefficient. cg ω is the controller bandwidth; res This refers to the harmonic frequency.

[0068] 42) The inner loop uses active damping control with capacitor current feedback: the capacitor branch current i in the LCL filter circuit is controlled by... Cabc i is obtained after abc / αβ coordinate transformation Cα and i Cβ Step 41) The output of the first quasi-proportional resonant controller and i Cα The difference is calculated, and the grid-connected voltage reference value u is obtained through the third PI controller. gα * Step 41) The output of the second quasi-proportional resonant controller and i Cβ The difference is calculated, and the grid-connected voltage reference value u is obtained through the fourth PI controller. gβ * ;

[0069] 43) will u gα * and u gβ * The SVPWM module generates a drive signal to drive the power switch of the grid-side converter and control the operation of the grid-side converter.

[0070] Compared with the prior art, the beneficial effects of the present invention are:

[0071] 1. By using Vienna rectifiers as the generator-side converters and employing sensorless algorithms, proportional-integral resonant controllers, and midpoint potential balance control methods, optimal control for multiple objectives was achieved, improving power quality, effectively suppressing stator current harmonic distortion of permanent magnet synchronous wind turbines, and reducing power losses in wind turbines and generator-side converters.

[0072] 2. The control strategy that combines active damping control with capacitor current feedback and quasi-proportional resonant control can not only greatly reduce the total harmonic distortion rate of the grid-connected current, but also allow for the use of smaller filter inductors, thereby achieving lightweight and miniaturized filters, and further reducing the cost and power loss of wind turbine units.

[0073] 3. It can achieve the goals of low power consumption, lightweight, miniaturization, low cost, and high efficiency of low wind speed permanent magnet direct drive wind turbine units. Attached Figure Description

[0074] Figure 1 This is a schematic diagram of the low wind speed permanent magnet direct drive wind turbine system described in this invention.

[0075] Figure 2 This is a schematic diagram of the topology of the machine-side converter described in this invention, which is a Vienna rectifier.

[0076] Figure 3 This is a schematic diagram illustrating the energy-saving control principle of the low-wind-speed permanent magnet direct-drive wind turbine converter described in this invention.

[0077] Figure 4 This is a schematic diagram of the sensorless algorithm control structure based on sliding mode observer described in this invention.

[0078] Figure 5 This is a schematic diagram of the control structure of the carrier pulse width modulation technology based on zero-sequence component injection and midpoint potential balance described in this invention.

[0079] Figure 6 This invention presents the simulated stator current waveform and its distortion rate at rated wind speed when using a Vienna rectifier and a traditional VSC as the machine-side converter.

[0080] Figure 7 This is an experimental graph showing the power generation efficiency curve when the present invention uses a Vienna rectifier and a traditional VSC as the generator-side converter.

[0081] Figure 8 This is a comparison diagram of the three-phase grid-connected current waveforms under the improved active damping method of the grid-side converter of this invention and the traditional voltage-oriented control strategy.

[0082] Figure 9 This is a comparison chart of the current distortion rate (THD) under the improved active damping method of the grid-side converter of this invention and the traditional voltage-oriented control strategy.

[0083] Figure 10 This is a comparison diagram of the three-phase grid-connected current waveforms under the improved active damping method of the grid-side converter of this invention and the traditional voltage-oriented control strategy.

[0084] Figure 11This is a comparison chart of the current distortion rate (THD) under the improved active damping method of the grid-side converter of this invention and the traditional voltage-oriented control strategy.

[0085] Among them, 1-Permanent magnet synchronous wind turbine, 2-Machine-side converter, 3-Grid-side converter, 4-Sensorless algorithm module, 5-First PI controller, 6-First proportional-integral-resonant (PI-RES) controller, 7-Second proportional-integral-resonant (PI-RES) controller, 8-Carrier pulse width modulation module based on zero-sequence component and midpoint potential balance, 9-Second PI controller, 10-First quasi-proportional-resonant (QPR) controller, 11-Second quasi-proportional-resonant (QPR) controller, 12-Third PI controller, 13-Fourth PI controller. Detailed Implementation

[0086] The present invention will now be described in further detail with reference to the accompanying drawings.

[0087] like Figure 1 As shown, the low wind speed permanent magnet direct drive wind turbine of the present invention includes a permanent magnet synchronous wind turbine generator 1, a machine-side converter 2, and a grid-side converter 3. One end of the machine-side converter 2 is connected to the stator of the permanent magnet synchronous wind turbine generator 1, and the other end is connected to the grid-side converter 3. The grid-side converter 3 is connected to the AC power grid through a power frequency transformer.

[0088] like Figure 2 As shown, the machine-side converter 2 is a Vienna rectifier, which includes three filter inductors (L... a L b L c ), three bidirectional switching units (S a S b S c Two output capacitors (C1, C2); one end of each of the three filter inductors is connected to the three-phase stator windings of the permanent magnet synchronous wind turbine generator 1; the two output capacitors are connected in series, wherein the output capacitor C1, which is connected to the positive terminal of the DC output of the Vienna rectifier, is called the upper capacitor, and the output capacitor C2, which is connected to the negative terminal of the DC output of the Vienna rectifier, is called the lower capacitor.

[0089] like Figure 3 As shown, grid-side converter 3 is a voltage source converter, which includes an LCL filter, consisting of a filter inductor L on the grid-side converter 3 (GSC) side, a filter capacitor C, and a grid-side filter inductor L... g constitute.

[0090] like Figure 3 As shown, the present invention provides a harmonic suppression and power consumption reduction control method for low-wind-speed permanent magnet direct-drive wind turbine generators, comprising the following steps:

[0091] Step 1: Obtain the d-axis position angle θ using a sensorless algorithm based on a sliding mode observer. se ;like Figure 3 , Figure 4 As shown, the specific process is as follows:

[0092] 11) The stator output of the permanent magnet synchronous wind turbine generator 1 is a three-phase alternating current with variable frequency and amplitude, and its voltage is denoted as u. a u b u c Their currents are denoted as i a i b i c ; will u a u b u c u is obtained through abc / αβ coordinate transformation α and u β ;change i a i b i c i is obtained through abc / αβ coordinate transformation α and i β ; will u α u β and i α i β The data is fed into sensorless algorithm module 4, which outputs the d-axis position angle θ. se The sensorless algorithm is as follows:

[0093] 12) The voltage equation of permanent magnet synchronous wind turbine generator 1 in the αβ coordinate system is:

[0094]

[0095] In the formula, L s For stator inductance; R s ω is the stator resistance; e Let ω be the electric angular velocity of the permanent magnet synchronous wind turbine generator 1. e =pω, where ω is the mechanical angular velocity measured by the encoder, and p is the number of pole pairs of permanent magnet synchronous wind turbine generator 1; ψ f For the rotor permanent magnet flux linkage of permanent magnet synchronous wind turbine generator 1; θ se Let be the d-axis phase angle, and have

[0096]

[0097] In the formula, e sα e sβ These are the α-axis and β-axis components of the back electromotive force, respectively;

[0098] 13) To facilitate the use of a sliding mode observer (SMO) to estimate the back electromotive force, the voltage equation in equation (1) is rewritten in the form of a current state equation:

[0099]

[0100] 14) To obtain the back electromotive force e sα e sβ The estimated value e sα,e e sβ,e The SMO is designed as follows:

[0101]

[0102] In the formula, i α,e i β,e The stator current i output by the SMO is respectively α i β The estimated value; u α u β For SMO input values.

[0103] 15) Subtracting equations (3) and (4), we obtain the stator current error equation as follows:

[0104]

[0105] 16) The sliding mode control rate is designed as follows:

[0106]

[0107] In the formula, sgn() is the sign function; k is the maximum amplitude of the generator's no-load back electromotive force, and satisfies:

[0108] k>max{e sα ,e sβ} (7)

[0109] 17) The sliding surface is designed as (i α -i α,e ), (i β - sβ,e By switching rapidly between positive and negative sliding surfaces, the back electromotive force error (e) is reduced. sα -e sα,e ), (e sβ -e sβ,e It quickly converges to 0, at which point we can obtain:

[0110]

[0111] In actual operation e sα,e e sβ,eIt is a discontinuous high-frequency switching signal. To obtain a continuous waveform, an external low-pass filter is required, which is expressed as:

[0112]

[0113] In the formula, τ0 is the time constant of the low-pass filter;

[0114] 18) In obtaining e sα e sβ Then, θ is obtained using the arctangent function method. se That is, it can be expressed as:

[0115]

[0116] Filtering will cause phase delay. Angle compensation is added to equation (10), which is expressed as:

[0117]

[0118] In the formula, ω c Let ω be the cutoff frequency of the low-pass filter. c =1 / τ0;

[0119] Step 2: A proportional-integral resonant controller with a variable resonant angular frequency is used to suppress the specified harmonics of the machine-side converter 2; the specific process is as follows:

[0120] 21) Based on the electric angular velocity ω of the permanent magnet synchronous wind turbine generator 1 e The fundamental frequency of the stator current is obtained, and then the stator current i is... s Perform Fast Fourier Decomposition and calculate the current distortion rate (THD) according to Equation (12):

[0121]

[0122] In the formula, I1 is the amplitude of the fundamental current, I k Let be the amplitude of the kth harmonic current.

[0123] 22) The stator voltage and current of permanent magnet synchronous wind turbine generator 1 are obtained by abc / dq coordinate transformation to obtain u respectively. d u q and i d i q ;

[0124] 23) Set a reference value for THD. * =0, take the difference between this reference value and its actual value and input it into the first PI controller 5. Its output is equal to T. e * / (1.5pψ f Add them together to get i q Reference value iq * T e * The electromagnetic torque T of the permanent magnet synchronous wind turbine generator 1 e The reference value, T e * =P * / ω,P * The output power reference value of the permanent magnet synchronous wind turbine generator 1 can be determined based on the maximum power point (MPP) curve of the wind turbine generator set, or based on the output power command of the wind farm.

[0125] 24) Set the reference value i of the stator current. q * Its actual value i q The difference is calculated by inputting the first proportional-integral-resonant (PI-RES) controller 6, whose output is decoupled from the q-axis feedforward decoupling compensation term -ω. e L s i d +ω e ψ f Adding them together, we obtain the q-axis component u of the stator voltage of the permanent magnet synchronous wind turbine generator 1. q Reference value u q * ;

[0126] The transfer function of the first proportional-integral resonant controller 6 is:

[0127]

[0128] In the formula, K p K is the proportionality coefficient. i ω is the integral coefficient; cs The controller bandwidth is given by h; the harmonic order is given by K. h is the h-th harmonic resonant gain coefficient.

[0129] 25) Let the d-axis component i of the stator current of permanent magnet synchronous wind turbine generator 1 be... d Reference value i d * =0, and compare it with its actual value i d The differential input second proportional-integral resonant controller 7, whose output is decoupled from the d-axis feedforward decoupling compensation term ω e L s i q Adding them together, we obtain the d-axis component u of the stator voltage of permanent magnet synchronous wind turbine generator 1. d Reference value u d * ;

[0130] The transfer function of the second proportional-integral resonant controller 7 is the same as that of the first proportional-integral resonant controller 6.

[0131] Step 3, put u d * u q * The input is a carrier modulation module 8 based on zero-sequence component and midpoint potential balance. It uses carrier pulse width modulation (CBPWM) technology based on zero-sequence component injection and midpoint potential balance to control the Vienna rectifier of the machine-side converter 2. The specific steps are as follows: Figure 5 As shown,

[0132] 31) The stator voltage reference value u obtained in steps 24) and 25) q * and u d * The voltage u is obtained after dq / abc transformation. a u b u c Reference value u a * u b * u c * This causes the DC-side voltage U of the Vienna rectifier to... dc Half of it is used as the baseline value, i.e., U dc / 2, for u a * u b * u c * Standardize to obtain the per-unit value u. ap * u bp * u cp * Find u ap * u bp * u cp * The maximum and minimum values ​​in the range are denoted as u. pmax u pmin ;

[0133] 32) Define the zero-order component as u0:

[0134] u0 = -0.5(u pmax +u pmin (14)

[0135] 33) Inject the zero-sequence component u0 into u respectively ap* u bp * u cp * , to obtain m ap * m bp * m cp * Find m ap * m bp * m cp * The maximum and minimum values ​​in the range are denoted as m. pmax m pmin ;

[0136] 34) Measure the voltage U1 across the upper capacitor and the voltage U2 across the lower capacitor of the Vienna rectifier 2; the difference between the two is input to the second PI controller 9, and the output of the second PI controller 9 is denoted as the suppression factor z:

[0137]

[0138] In the formula, K zP K zI These represent the proportional and integral parameters of the midpoint voltage balance loop, respectively.

[0139] 35) The inhibition factor z is limited by a limiter, with an upper limit of 1-m. pmax Its lower limit is -1-m pmin ;

[0140] 36) The suppression factor z after the limiter is compared with the m obtained in step 33). ap * m bp * m cp * Add them together to obtain the corrected voltage reference value u. am * u bm * u cm * The role of the inhibitor z is to make m ap * m bp * m cp * The vertical translation adjusts the duration of the P-type or N-type small vector within the switching cycle, thereby effectively balancing U1 and U2 and reducing the fluctuation of the midpoint potential U0 (=U2-U1).

[0141] 37) The corrected voltage reference value u am * u bm * u cm * The drive signals s of the three bidirectional switching units of the Vienna rectifier are obtained through carrier pulse width modulation (CBPWM). a s b s c It drives three bidirectional switching units to operate and controls the permanent magnet synchronous wind turbine generator 1 to operate.

[0142] Step 4: The grid-side converter 3 adopts a control strategy combining active damping control with capacitor current feedback and quasi-proportional resonant control. The outer loop uses quasi-resonant PR control, and the inner loop uses active damping control with capacitor current feedback to filter the grid-side current.

[0143] 41) The outer loop adopts quasi-resonant PR control: the three-phase current that meets the grid connection requirements is used as the reference value i of the grid-side three-phase current. * gabc (i.e. i * ga i * gb i * gc The control objective is to make the filter output current in phase with the grid voltage, that is:

[0144] The reference current i * gabc i is obtained through abc / αβ coordinate transformation gα * and i gβ * , respectively with their actual current i gα and i gβ After the difference is calculated, the results are input to the first quasi-proportional resonance (QPR) controller 10 and the second quasi-proportional resonance (QPR) controller 11, respectively.

[0145] The transfer functions of the first QPR controller 10 and the second QPR controller 11 are both:

[0146]

[0147] In the formula, K p K is the proportional gain coefficient; r ω is the resonant gain coefficient. cg ω is the controller bandwidth; res This refers to the harmonic frequency.

[0148] 42) The inner loop uses active damping control with capacitor current feedback: the capacitor branch current i in the LCL filter circuit is controlled by...Cabc i is obtained after abc / αβ coordinate transformation Cα and i Cβ Step 41) The output of the first QPR controller 10 and i Cα The difference is calculated, and the grid-connected voltage reference value u is obtained through the third PI controller 12. gα * Step 41) The output of the second QPR controller 11 and i Cβ The difference is calculated, and the grid-connected voltage reference value u is obtained through the fourth PI controller 13. gβ * ;

[0149] 43) will u gα * and u gβ * After modulation by the SVPWM module, a drive signal is generated to drive the power switch of the grid-side converter 3 and control the operation of the grid-side converter 3.

[0150] The present invention will be further described below using a preferred embodiment.

[0151] 1. To verify the harmonic suppression advantage of using Vienna rectifier as machine-side converter in this invention, simulations and experiments were conducted on machine-side converter 2 using Vienna rectifier and traditional voltage source converter (VSC) respectively, and the harmonic content and efficiency of each were compared and analyzed.

[0152] Figure 6 The image shows the simulated stator current waveform of the permanent magnet synchronous wind turbine generator 1 at rated wind speed (9 m / s) when the generator-side converter 2 uses a Vienna rectifier (VR) and a VSC. From... Figure 6 It can be seen that with VSC, the stator current distortion rate of phase a is 5.75%, while with Vienna rectifier, the stator current distortion rate of phase a is 2.92%. Clearly, Vienna rectifier has a better harmonic suppression effect.

[0153] Figure 7 The figure shows an experimental graph illustrating the power generation efficiency of the permanent magnet synchronous wind turbine generator 1 when the generator-side converter 2 uses a Vienna rectifier (VR) and a VSC. From... Figure 7It can be seen that when using the Vienna rectifier as the topology for the machine-side converter 2, the overall power generation efficiency η is greater than 88%, reaching a maximum of 91.46% near the rated wind speed. However, when using the VSC as the topology for the machine-side converter 2, η reaches a maximum of 89.03%, which is 2.43% lower than that of the Vienna rectifier. Furthermore, as the wind speed increases, η first increases and then stabilizes (7.6 m / s). Using the Vienna rectifier allows η to stabilize earlier (6.4 m / s). The results indicate that the PMSG experimental platform based on the Vienna rectifier has lower power loss.

[0154] 2. In order to verify the advantages of the combined active damping control and quasi-proportional resonant control control strategy (hereinafter referred to as "improved active damping method") of the grid-side converter 3 of the present invention, a comparative analysis is conducted with the voltage-oriented control strategy.

[0155] 1) First, compare the harmonic suppression capabilities of the two control strategies under normal filter parameters. The simulation parameters are shown in Table 1.

[0156] Table 1 Main Technical Parameters of Grid-Side Converter

[0157]

[0158] Figure 8 , Figure 9 The figures show the three-phase grid-connected current waveforms, current harmonics, and distortion rates under two different control strategies. Figure 9 It is known that the grid-connected current waveform using voltage-oriented control has a relatively high harmonic content, with a total harmonic distortion rate of 4.37%. The high-content harmonics include odd-order harmonics such as the 5th, 7th, and 11th harmonics. After adopting the improved active damping method of this invention, the total harmonic distortion rate of the grid-connected current waveform is only 1.67%, the 5th harmonic is almost reduced to 0, and the 7th harmonic content is significantly reduced.

[0159] 2) Next, compare the harmonic suppression capabilities of the two control strategies when the filter parameters are reduced. In this case, the filter inductance is reduced by a factor of 6, i.e., L = 1mH, L g =0.25mH. The remaining simulation parameters in Table 1 remain unchanged.

[0160] Figure 10 , Figure 11 The figures show the three-phase grid-connected current waveforms, current harmonics, and distortion rates after the filter inductance is reduced by a factor of 6 under two different control strategies. Figure 11 It is evident that the grid-connected current waveform using voltage-oriented control has a relatively high harmonic content, with a total harmonic distortion rate of 4.37%. The most prevalent harmonics are odd-order harmonics such as the 5th, 7th, and 11th harmonics. After applying the improved active damping method of this invention, the total harmonic distortion rate of the grid-connected current is 3%, the 11th harmonic is almost reduced to zero, and the contents of the 5th and 7th harmonics are significantly decreased.

[0161] Based on the simulation results above, it can be seen that compared with the voltage-oriented control strategy, the improved active damping method of this invention can not only greatly reduce the total harmonic distortion rate of the grid-connected current, but also use smaller filter inductors, thereby achieving lightweight and miniaturized filters, and further reducing costs and losses.

[0162] In summary, this invention, by employing a control strategy that combines Vienna rectifiers, sensorless algorithms, proportional-integral resonant controllers, active damping control with capacitor current feedback, and quasi-proportional resonant control, can effectively suppress stator current harmonic distortion and grid-side converter current harmonic distortion in permanent magnet synchronous wind turbines, thereby reducing the power consumption of the entire wind turbine unit. It can achieve the goals of low power consumption, lightweight, miniaturization, low cost, and high efficiency in low-wind-speed permanent magnet direct-drive wind turbine units.

Claims

1. A harmonic suppression and power consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine generator, wherein the low-wind-speed permanent magnet direct-drive wind turbine generator comprises: Permanent magnet synchronous wind turbine generator, machine-side converter, grid-side converter; One end of the generator-side converter is connected to the stator of the permanent magnet synchronous wind turbine, and the other end is connected to the grid-side converter; the grid-side converter is connected to the power grid through a power frequency transformer; the generator-side converter is a Vienna rectifier, which includes three bidirectional switching units and two output capacitors; the two output capacitors are connected in series, wherein the output capacitor connected to the positive terminal of the DC output of the Vienna rectifier is called the upper capacitor, and the output capacitor connected to the negative terminal of the DC output of the Vienna rectifier is called the lower capacitor; the grid-side converter is a voltage source converter, which includes an LCL filter; characterized by the following steps: Step 1: Obtain the d-axis position angle θ using a sensorless algorithm based on a sliding mode observer. se ; Step 2: For the machine-side converter, a proportional-integral resonant controller with variable resonant angular frequency is used to suppress the specified harmonics; Step 3: The Vienna rectifier is controlled using carrier pulse width modulation technology based on zero-sequence component injection and midpoint potential balance. Step 4: The grid-side converter adopts a control strategy that combines active damping control with capacitor current feedback and quasi-proportional resonant control to filter the grid-side current.

2. The harmonic suppression and power consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine according to claim 1, characterized in that, The specific process of step 1 is as follows: 11) The stator output of the permanent magnet synchronous wind turbine generator is a three-phase alternating current with variable frequency and amplitude, the voltage of which is denoted as u. a u b u c Their currents are denoted as i a i b i c ; will u a u b u c u is obtained through abc / αβ coordinate transformation α and u β ;change i a i b i c i is obtained through abc / αβ coordinate transformation α and i β ; 12) The voltage equation of the permanent magnet synchronous wind turbine in the αβ coordinate system is: In the formula, L s For stator inductance; R s ω is the stator resistance; e ψ is the electric angular velocity of the permanent magnet synchronous wind turbine. f θ is the rotor permanent magnet flux linkage of the permanent magnet synchronous wind turbine generator; se Let be the d-axis phase angle, and have In the formula, e sα e sβ These are the α-axis and β-axis components of the back electromotive force, respectively; 13) To facilitate the use of a sliding mode observer to estimate the back electromotive force, the voltage equation in equation (1) is rewritten in the form of a current state equation: 14) To obtain the back electromotive force e sα e sβ The estimated value e sα,e e sβ,e The sliding mode observer is designed as follows: In the formula, i α,e i β,e The stator current i output by the sliding mode observer is respectively α i β The estimated value; u α u β The input value for the sliding mode observer; 15) Subtracting equations (3) and (4), we obtain the stator current error equation as follows: 16) The sliding mode control rate is designed as follows: In the formula, sgn() is the sign function; k is the maximum amplitude of the generator's no-load back electromotive force, and satisfies: k>max{e sα ,and sβ } (7) 17) The sliding surface is designed as (i α -i α,e ), (i β -i β,e By switching rapidly between positive and negative sliding surfaces, the back electromotive force error (e) is reduced. sα -e sα,e ), (e sβ -e sβ,e It quickly converges to 0, at which point we can obtain: In actual operation e sα,e e sβ,e It is a discontinuous high-frequency switching signal. To obtain a continuous waveform, an external low-pass filter is required, which is expressed as: In the formula, τ0 is the time constant of the low-pass filter; 18) In obtaining e sα e sβ Then, θ is obtained using the arctangent function method. se That is, it can be expressed as: Filtering will cause phase delay. Angle compensation is added to equation (10), which is expressed as: In the formula, ω c Let ω be the cutoff frequency of the low-pass filter. c =1 / τ0.

3. The harmonic suppression and power consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine according to claim 1, characterized in that, The specific process of step 2 is as follows: 21) Based on the electric angular velocity ω of the permanent magnet synchronous wind turbine generator e The fundamental frequency of the stator current is obtained, and then the stator current i is... s Perform Fast Fourier Decomposition and calculate the current distortion rate (THD) according to Equation (12): In the formula, I1 is the amplitude of the fundamental current, I k The value of the kth harmonic current; 22) The stator voltage and current of the permanent magnet synchronous wind turbine generator are obtained by abc / dq coordinate transformation to obtain u respectively. d u q and i d i q ; 23) Set a reference value for THD. * =0, the difference between this reference value and its actual value is input to the first PI controller, and the output of the first PI controller is equal to T. e * / (1.5pψ f Add them together to get i q Reference value i q * T e * The electromagnetic torque T of the permanent magnet synchronous wind turbine generator e Reference value; 24) Set the reference value i of the stator current. q * Its actual value i q The difference is input to the first proportional-integral resonant controller, and the output of the first proportional-integral resonant controller is decoupled from the q-axis feedforward decoupling compensation term -ω. e L s i d +ω e ψ f By adding them together, we obtain the q-axis component u of the stator voltage of the permanent magnet synchronous wind turbine. q Reference value u q * ; The transfer function of the first proportional-integral resonant controller is: In the formula, K p K is the proportionality coefficient. i ω is the integral coefficient; cs The controller bandwidth is given by h; the harmonic order is given by K. h The h-th harmonic resonant gain coefficient; 25) Let the d-axis component i of the stator current of the permanent magnet synchronous wind turbine be... d Reference value i d * =0, and compare it with its actual value i d The second proportional-integral resonant controller is used as the input, and the output of the second proportional-integral resonant controller is decoupled from the d-axis feedforward decoupling compensation term ω. e L s i q By adding them together, we obtain the d-axis component u of the stator voltage of the permanent magnet synchronous wind turbine. d Reference value u d * The transfer function of the second proportional-integral resonant controller is the same as that of the first proportional-integral resonant controller.

4. The harmonic suppression and power consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine according to claim 1, characterized in that, The specific steps of step 3 are as follows: 31) The stator voltage reference value u obtained in steps 24) and 25) q * and u d * The voltage u is obtained after dq / abc transformation. a u b u c Reference value u a * u b * u c * Let the DC side voltage U of the Vienna rectifier be... dc Half of it is used as the baseline value, i.e., U dc / 2, for u a * u b * u c * Standardize to obtain the per-unit value u. ap * u bp * u cp * Find u ap * u bp * u cp * The maximum and minimum values ​​in the range are denoted as u. pmax u pmin ; 32) Define the zero-order component as u0: u0=-0.5(u pmax +u pmin ) (14) 33) Inject the zero-sequence components into u respectively. ap * u bp * u cp * , to obtain m ap * m bp * m cp * Find m ap * m bp * m cp * The maximum and minimum values ​​in the range are denoted as m. pmax m pmin ; 34) Measure the voltage U1 across the upper capacitor and the voltage U2 across the lower capacitor of the Vienna rectifier; the difference between the two is input to the second PI controller, and the output of the second PI controller is recorded as the suppression factor z; 35) The inhibition factor z is limited by a limiter, with an upper limit of 1-m. pmax Its lower limit is -1-m pmin ; 36) The suppression factor z after the limiter is compared with the m obtained in step 33). ap * m bp * m cp * Add them together to obtain the corrected voltage reference value u. am * u bm * u cm * ; 37) The corrected voltage reference value u am * u bm * u cm * The drive signals for the three bidirectional switching units of the Vienna rectifier are obtained by carrier pulse width modulation, and the three bidirectional switching units are driven to work.

5. The harmonic suppression and power consumption reduction control method for a low-wind-speed permanent magnet direct-drive wind turbine according to claim 1, characterized in that, The specific process of step 4 is as follows: 41) Quasi-resonant control is adopted in the outer loop: the three-phase current value that meets the grid connection requirements is used as the reference value i of the grid-side three-phase current. * gabc The control objective is to make the filter output current in phase with the grid voltage, that is: The reference current i * gabc i is obtained through abc / αβ coordinate transformation gα * and i gβ * , respectively with their actual current i gα and i gβ After the difference is calculated, the values ​​are input to the first quasi-proportional resonant controller and the second quasi-proportional resonant controller, respectively. The transfer functions of both the first quasi-proportional resonant controller and the second quasi-proportional resonant controller are: In the formula, K pr K is the proportional gain coefficient; r ω is the resonant gain coefficient. cg ω is the controller bandwidth; res Harmonic frequencies; 42) The inner loop adopts active damping control with capacitor current feedback: the capacitor branch current i in the LCL filter circuit of the grid-side converter is controlled by... Ca i Cb i Cc i is obtained through abc / αβ coordinate transformation Cα and i Cβ Step 41) The output of the first quasi-proportional resonant controller and i Cα The difference is calculated, and the grid-connected voltage reference value u is obtained through the third PI controller. gα * Step 41) The output of the second quasi-proportional resonant controller and i Cβ The difference is calculated, and the grid-connected voltage reference value u is obtained through the fourth PI controller. gβ * ; 43) will u gα * and u gβ * The SVPWM module generates a drive signal to drive the power switch of the grid-side converter and control the operation of the grid-side converter.

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