Motor control method and system for a non-inductive control sensor

Abstract

The application relates to the technical field of motor control, and discloses a motor control method and system of a non-inductive control sensor, which comprises the following steps: performing pulse voltage signal injection processing on an estimated d-axis of a permanent magnet synchronous motor and extracting three-phase high-frequency currents; performing adaptive dead-zone compensation based on the three-phase high-frequency currents to obtain high-frequency currents after dead-zone compensation; inputting the high-frequency currents after dead-zone compensation into a position phase-locked loop for synchronous demodulation and PI control to obtain a rotor position angle; establishing a q-axis starting current according to the rotor position angle to obtain a starting torque instruction; driving the permanent magnet synchronous motor to rotate by executing the starting torque instruction and performing real-time detection on a wheel hub rotating speed; when the rotating speed reaches a preset target value, performing switching processing from a high-frequency injection mode to an observation mode of back electromotive force to output a double-mode control switching signal; and the application solves the step-out problem of traditional non-inductive control under variable load starting, and improves the starting success rate.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, and in particular to a motor control method and system without a sensor. Background Technology

[0002] Traditional permanent magnet synchronous motor control systems rely on position sensors to obtain rotor position information. This not only increases system cost and complexity but also reduces system reliability. In particular, position sensors are prone to failure under harsh operating conditions such as frequent starts and stops of scooters and vibration and shock.

[0003] Existing sensorless control methods mainly fall into two categories: back-EMF observation and high-frequency signal injection. Back-EMF observation performs well at medium and high speeds, but the back-EMF signal is weak or even zero at low speeds or zero-speed starts, making accurate rotor position detection impossible. While high-frequency signal injection can operate at low speeds, existing technologies are mostly designed for high-voltage, high-power motor systems. In applications with 48V low-voltage, low-power permanent magnet synchronous hub motors, they suffer from low signal-to-noise ratios and insufficient position detection accuracy. Furthermore, the dead-zone effect is significantly amplified when the low-power motor operates under light loads, leading to position estimation drift and severely impacting start-up reliability. Summary of the Invention

[0004] This invention provides a motor control method and system with a sensorless control system. This invention solves the problem of step loss during variable load startup in traditional sensorless control and improves the startup success rate.

[0005] In a first aspect, the present invention provides a motor control method using a non-contact control sensor, the motor control method comprising: Pulsed voltage signal injection processing is performed on the estimated d-axis of the permanent magnet synchronous motor, and three-phase high-frequency current is extracted. Adaptive dead-zone compensation is performed based on the three-phase high-frequency current to obtain the high-frequency current after dead-zone compensation. The high-frequency current input position phase-locked loop after dead-zone compensation is synchronously demodulated and PI controlled to obtain the rotor position angle. The q-axis starting current is established based on the rotor position angle to obtain the starting torque command; The starting torque command is executed to drive the permanent magnet synchronous motor to rotate and the hub speed is detected in real time. When the speed reaches the preset target value, the high-frequency injection mode is switched to the back electromotive force observation mode, and a dual-mode control switching signal is output.

[0006] In conjunction with the first aspect, in a first implementation of the first aspect of the present invention, the step of injecting a pulsed voltage signal into the estimated d-axis of the permanent magnet synchronous motor and extracting the three-phase high-frequency current includes: A standard pulsed fundamental frequency signal is generated by the internal timer of the DSP controller; The standard pulse fundamental frequency signal is input into the amplitude modulation circuit for 2V amplitude control to obtain an amplitude-controlled pulse voltage signal; The amplitude-controlled pulsating voltage signal is isolated by an isolation transformer and bandpass filtered to obtain a clean high-frequency injection signal; Based on the current rotor position angle, the pure high-frequency injection signal is transformed to the estimated d-axis coordinate system and injected to obtain the pulsating high-frequency injection voltage acting on the d-axis winding of the permanent magnet synchronous motor. The high-frequency components of the three-phase winding current of the permanent magnet synchronous motor under the action of the pulsed high-frequency injection voltage are extracted to obtain the three-phase high-frequency current.

[0007] In conjunction with the first aspect, in a second implementation of the first aspect of the present invention, the step of extracting the high-frequency components of the three-phase winding current of the permanent magnet synchronous motor under the action of the pulsed high-frequency injection voltage to obtain the three-phase high-frequency current includes: The three-phase winding current of the permanent magnet synchronous motor under the pulsed high-frequency injection voltage is synchronously sampled to obtain the three-phase current sampling signal. The three-phase current sampling signal is bandpass filtered to obtain three-phase high-frequency current components, and the three-phase high-frequency current components are lowpass filtered and signal conditioned to obtain a clean three-phase high-frequency current signal. Amplitude and phase information are extracted based on the pure three-phase high-frequency current signal to obtain a three-phase high-frequency current containing rotor position information.

[0008] In conjunction with the first aspect, in a third implementation of the first aspect of the present invention, the step of performing adaptive dead-zone compensation based on the three-phase high-frequency current to obtain the dead-zone compensated high-frequency current includes: The average value and unbalance index of the three-phase high-frequency current are calculated to obtain the unbalance value of the three-phase high-frequency current. The unbalance value of the three-phase high-frequency current is compared with a preset threshold to obtain a target comparison result. Based on the target comparison result, an adaptive compensation coefficient is calculated to obtain a dynamic dead zone compensation coefficient. The dead zone compensation voltage is calculated based on the dynamic dead zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead zone compensation voltage signal. The three-phase dead-zone compensation voltage signal is superimposed on the three-phase high-frequency current for compensation and correction, resulting in the high-frequency current after dead-zone compensation.

[0009] In conjunction with the first aspect, in the fourth implementation of the first aspect of the present invention, the step of calculating the dead-zone compensation voltage based on the dynamic dead-zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead-zone compensation voltage signal includes: The sign of the three-phase high-frequency currents is determined to obtain the sign functions of the A-phase, B-phase, and C-phase currents; The dead-time voltage is calculated based on the preset ideal dead-time and PWM switching frequency to obtain the ideal dead-time compensation voltage reference value. The dynamic dead zone compensation coefficient is multiplied by the ideal dead zone compensation voltage reference value to obtain the compensation voltage amplitude. The three-phase dead zone compensation voltage signal is obtained by performing sign matching and amplitude synthesis with the sign functions of the currents in phases A, B, and C, respectively, based on the compensation voltage amplitude.

[0010] In conjunction with the first aspect, in the fifth implementation of the first aspect of the present invention, the step of synchronously demodulating and PI controlling the high-frequency current input position phase-locked loop after dead-zone compensation to obtain the rotor position angle includes: The high-frequency current input position phase-locked loop after dead-zone compensation is synchronously multiplied and demodulated with the reference sine signal and the reference cosine signal to obtain the demodulated signal. The demodulated signal is subjected to low-pass filtering and quadrature component separation to obtain a first position detection component and a second position detection component; The rotor position error angle is obtained by performing an arctangent function operation based on the first position detection component and the second position detection component. The rotor position error angle is input into a variable gain PI controller for segmented gain adjustment and closed-loop control to obtain the rotor position angle.

[0011] In conjunction with the first aspect, in a sixth implementation of the first aspect of the present invention, the step of performing arctangent function calculation based on the first position detection component and the second position detection component to obtain the rotor position error angle includes: The first position detection component and the second position detection component are subjected to amplitude verification and division-by-zero protection preprocessing to obtain orthogonal position detection components within a safe range; The orthogonal position detection components within the safe range are input into the four-quadrant arctangent function to calculate the angle, thus obtaining the initial position error angle. Based on the sign and magnitude of the initial position error angle, quadrant correction and angle range adjustment are performed to obtain the corrected position error angle; The rotor position error angle is obtained by performing precision limiting and numerical stability filtering on the corrected position error angle.

[0012] In conjunction with the first aspect, in the seventh implementation of the first aspect of the present invention, the step of establishing the q-axis starting current based on the rotor position angle to obtain the starting torque command includes: The absolute value of the position error is calculated based on the rotor position angle and compared with the radian threshold to obtain a position locking success confirmation signal. Based on the successful confirmation signal of position locking, a three-stage q-axis current establishment strategy is initiated and segmented current increment processing is performed to obtain a segmented q-axis current command that includes preload stage current, acceleration stage current and steady-state stage adjustable current. The d-axis current is set to zero and kept constant to obtain a d-axis zero current command signal. Based on load current change rate monitoring, and when the change rate exceeds the target current value, the current slope is adaptively adjusted to obtain a fast current slope command. Based on the fast current slope command, the phased q-axis current command and the d-axis zero current command are combined in coordinates and torque is calculated to obtain the starting torque command.

[0013] In conjunction with the first aspect, in the eighth implementation of the first aspect of the present invention, the step of executing the starting torque command to drive the permanent magnet synchronous motor to rotate and real-time detection of the hub speed, and when the speed reaches a preset target value, performing a switching process from high-frequency injection mode to back electromotive force observation mode, and outputting a dual-mode control switching signal, includes: The starting torque command is input to the drive controller of the permanent magnet synchronous motor for PWM modulation and three-phase drive processing to obtain the three-phase drive voltage that drives the permanent magnet synchronous motor to rotate. Based on the rotor position angle, a multi-point difference algorithm is used for calculation and numerical differentiation to obtain the real-time detection value of the hub speed. The real-time detected value of the wheel hub speed is compared with the preset speed threshold and the time is confirmed to obtain a speed compliance confirmation signal; Based on the speed target confirmation signal, the back EMF observer is started and runs in parallel with the high-frequency injection position detector and the position error is compared to obtain the mode switching ready signal. Based on the mode switching readiness signal, the amplitude of the high-frequency injection signal is linearly reduced from 2V to 0V, and the back-cut threshold monitoring is activated at the same time. The output is a dual-mode control switching signal that includes forward switching and back-cut protection.

[0014] Secondly, the present invention provides a motor control system for a sensorless control system, the motor control system for the sensorless control system comprising: The injection module is used to inject pulse voltage signals into the estimated d-axis of the permanent magnet synchronous motor and extract the three-phase high-frequency current. The compensation module is used to perform adaptive dead-zone compensation based on the three-phase high-frequency current to obtain the high-frequency current after dead-zone compensation. The control module is used to synchronously demodulate and PI control the high-frequency current input position phase-locked loop after dead-zone compensation to obtain the rotor position angle. A module is established to generate the q-axis starting current based on the rotor position angle, thereby obtaining the starting torque command. The output module is used to execute the starting torque command to drive the permanent magnet synchronous motor to rotate and to detect the hub speed in real time. When the speed reaches the preset target value, it performs the switching process from high frequency injection mode to back electromotive force observation mode and outputs a dual-mode control switching signal.

[0015] The technical solution provided by this invention, through precise design of a 300Hz injection frequency and a 2V injection amplitude, ensures position detection accuracy under low-voltage systems while avoiding the impact of high-frequency noise on the riding experience, thus solving the problem of low signal-to-noise ratio in existing high-frequency injection methods for low-power hub motors. By real-time detection of three-phase high-frequency current imbalance and dynamic adjustment of the compensation coefficient, the position estimation drift problem caused by the amplification of dead zone effect under light load in low-power motors is effectively eliminated, significantly improving starting reliability. An improved PI control structure and variable gain strategy are adopted to compress the position convergence time to 30ms, meeting the expectations of scooter users for rapid start-up response and improving the riding experience. A three-stage linear slope current establishment method ensures sufficient starting torque while avoiding current surges, achieving smooth and rapid starts and solving the starting shock and vibration problems caused by traditional step torque setting. The problem of switching point drift under different loads, which is a problem of traditional voltage detection methods, is avoided, ensuring the accuracy and stability of control mode switching and improving control performance throughout the entire operating range. By monitoring load changes in real time and adaptively adjusting control parameters, and being optimized for high-load conditions such as uphill runs, the problem of step loss during variable load startup in traditional sensorless control has been solved, thus improving the startup success rate. Attached Figure Description

[0016] To more clearly illustrate the technical solutions of the embodiments of the present invention, the drawings used in the description of the embodiments will be briefly introduced below. Obviously, the drawings described below are some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0017] Figure 1 This is a schematic diagram of the steps of the motor control method using a sensorless control sensor in an embodiment of the present invention; Figure 2 This is a schematic diagram of the motor control system of the sensorless control system in an embodiment of the present invention. Detailed Implementation

[0018] This invention provides a motor control method and system using a non-sensory control sensor. The terms "first," "second," "third," "fourth," etc. (if present) in the specification, claims, and accompanying drawings are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that such data can be interchanged where appropriate so that the embodiments described herein can be implemented in a sequence other than that illustrated or described herein. Furthermore, the terms "comprising" or "having" and any variations thereof are intended to cover a non-exclusive inclusion; for example, a process, method, system, product, or device that includes a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units not explicitly listed or inherent to such processes, methods, products, or devices.

[0019] For ease of understanding, the specific process of the embodiments of the present invention is described below. Please refer to [link / reference]. Figure 1 One embodiment of the motor control method using a sensorless control system according to the present invention includes: Step S1: Inject pulse voltage signal into the estimated d-axis of the permanent magnet synchronous motor and extract the three-phase high-frequency current; It is understood that the executing entity of this invention can be a motor control system with a sensorless control system, or it can be a terminal or a server; no specific limitation is made here. This embodiment of the invention will be described using a server as an example.

[0020] Specifically, the control system utilizes a high-precision timer module integrated within the DSP controller to generate a set of standard pulsed fundamental frequency signals. The frequency of this signal is designed to match the winding inductance characteristics of the permanent magnet synchronous motor and the system sampling period, and is set to 300Hz to ensure sufficient response bandwidth for the high-frequency signal within the PWM cycle. The pulsed fundamental frequency signal output from the timer is input to the amplitude modulation circuit module. This module, through the combined action of the control voltage reference channel and the modulation logic circuit, limits the signal amplitude to a 2V voltage range, resulting in a pulsed voltage signal with controlled amplitude. The controlled pulsed voltage signal is then input to an isolation transformer with electrical isolation function for signal isolation. Magnetic coupling is used to block DC potential coupling between the main circuit and the high-frequency signal path, improving the overall anti-interference capability of the control system and ensuring that the injected signal path does not affect the main power circuit. The isolated signal is then subjected to frequency domain shaping processing through a bandpass filter circuit to filter out stray interference and broadband noise outside the designed injection frequency range, retaining only the target spectral components mainly concentrated around 300Hz, resulting in a high-frequency injection signal with a clean spectrum and stable waveform. Based on the rotor position angle estimated in the previous control cycle of the current control system, the high-frequency injected signal is transformed in real time to the estimated rotating coordinate system of the d-axis using a coordinate transformation algorithm (such as inverse Park transform). This ensures that the injected signal direction is precisely aligned with the permanent magnet synchronous motor's pole axis, thereby maximizing the modulation effect of reluctance difference on the injected response. By injecting the transformed high-frequency signal as the d-axis direction control voltage into the motor stator windings, current response characteristics reflecting changes in pole position are excited without relying on position sensors. At this time, under the action of the high-frequency injected voltage, the three-phase stator windings of the motor will generate transient current signals containing high-frequency response characteristics. The system extracts the i-th current signal from the three-phase windings through the three-phase current sampling channels. a i b i c The current data is analyzed, and a high-pass filtering algorithm or a phase-locked bandpass filter is used to extract the high-frequency current component centered at 300Hz. a_hf i b_hf with i c_hf This constitutes a three-phase high-frequency current output signal.

[0021] Step S2: Perform adaptive dead-zone compensation based on the three-phase high-frequency current to obtain the high-frequency current after dead-zone compensation; Specifically, i is obtained from the three-phase high-frequency current. a_hf i b_hf and i c_hfAs the basic input data for compensation calculation, the average value of the three-phase high-frequency current is calculated based on this. By taking the arithmetic mean of the three-phase current, an average current value representing the overall trend of the high-frequency current center is obtained. Then, based on this average value, the deviation of each phase current from the average value is calculated, and the sum of squares of each deviation term is performed to obtain the unbalance value of the three-phase high-frequency current, reflecting the current distortion phenomenon caused by the dead zone effect when the motor is running under light load or at rest. The unbalance value is compared in real time with a preset dead zone drift detection threshold, where the threshold is set to 0.1A, to define whether the high-frequency current deviation reaches the judgment standard for the compensation level. When the unbalance value exceeds this threshold, it is determined that the dead zone compensation trigger condition is met, and the adaptive compensation coefficient calculation mechanism is activated accordingly. This mechanism dynamically adjusts the compensation coefficient through a nonlinear gain function related to the unbalance value, so that the dead zone compensation can continuously change with the actual degree of the current response offset, resulting in a dynamic dead zone compensation coefficient, thereby avoiding the problems of miscompensation or overcompensation caused by changes in the size of the dead zone. Based on the dynamic dead-zone compensation coefficient and the sign and direction information of the current three-phase instantaneous current, i.e., the conduction direction of the current half-bridge power device is determined according to the polarity state of each phase current. This determines the voltage offset direction of the dead-zone effect. The dynamic compensation coefficient is then multiplied and combined with this polarity function to obtain the corresponding dead-zone compensation voltage signal for each phase. This signal is then used to construct the voltage offset for i... a_hf i b_hf and i c_hf The compensation correction amount is then applied. Finally, the three-phase dead-time compensation voltage signals are superimposed onto the original three-phase high-frequency current signals to form a dead-time corrected high-frequency current output, i.e., a dead-time compensated high-frequency current. This effectively suppresses the nonlinear error introduced by the driver's dead time and improves the symmetry and linearity of the current response.

[0022] Step S3: Input the high-frequency current after dead-zone compensation into the position phase-locked loop for synchronous demodulation and PI control to obtain the rotor position angle; Specifically, the dead-zone-compensated high-frequency current is input to the position phase-locked loop (PLL) module and synchronously multiplied with a set of reference sine and cosine signals. These two reference signals have the same frequency characteristics as the high-frequency injected signal, and the frequency is set to 300Hz to complete the orthogonal synchronous demodulation process for the injected frequency components. During the synchronous multiplication stage, the dead-zone-compensated high-frequency current is modulated onto the sine and cosine reference bases respectively, thus obtaining two corresponding original demodulated signals. These signals contain the amplitude change information caused by the relative position relationship between the rotor magnetic poles and the injected signal, which is a direct reflection of the rotor position change in the high-frequency response. The demodulated signals are input to the low-pass filter module for processing. By selecting a filter with a reasonable cutoff frequency, the high-frequency beat frequency components and harmonic interference generated during the modulation process are suppressed, and only the low-frequency signal components related to the rotor magnetic pole position change are retained. The filtered output signal is the orthogonal component containing rotor position information, one of which is the sine component and the other is the cosine component, which are denoted as the position detection component and its corresponding orthogonal position detection component, respectively. These two orthogonal signals form an orthogonal vector reflecting the relative angle information of the rotor. After obtaining the orthogonal components, the two signals are processed using an arctangent function. Specifically, the position detection component is used as the numerator, and the orthogonal component as the denominator, to perform an arctangent calculation, resulting in a rotor position error angle value containing relative phase deviation information. This angle value reflects the relative position difference between the currently injected signal and the rotor magnetic poles, representing the phase-locked offset that the system should correct within the current control cycle. This position error angle is input to a variable-gain PI controller. The PI controller sets multiple proportional gain parameters for different angle error amplitude ranges. A larger proportional gain is used when the error angle is large to achieve rapid locking, while the proportional gain is gradually reduced as the error converges towards zero to avoid overshoot oscillations. Simultaneously, its integral gain uses an adaptive strategy based on the error amplitude for real-time adjustment, ensuring that the system maintains good response and stability at different control stages. The output after processing by the PI controller is the currently estimated rotor position angle.

[0023] Step S4: Establish the q-axis starting current based on the rotor position angle to obtain the starting torque command; Specifically, based on the rotor position angle output by the position phase-locked loop (PLL), the absolute value of the current position error is calculated and compared with a preset radian threshold, which is set to π / 36, or approximately 5 degrees, to determine whether the rotor position has been stably locked. When the system detects that the position error is consistently below this threshold for a continuous period of more than 10 ms, it considers the rotor position to have been reliably locked, thus generating a position lock success confirmation signal. After the position lock success signal is triggered, a three-stage q-axis current establishment strategy is initiated. This strategy gradually increases the q-axis current in a linear, piecewise manner to ensure that the current rise during the initial start-up process of the motor can overcome static friction without causing current surges or vibrations. The process involves three stages: First, a pre-loading stage, where the current linearly increases from 0 to 1A to preload the hub motor's rotational inertia and eliminate initial static resistance. Second, an acceleration stage, where the q-axis current continues to linearly increase from 1A to 2.5A, providing the main starting torque to initiate the scooter's start-up. Third, a steady-state stage, where the q-axis current is maintained within a dynamically adjustable range based on actual load requirements to match the vehicle's operating conditions after startup. These three stages form a complete and gradual current increase process, avoiding the impact load problem caused by traditional step-current commands. Simultaneously, to ensure optimal torque output efficiency, the d-axis current is set to zero and remains constant throughout the entire startup process. This is achieved by directly setting the d-axis current to zero using a constant-value command and locking the controller output of that channel, concentrating all current energy on the q-axis to maximize electromagnetic torque generation. During the execution of the aforementioned current strategy, the real-time rate of change of the q-axis current is monitored synchronously. When the current growth rate exceeds the preset target rate of change threshold, such as 5A / s, it is determined to be a load change event, and the current slope parameter is immediately dynamically adjusted according to the rate of change, increasing the default linear rising slope from 0.5A / 15ms to 1A / 10ms. This allows for rapid adaptation to complex operating conditions such as uphill or sudden load changes, ensuring that the motor does not lose synchronization during startup and maintains a stable operating state. The controller synthesizes the phased q-axis current command and the constant d-axis zero current command through coordinate transformation, that is, transforming from the qd synchronous rotating coordinate system back to the stationary abc three-phase coordinate system. During the transformation, the torque contribution of the current vector direction and amplitude is calculated, and combined with the current estimated position angle, the final starting torque command is generated.

[0024] Step S5: Execute the starting torque command to drive the permanent magnet synchronous motor to rotate and detect the hub speed in real time. When the speed reaches the preset target value, switch from high frequency injection mode to back EMF observation mode and output dual-mode control switching signal.

[0025] Specifically, the start torque command is input to the drive controller of the permanent magnet synchronous motor. Within the drive controller, the PWM modulation module performs spatial voltage vector deconstruction and modulation encoding on this command, forming a PWM signal sequence capable of driving power switching devices. Simultaneously, the modulated PWM signal drives the three-phase bridge arms to generate continuous three-phase drive voltage waveforms, thereby acting on the three-phase windings of the permanent magnet synchronous motor in a high-precision and high-response manner. This drives the motor to start slowly according to the preset torque, achieving accelerated rotor rotation and linear hub displacement response. During the controlled start and gradual acceleration of the motor rotor, the previously established rotor position angle is used as the input basis. A multi-point differential algorithm is employed to perform numerical differentiation processing to obtain an estimated rotational speed. This differential method, based on a five-point central differential structure, constructs a difference quotient expression, which improves the noise resistance and transient change capture capability of the differential calculation, thereby achieving high-precision real-time detection of the actual hub rotational speed. The system continuously compares the obtained real-time speed data with a pre-set target speed threshold of 100 rpm, serving as the lower limit for sufficient signal-to-noise ratio operation in the back EMF observation mode. A time confirmation mechanism is also implemented, requiring the speed to continuously exceed this threshold and remain above it for at least 50 ms before outputting a speed compliance confirmation signal to eliminate false triggering caused by transient fluctuations. Once the speed is determined to have stabilized at the set threshold, the back EMF observer is activated and runs in parallel with the previously running high-frequency injection position detector. At this time, both estimators simultaneously estimate the current motor position. The system continuously collects the position angle data output by both estimators and performs error calculation and amplitude comparison. If the position error between the two is determined to be below the set threshold, the system issues a mode switching readiness signal, indicating that the system is ready to smoothly transition from the high-frequency injection estimation mode to the back EMF observation mode. To ensure that the mode switching process does not cause control jitter or estimation interruption, based on the mode switching readiness signal, a high-frequency injection amplitude linearly decreasing control strategy is executed. The original injection voltage amplitude is gradually reduced from 2V to 0V at a stable linear rate, while the back EMF observer continues to operate during this period, allowing it to independently perform the position estimation task. Simultaneously, to prevent the motor speed from dropping due to subsequent load changes or downhill deceleration, a back-cut threshold monitoring mechanism is activated. When the hub speed drops below 80rpm again and remains below this level for a certain period, the high-frequency injection mode is automatically restarted, switching back to the low-speed, high-precision position estimation channel. The output includes a dual-mode control switching signal containing two logics: forward switching to back EMF observation and backward switching back to high-frequency injection.

[0026] In one specific embodiment, the process of performing step S1 may specifically include the following steps: A standard pulsed fundamental frequency signal is generated by the internal timer of the DSP controller; The standard pulse fundamental frequency signal is input into the amplitude modulation circuit for 2V amplitude control to obtain a pulse voltage signal with controlled amplitude. An isolation transformer and bandpass filtering are applied to the amplitude-controlled pulsating voltage signal to obtain a clean high-frequency injection signal. Based on the current rotor position angle, the pure high-frequency injection signal is transformed to the estimated d-axis coordinate system and injected to obtain the pulsating high-frequency injection voltage acting on the d-axis winding of the permanent magnet synchronous motor. The high-frequency components of the three-phase winding current of a permanent magnet synchronous motor under the action of pulsating high-frequency injected voltage are extracted to obtain the three-phase high-frequency current.

[0027] Specifically, a high-frequency pulse signal generation channel is constructed within the system from the source of signal generation. The core of this channel is a timer within the DSP controller, which serves as the reference timing generator. By configuring the timer's operating frequency, period parameters, and duty cycle output mode, a set of standard rectangular wave pulse signals is output, with the frequency set to 300Hz. The standard pulse signal undergoes amplitude tuning to form a high-frequency voltage signal with controlled amplitude. This signal is then input to an amplitude modulation circuit, which consists of a proportional regulator, a voltage follower, and a protection clamping channel. The control logic locks the signal amplitude at 2V, the minimum effective amplitude selected under 48V bus power supply conditions. This ensures that the high-frequency response energy effectively excites the motor's inductor response while avoiding interference or high-frequency noise generation caused by high-frequency overshoot. The waveform of the modulated output pulse signal maintains the original timing structure, while the voltage amplitude is stably controlled within the 2V range. To improve the anti-interference capability and electrical isolation safety of the signal transmission in the motor system, this amplitude-controlled pulse signal is processed by an isolation transformer. The isolation transformer employs a high-frequency ferrite core winding structure. Its function is twofold: firstly, to block the DC path between the signal source and the motor drive side, effectively preventing signal feedback interference to the main circuit; and secondly, to enhance the signal current driving capability through a transformer coupling mechanism, enabling stable injection into the motor windings. Simultaneously, after isolation, to ensure that only the target frequency band signal components are retained, the isolated signal undergoes bandpass filtering. The center frequency of the bandpass filter is set at 300Hz, with a bandwidth range of ±20Hz, effectively filtering out high-frequency stray components from system power supply harmonics, electromagnetic interference noise, and PWM secondary-side interference bands, thus outputting an injection signal with high spectral purity. During real-time operation of the control system, the system updates the currently estimated rotor spatial attitude angle information based on the estimated angle provided by the position phase-locked loop in each control cycle, and uses this angle for coordinate transformation calculations. This transformation process is based on a typical Park transform inverter process, rotating the high-frequency injected voltage signal in the stationary reference coordinate system to the currently estimated d-axis direction of the motor in real time, ensuring that the injected signal is consistent with the rotor magnetic pole direction, thereby improving the response sensitivity of the injected signal in the inductive anisotropic direction. After this angle transformation, the injected signal is mapped onto a synchronous rotating coordinate system. In this coordinate system, it is converted into a three-phase injection voltage that can be executed by the drive circuit through SVPWM or equivalent pulse width control channel. The control system periodically applies this high-frequency injection voltage to the d-axis winding of the motor at a frequency of 300Hz, that is, it forms a modulated injection in the actual three-phase current control quantity in the form of superposition or overlay, thereby exciting high-frequency response characteristics related to the magnetic pole position in the inductive response path of the motor.Under the influence of high-frequency injected voltage, the current in the windings of a permanent magnet synchronous motor exhibits a response characteristic corresponding to the frequency. Due to the directional differences in inductance and the anisotropic nature of the rotor magnetic poles, the amplitude and phase of the current excited by the injected signal in the three-phase windings will differ depending on the rotor angle, forming high-frequency current characteristic components that can be identified by the sampling system. The control system uses a high-speed sampling module to collect the three-phase winding current i respectively. a i b i c And by using a digital bandpass filter, the frequency band component centered at 300Hz in each phase is extracted, ultimately forming i a_hf i b_hf i c_hf Three independent high-frequency current signals.

[0028] In one specific embodiment, the process of extracting the high-frequency components of the three-phase winding current of the permanent magnet synchronous motor under the action of pulsating high-frequency injection voltage to obtain the three-phase high-frequency current can specifically include the following steps: The three-phase winding current of a permanent magnet synchronous motor under the action of pulsating high-frequency injected voltage is synchronously sampled to obtain the three-phase current sampling signal; The three-phase current sampling signal is bandpass filtered to obtain the three-phase high-frequency current components. The three-phase high-frequency current components are then lowpass filtered and conditioned to obtain a clean three-phase high-frequency current signal. Amplitude and phase information are extracted from the pure three-phase high-frequency current signal to obtain a three-phase high-frequency current containing rotor position information.

[0029] Specifically, when a 300Hz pulsating high-frequency voltage signal is injected into the estimated d-axis direction through coordinate transformation and applied to the stator windings of a permanent magnet synchronous motor by the PWM controller, current components with different frequency responses will appear in different windings due to the motor's inductance anisotropy and the differences in the distribution of electrical parameters such as stator resistance and inductance. To capture this current response characteristic, the system performs synchronous sampling of the three-phase winding current signals at a high sampling rate under the excitation of the high-frequency injected signal. The controller samples the i-phase current signals at different frequencies. a i b and i cThe three current signals are synchronously captured at preset sampling times within each PWM cycle to ensure phase alignment and time consistency, resulting in three-phase current sampling signals. These signals are then bandpass filtered. The bandpass filter used has a center frequency of 300Hz and a passband width of ±20Hz to cover the slight frequency drift caused by system dynamics and the injection process, while also providing strong suppression of noise components and PWM harmonic frequencies. After bandpass filtering, irrelevant frequency components in each current signal are effectively filtered out, retaining only the high-frequency response concentrated near the injection frequency, thus obtaining frequency-specific three-phase high-frequency current components. The three-phase high-frequency current components are then low-pass filtered and conditioned to remove unwanted high-frequency glitches and restore a stable signal profile. The low-pass filter's cutoff frequency is selected to be slightly higher than twice the target injection frequency to ensure effective smoothing of high-frequency beat frequencies without weakening the main high-frequency components. Next, signal conditioning is performed, including automatic elimination of DC bias, signal amplitude normalization, waveform correction, and discrete reconstruction of sampled values. These operations yield a high-fidelity, pure three-phase high-frequency current signal. Amplitude and phase information are extracted based on this pure three-phase high-frequency current signal. Amplitude information reflects the response intensity of the high-frequency injected signal in different windings, while phase information reflects the relative timing offset between the injected signal and the stator magnetic field or rotor poles. The amplitude characteristics are obtained by calculating the maximum peak value, effective value, or root mean square value of the three-phase high-frequency current in each cycle. Simultaneously, using a synchronous demodulation mechanism, these values ​​are multiplied and integrated with reference sine and cosine signals respectively to extract the orthogonal components between each signal and the injection reference, thereby constructing a phase relationship reflecting the spatial position of the magnetic poles. This is achieved by constructing i a_hf i b_hf with i c_hf By establishing a vector space model among the three components and combining it with coordinate transformation formulas, the spatial vector rotation trajectory of the current response is constructed, and the relative orientation of the current rotor magnetic poles in the synchronous rotating coordinate system is calculated based on this trajectory. The resulting three-phase high-frequency current signal incorporates the amplitude asymmetry and phase difference caused by the change in inductance due to the magnetic pole position in response to the injected signal.

[0030] In one specific embodiment, the process of performing step S2 may specifically include the following steps: The average value and unbalance index of the three-phase high-frequency current are calculated to obtain the unbalance value of the three-phase high-frequency current. The unbalance value of the three-phase high-frequency current is compared with a preset threshold to obtain the target comparison result. Based on the target comparison result, the adaptive compensation coefficient is calculated to obtain the dynamic dead zone compensation coefficient. The dead zone compensation voltage is calculated based on the dynamic dead zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead zone compensation voltage signal. The three-phase dead-zone compensation voltage signal is superimposed on the three-phase high-frequency current for compensation and correction, thus obtaining the high-frequency current after dead-zone compensation.

[0031] Specifically, i is obtained from a permanent magnet synchronous motor that has undergone high-frequency injection processing. a_hf i b_hf i c_hf The pure high-frequency current signals of the three-phase windings have been extracted through processes such as synchronous sampling, bandpass filtering, and low-pass conditioning to remove the current components mainly concentrated around the injection frequency (e.g., 300Hz), while preserving the amplitude and phase response characteristics caused by rotor position changes. Because the power devices in the drive arm of the motor have a fixed dead time under low speed or light load conditions, and this dead time is not fully compensated for in the drive voltage, it will cause current distortion at different phases, manifesting as asymmetry or offset in the three-phase current response. Therefore, this error source is dynamically identified and compensated. The average value of the three-phase high-frequency current signals is calculated, specifically by averaging i... a_hf i b_hf i c_hf The three phase currents are sampled separately within the current sampling period, and then their arithmetic average is calculated to obtain the center value of the three-phase high-frequency current within the current period. Based on this average value, the difference between each phase current and the average value is calculated, squared, and summed to form the unbalance index value UIB, which is used to quantify the consistency of the three-phase current response under high-frequency excitation conditions. The smaller the UIB value, the more symmetrical and balanced the current response tends to be; the larger the UIB value, the more significant the current asymmetric distortion caused by the dead zone effect, especially when the motor is stationary or under slight vibration, this value will be significantly higher than in the moving state. The unbalance index UIB is compared with the dead zone trigger threshold set in the system. This threshold is set to 0.1A and is used as the activation threshold for dead zone distortion identification. If the current UIB value is less than the threshold, the current three-phase signal is determined to be within the acceptable error range, and no additional compensation is required; the dead zone compensation voltage remains at the default value or minimum value. Conversely, if the UIB exceeds the set value, the system activates the adaptive compensation coefficient calculation mechanism and enters the dynamic compensation range. In this mechanism, the controller extracts compensation coefficients from a predefined functional relationship based on the current magnitude and incremental trend of the UIB. This function is established through a polynomial or interpolation model fitted to experimental data and has automatic adjustment capabilities. The core function of the compensation coefficients is to dynamically weight the dead-zone voltage compensation ratio according to the severity of the current imbalance, enabling the system to adaptively correct the current offset under various operating conditions instead of using a uniform static compensation value. The dead-zone compensation voltage is calculated based on the dynamic dead-zone compensation coefficients and the sign function of each phase current. Since the direction of dead-zone voltage compensation depends on the current conduction state of the power transistor, the system determines i in each sampling period. a_hf ib_hf i c_hf The polarity of each phase's current is determined by the sign function, which identifies whether the current is conducting during the positive or negative half-cycle. Only by identifying the conduction direction can the polarity of the compensation voltage be correctly generated. The compensation coefficient is multiplied by the sign function and weighted with the ideal dead-zone voltage reference value to obtain the dead-zone compensation voltage signal V for each phase. dt_a V dt_b V dt_c These three components constitute the three-phase dead-zone compensation vector, which is continuously refreshed with the change in UIB during each control cycle, possessing dynamic response capability. The aforementioned three-phase dead-zone compensation voltage signals are superimposed onto the original three-phase high-frequency current signals to achieve real-time compensation and correction of the high-frequency current. During the superposition process, the sampling, calculation, and compensation processes are kept synchronized to avoid timing mismatches causing cumulative compensation errors. The high-frequency current signal after compensation is the current response after dead-zone compensation.

[0032] In one specific embodiment, the process of calculating the dead-zone compensation voltage based on the dynamic dead-zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead-zone compensation voltage signal can specifically include the following steps: The sign of the three-phase high-frequency currents is determined to obtain the sign functions of the A-phase, B-phase, and C-phase currents; The dead-time voltage is calculated based on the preset ideal dead-time and PWM switching frequency to obtain the ideal dead-time compensation voltage reference value. The dynamic dead zone compensation coefficient is multiplied by the ideal dead zone compensation voltage reference value to obtain the compensation voltage amplitude. The three-phase dead-zone compensation voltage signal is obtained by performing sign matching and amplitude synthesis between the compensation voltage amplitude and the sign functions of the currents in phases A, B, and C, respectively.

[0033] Specifically, a three-phase signal polarity determination mechanism based on the current sampling period is constructed to determine the conduction direction and power transistor operating state of each phase. The system utilizes the pure high-frequency current signal i before three-phase dead-time compensation. a_hf i b_hf i c_hfThe system sequentially determines the positive and negative polarity of the current values ​​within the control cycle. A standard symbol function extraction algorithm is used to perform polarity conversion logic on each phase current value: +1 is output when the current value is greater than zero, -1 is output when the current value is less than zero, and zero is defaulted to the previous cycle state or defined as 0 to avoid calculation uncertainties. Through this determination process, the system obtains the current symbol functions for phases A, B, and C respectively. These three function values ​​are updated in each control cycle to track real-time changes in current direction. After constructing the symbol functions, the system derives the voltage compensation reference value based on the preset ideal dead time and the ratio between the ideal dead time and the PWM switching frequency. Dead time refers to the intentionally left short non-conducting time between the upper and lower bridge arm switches in the driver to avoid short-circuit breakdown risk; its typical setting range is 1μs to 3μs. The PWM switching frequency is the carrier frequency during system modulation operations, ranging from 10kHz to 20kHz. Based on the dead time and PWM frequency pre-set in the control system, the equivalent impact of dead time on the drive voltage, i.e., the ideal dead-time compensation voltage reference value, is obtained through mathematical conversion. This value is pre-set in the controller as a fixed reference through simulation modeling or experimental calibration, or dynamically calculated based on real-time PWM parameters, to reflect the voltage compensation amplitude required to completely offset the dead-time window effect under ideal conditions. The dynamic dead-time compensation coefficient calculated in the previous stage is multiplied by the ideal dead-time compensation voltage reference value to obtain the dead-time compensation voltage amplitude under the current control cycle. The multiplication process is implemented in hardware using a fixed-point multiplier to ensure that the calculation speed and amplitude accuracy meet the real-time requirements under high-frequency control cycles. The compensation voltage amplitude is matched and synthesized with the sign functions corresponding to phases A, B, and C to form three dead-time compensation voltage signals with directional and amplitude modulation capabilities. During the compensation voltage synthesis process, each phase is updated synchronously, and its output is denoted as V. dt_a V dt_b V dt_c , representing the voltage correction amounts applied to phases A, B, and C, respectively.

[0034] In one specific embodiment, the process of performing step S3 may specifically include the following steps: The high-frequency current input position phase-locked loop after dead-zone compensation is synchronously multiplied and demodulated with the reference sine signal and the reference cosine signal to obtain the demodulated signal. The demodulated signal is low-pass filtered and orthogonal component separation is performed to obtain the first position detection component and the second position detection component; The rotor position error angle is obtained by performing arctangent function calculation based on the first position detection component and the second position detection component. The rotor position error angle is input into a variable gain PI controller for segmented gain adjustment and closed-loop control to obtain the rotor position angle.

[0035] Specifically, the high-frequency current after dead-zone compensation is used as the main input and synchronously multiplied with two reference signals. These two reference signals are a sinusoidal reference signal and a cosine reference signal constructed internally by the system using a high-frequency injection fundamental frequency. They are orthogonal, with constant frequency and stable phase. During synchronous multiplication, the high-frequency current signal is projected onto these two standard orthogonal reference bases to extract the modulation components related to position information. The two product signals obtained after demodulation still contain beat frequency components related to the injection frequency and other high-frequency harmonic interference. Therefore, low-pass filters are introduced to process them separately. The cutoff frequency of the filter needs to be slightly higher than the expected mechanical change frequency of the motor to completely retain the low-frequency signal generated during the slow change of rotor angle, while effectively suppressing unwanted high-frequency noise. After low-pass filtering, two stable orthogonal low-frequency signals are obtained, representing the projection intensity of the high-frequency current in the reference sine and reference cosine directions, respectively. Angle calculation is performed using these two signals. The phase relationship between the two orthogonal signals is established through a standard arctangent function to calculate the current position error angle. This angle represents the deviation between the currently estimated rotor position and the actual magnetic pole position, and is a key parameter for the controller to determine whether the true position has been locked. Because this angle exhibits significant dynamic changes at different operating stages, it is processed by a closed-loop controller with piecewise gain adjustment capability. This controller automatically adjusts its control parameters based on the absolute magnitude of the angle error. When the error is large, it increases the proportional gain to accelerate the response speed, while decreasing the gain when the error is small to avoid oscillation or misadjustment. It also tracks the cumulative offset trend through a corresponding integral gain to stabilize the control result. After processing the position error angle, the controller outputs the rotor position angle value estimated for the current cycle.

[0036] In one specific embodiment, the process of performing arctangent function calculation based on the first position detection component and the second position detection component to obtain the rotor position error angle can specifically include the following steps: The amplitude of the first position detection component and the second position detection component are checked and preprocessed with division by zero protection to obtain the orthogonal position detection component within the safe range; The orthogonal position detection components within the safe range are input into the four-quadrant arctangent function to calculate the angle, thus obtaining the initial position error angle; Quadrant correction and angle range adjustment are performed based on the sign and magnitude of the initial position error angle to obtain the corrected position error angle; The rotor position error angle is obtained by applying precision limits and numerical stability filtering to the correction position error angle.

[0037] Specifically, amplitude verification is performed on the first and second position detection components. This involves comparing the absolute values ​​of each component and setting a minimum amplitude threshold as the lower limit for valid signal recognition. When the absolute value of any position detection component is detected to be lower than the preset amplitude safety threshold (e.g., 0.005), the system activates a division-by-zero protection mechanism. This clamps the component value to a smaller but non-zero safe alternative value, or performs interpolation smoothing based on the state of the previous control cycle, to avoid directly inputting a zero value into the trigonometric function processing path. After amplitude detection and division-by-zero preprocessing, a pair of orthogonal position detection components with numerical controllability and amplitudes within the defined range are obtained. The orthogonal position detection components within the safe range are sent to the four-quadrant arctangent function processing module to perform angle calculation. The four-quadrant arctangent function can identify the specific region where the input component is located in the first quadrant (positive-positive), second quadrant (negative-positive), third quadrant (negative-negative), and fourth quadrant (positive-negative), and return the accurate corresponding spatial angle value, with the angle range between -π and +π. This processing step yields an initial position error angle relative to the reference injected signal base direction. This angle reflects the spatial deviation of the motor's magnetic poles relative to the reference orthogonal base in the rotating coordinate system. Quadrant correction and angle range adjustment are performed based on the sign and magnitude of the initial position error angle. Due to the requirements for angle compression, mapping of positive and negative torque polarity intervals, and optimization of calculation accuracy in the control system of permanent magnet synchronous motors, the quadrant of the initial angle is determined based on its sign. According to the coordinate system logic rules set internally by the controller, the angle value is converted to an effective control range under a unified reference coordinate system. For example, when the angle is in the negative range but close to the -π boundary, the system adds 2π to pull it back to the 0-2π range; if the angle exceeds the upper limit of π but does not exceed 2π, it is mapped back to the -π to π control range by subtracting 2π. Furthermore, the system determines whether reverse rotation mapping or symmetrical conversion is needed based on the magnitude of the initial angle to ensure that the final angle remains consistent with the actual physical rotor pole position. The angle obtained through the above quadrant determination, sign determination, and interval conversion operations is the corrected position error angle. The accuracy limit and numerical stability filtering of the corrected position error angle are applied to prevent small amplitude disturbances from causing control command oscillations or discontinuous jumps in the response curve. For accuracy limiting, the system introduces an angle change rate threshold or a maximum single-cycle change amplitude limiter. When the angle change amplitude exceeds this threshold, it is determined as a sudden abnormal data point, and the previous cycle value is held, or interpolation delay smoothing is used to gradually transition the actual output value to the new angle to ensure response consistency. For numerical stability filtering, methods such as moving average, exponential weighted filtering, or low-pass recursive filtering are used to combine the angle estimates from nearly several control cycles into a stable output, ultimately yielding the rotor position error angle.

[0038] In one specific embodiment, the process of performing step S4 may specifically include the following steps: The absolute value of the position error is calculated based on the rotor position angle and compared with the radian threshold to obtain a position locking success confirmation signal; Based on the successful confirmation signal of position locking, a three-stage q-axis current establishment strategy is initiated and segmented current increment processing is performed to obtain a segmented q-axis current command that includes preload stage current, acceleration stage current and steady-state adjustable current. The d-axis current is set to zero and kept constant to obtain a d-axis zero current command signal. Based on load current change rate monitoring, and when the change rate exceeds the target current value, the current slope is adaptively adjusted to obtain a fast current slope command. Based on the fast current slope command, the phased q-axis current command and d-axis zero current command are combined by coordinate synthesis and torque conversion to obtain the starting torque command.

[0039] Specifically, the rotor position estimation results undergo effective error analysis and judgment. After receiving the rotor position angle output from the phase-locked loop structure, the control system calculates the trend of this angle's change within consecutive control cycles and performs a difference calculation with the estimation result of the previous cycle to obtain the absolute value of the position error in the current control cycle. This error value serves as the primary basis for position locking determination and is compared with a pre-set radian threshold in the system. The threshold value is set to π / 36, approximately 5 degrees, to define whether the change in rotor angle within a short period is within an acceptable locking accuracy range. If the absolute value of the position error in multiple consecutive control cycles is less than this radian threshold, it indicates that the rotor angle estimation has stabilized, with small fluctuations and high response accuracy. Based on this, the system generates a position locking success confirmation signal and uses this signal as the trigger condition for starting the next stage of the control process. The generation logic of this confirmation signal incorporates a time criterion, requiring that the duration for which the error value meets the threshold condition exceeds a set time window, such as 10ms, to ensure that the locking is not caused by short-term accidental jitter, thereby avoiding misjudgment. Once the position lock success signal is activated, the system initiates a three-stage q-axis current establishment strategy. This strategy allocates target current values ​​and incremental growth rates across different time intervals to smoothly and linearly establish the drive current during startup, outputting a three-segment q-axis current command curve. The first stage is the pre-loading stage, where the goal is to gradually increase the q-axis current from 0 to 1A with a small slope. This overcomes the initial static friction of the permanent magnet synchronous motor in a stationary state and the inertial load of the wheel hub system, ensuring the system has basic driving capability during the transition from zero speed to micro-rotation. The second stage is the acceleration stage, where the q-axis current continues to increase linearly with a moderate slope to approximately 2.5A, forming the main acceleration drive torque to complete the kinetic energy conversion required for vehicle start-up. The third stage is the steady-state stage, where the current no longer increases at a fixed rate but is dynamically fine-tuned based on the current load, wheel speed trends, and torque demand targets. This keeps the q-axis current fluctuating slightly around 2.5A or adjusting it to higher or lower levels as needed, thus achieving continuous control after stable operation. Simultaneously, to maximize torque output efficiency in the flux direction, the system applies a zero-value setting to the d-axis current channel throughout the entire q-axis current establishment process and maintains this constant value through the controller's closed-loop channel structure. That is, regardless of changes in the current q-axis current, the d-axis current is always set to 0 and remains static, ensuring that all control energy is concentrated on the q-axis direction that generates the driving torque, without introducing flux disturbances or wasted energy. This d-axis zero-current command signal and the q-axis current form a current vector control basis in a synchronously rotating coordinate system, constituting the primary reference source for torque conversion and control law output. During the q-axis current establishment process, the system continuously monitors the rate of change of the current q-axis current to determine in real time whether a sudden change in the load state occurs.For example, when an electric scooter encounters uphill, steep slopes, heavy loads, or complex road surface changes, the torque demand of the load will cause a rapid increase in the q-axis current. To adapt to this situation, the system constructs a load change rate detector. It calculates the current difference between the current control cycle and the previous cycle by dividing the value by the sampling period, forming a real-time current change rate value, which is then compared with a preset target change rate threshold, set at 5A / s. If the detected actual change rate exceeds the threshold, the system automatically adjusts the slope of the current establishment curve of the current in the current phase, increasing the current increment rate of the first and second stages, for example, from 0.5A / 15ms to 1A / 10ms. This allows the motor to quickly build sufficient torque under high load conditions without experiencing step loss, oscillation, or starting failure. This slope adjustment mechanism creates a dynamic response capability to actual load changes, improving the adaptability of the sensorless control strategy and the start-up success rate. The three-stage q-axis current command and d-axis zero current command are input into the coordinate synthesis module. Through orthogonal rotation coordinate transformation, they are mapped from the dq coordinate system back to the stationary three-phase coordinate system abc. Combined with the current position angle, spatial voltage vector conversion is performed to form the three-phase drive voltage target. At the same time, the system synchronously performs the projection calculation of current onto electromagnetic torque based on magnetic pole direction in this transformation path to complete the mapping of current command to torque command, thereby obtaining the starting torque command.

[0040] In one specific embodiment, the process of performing step S5 may specifically include the following steps: The starting torque command is input to the drive controller of the permanent magnet synchronous motor for PWM modulation and three-phase drive processing to obtain the three-phase drive voltage that drives the permanent magnet synchronous motor to rotate. Based on the rotor position angle, a multi-point difference algorithm is used for calculation and numerical differentiation to obtain the real-time detection value of the hub speed. The real-time detected value of the wheel hub speed is compared with the preset speed threshold and the time is confirmed to obtain a speed compliance confirmation signal; Based on the speed target confirmation signal, the back EMF observer is started and runs in parallel with the high-frequency injection position detector and the position error is compared to obtain the mode switching ready signal. Based on the mode switching readiness signal, the amplitude of the high-frequency injection signal is linearly reduced from 2V to 0V and the back-cut threshold monitoring is activated at the same time. The output includes a dual-mode control switching signal that includes forward switching and back-cut protection.

[0041] Specifically, the starting torque command is input to the drive controller of the permanent magnet synchronous motor for PWM modulation. The PWM modulator, based on the target current magnitude and the system-set power bus voltage, uses space vector pulse width modulation (SPWM) or three-phase symmetrical sinusoidal pulse width modulation (SPWM) to convert the target current into PWM control pulse width signals for the three-phase bridge arms. This PWM signal directly acts on the power switching transistors of the motor drive controller, adjusting the actual applied drive voltage in the three-phase windings within each PWM cycle by controlling the on-time, thereby forming a three-phase voltage waveform consistent with the desired current. This set of three-phase voltages generated by PWM modulation drives the stator to generate a rotating magnetic field in the electromagnetic field, and guides the rotor to start and accelerate rotation in the desired manner by controlling the direction and frequency, constituting the electromagnetic energy realization path for the entire motor starting process. As the motor begins to rotate, the system monitors the actual rotational speed of the hub in real time to determine whether the conditions for switching from high-frequency injection mode to back EMF observation mode are met. Since this stage is still sensorless, the system cannot obtain wheel speed information through direct measurement; therefore, indirect speed estimation is performed based on the rotor angle output by the position phase-locked loop (PLL). The controller employs a multi-point differential method to numerically differentiate the rotor angle θ within a continuous cycle. By constructing a five-point central difference formula, such as using θ(k), θ(k-1), θ(k-2), θ(k-3), and θ(k-4) as inputs, and combining this with the sampling period, a speed approximation calculation is performed to obtain an estimated hub speed for the current cycle. The real-time speed estimate is compared with a preset speed threshold to determine if the current speed meets the basic conditions for switching to the back EMF observation mode. This threshold is set to 100 rpm. During the judgment process, the system requires the current wheel speed to be higher than this threshold and maintain this state for a confirmation time, such as 50 ms, to eliminate the possibility of falsely high speeds caused by short-term disturbances or measurement errors. If the speed consistently meets the conditions, the system issues a speed compliance confirmation signal. Upon receiving the speed compliance confirmation signal, the control system activates the back EMF observer. This observer extracts back EMF information by detecting the relationship between the voltage and current at the motor winding terminals and calculates the current position of the motor rotor accordingly. Essentially, it is a sensorless estimation method based on a flux linkage model or a back EMF integral model. Simultaneously, the system does not immediately shut down the high-frequency injection mode at this stage. Instead, it runs the back EMF observer and the high-frequency injection position detector in parallel. The two position estimation channels output their respective angle estimates. The system compares these two sets of estimates to calculate the angle deviation between them. When the angle error between the two estimation methods is detected to be less than the set allowable value (e.g., ±5°), it indicates that the back EMF estimation has achieved a positioning capability equivalent to high-frequency injection, and the system generates a "mode switching ready signal." This signal signifies that the motor is in a switchable state, and that switching will not cause problems such as control discontinuity, response jumps, or positioning loss due to different estimation methods.After the mode switching readiness signal is activated, the system implements a linear amplitude reduction strategy control for the high-frequency injection signal channel, gradually reducing the injection amplitude, originally set at 2V, to 0V with a linear slope within a predetermined period, thereby achieving a "soft exit" of the injection process. This method avoids current disturbances or system oscillations caused by abrupt termination of the injection signal, ensuring a smooth transition of the control state from injection drive to back EMF estimation while maintaining the electromagnetic stability of motor operation. Simultaneously, while performing linear attenuation, the system activates the back-cut threshold monitoring channel to continuously track the wheel hub speed. If the wheel speed is detected to drop below 80rpm again during operation, i.e., below the switching lower limit, the system automatically restores the high-frequency injection path and restarts the phase-locked position detector, causing the control strategy to automatically switch back from back EMF observation to high-frequency injection mode. The system output includes a dual-mode control switching signal containing a "forward switching control signal" (indicating successful switching from high-frequency injection to back EMF estimation mode) and a "back-cut protection control signal" (indicating low-speed return to high-frequency injection mode).

[0042] The above describes the motor control method of the sensorless control system in the embodiments of the present invention. The following describes the motor control system of the sensorless control system in the embodiments of the present invention. Please refer to [link / reference]. Figure 2 One embodiment of the sensorless motor control system of the present invention includes: The injection module is used to inject pulse voltage signals into the estimated d-axis of the permanent magnet synchronous motor and extract the three-phase high-frequency current. The compensation module is used to perform adaptive dead-zone compensation based on the three-phase high-frequency current to obtain the high-frequency current after dead-zone compensation. The control module is used to synchronously demodulate and PI control the high-frequency current input to the position phase-locked loop after dead-zone compensation to obtain the rotor position angle. A module is established to generate the q-axis starting current based on the rotor position angle, thereby obtaining the starting torque command. The output module is used to execute the starting torque command to drive the permanent magnet synchronous motor to rotate and to detect the hub speed in real time. When the speed reaches the preset target value, it performs the switching process from high frequency injection mode to back EMF observation mode and outputs a dual-mode control switching signal.

[0043] Through the coordinated efforts of the aforementioned components, and by precisely designing a 300Hz injection frequency and a 2V injection amplitude, position detection accuracy under low-voltage systems is ensured while avoiding the impact of high-frequency noise on the riding experience. This solves the problem of low signal-to-noise ratio in existing high-frequency injection methods for low-power hub motors. By real-time detection of three-phase high-frequency current imbalance and dynamic adjustment of the compensation coefficient, the position estimation drift problem caused by the amplified dead-zone effect of low-power motors under light loads is effectively eliminated, significantly improving starting reliability. An improved PI control structure and variable gain strategy are adopted, compressing the position convergence time to 30ms, meeting scooter users' expectations for rapid start-up response and improving the riding experience. A three-stage linear slope current establishment method ensures sufficient starting torque while avoiding current surges, achieving a smooth and rapid start and solving the starting shock and vibration problems caused by traditional step torque setting. The problem of switching point drift under different loads, a problem inherent in traditional voltage detection methods, is avoided, ensuring the accuracy and stability of control mode switching and improving control performance throughout the entire operating range. By monitoring load changes in real time and adaptively adjusting control parameters, and being optimized for high-load conditions such as uphill runs, the problem of step loss during variable load startup in traditional sensorless control has been solved, thus improving the startup success rate.

[0044] Those skilled in the art will clearly understand that, for the sake of convenience and brevity, the specific working processes of the systems and units described above can be referred to the corresponding processes in the foregoing method embodiments, and will not be repeated here.

[0045] If the integrated unit is implemented as a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of the present invention, in essence, or the part that contributes to the prior art, or all or part of the technical solution, can be embodied in the form of a software product. This computer software product is stored in a storage medium and includes several instructions to cause a computer device (which may be a personal computer, server, or network device, etc.) to execute all or part of the steps of the methods described in the various embodiments of the present invention. The aforementioned storage medium includes various media capable of storing program code, such as USB flash drives, portable hard drives, read-only memory (ROM), random access memory (RAM), magnetic disks, or optical disks.

[0046] The above-described embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A motor control method using a sensorless control system, characterized in that, include: Pulsed voltage signal injection processing is performed on the estimated d-axis of the permanent magnet synchronous motor, and three-phase high-frequency current is extracted. Adaptive dead-zone compensation is performed based on the three-phase high-frequency current to obtain the dead-zone compensated high-frequency current. Specifically, this includes: calculating the average value and unbalance index of the three-phase high-frequency current to obtain the unbalance value of the three-phase high-frequency current; comparing the unbalance value of the three-phase high-frequency current with a preset threshold to obtain a target comparison result, and calculating the adaptive compensation coefficient based on the target comparison result to obtain the dynamic dead-zone compensation coefficient; calculating the dead-zone compensation voltage according to the dynamic dead-zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead-zone compensation voltage signal; and superimposing the three-phase dead-zone compensation voltage signal onto the three-phase high-frequency current for compensation correction to obtain the dead-zone compensated high-frequency current. The high-frequency current input position phase-locked loop after dead-zone compensation is synchronously demodulated and PI controlled to obtain the rotor position angle. The q-axis starting current is established based on the rotor position angle to obtain the starting torque command; The starting torque command is executed to drive the permanent magnet synchronous motor to rotate and the hub speed is detected in real time. When the speed reaches the preset target value, the high-frequency injection mode is switched to the back electromotive force observation mode, and a dual-mode control switching signal is output.

2. The motor control method using a sensorless control system according to claim 1, characterized in that, The process of injecting pulsating voltage signals into the estimated d-axis of the permanent magnet synchronous motor and extracting the three-phase high-frequency current includes: A standard pulsed fundamental frequency signal is generated by the internal timer of the DSP controller; The standard pulse fundamental frequency signal is input into the amplitude modulation circuit for 2V amplitude control to obtain an amplitude-controlled pulse voltage signal; The amplitude-controlled pulsating voltage signal is isolated by an isolation transformer and bandpass filtered to obtain a clean high-frequency injection signal; Based on the current rotor position angle, the pure high-frequency injection signal is transformed to the estimated d-axis coordinate system and injected to obtain the pulsating high-frequency injection voltage acting on the d-axis winding of the permanent magnet synchronous motor. The high-frequency components of the three-phase winding current of the permanent magnet synchronous motor under the action of the pulsed high-frequency injection voltage are extracted to obtain the three-phase high-frequency current.

3. The motor control method using a sensorless control according to claim 2, characterized in that, The extraction of high-frequency components of the three-phase winding current of the permanent magnet synchronous motor under the action of the pulsed high-frequency injection voltage to obtain the three-phase high-frequency current includes: The three-phase winding current of the permanent magnet synchronous motor under the pulsed high-frequency injection voltage is synchronously sampled to obtain the three-phase current sampling signal. The three-phase current sampling signal is bandpass filtered to obtain three-phase high-frequency current components, and the three-phase high-frequency current components are lowpass filtered and signal conditioned to obtain a clean three-phase high-frequency current signal. Amplitude and phase information are extracted based on the pure three-phase high-frequency current signal to obtain a three-phase high-frequency current containing rotor position information.

4. The motor control method using a non-contact control sensor according to claim 1, characterized in that, The step of calculating the dead-zone compensation voltage based on the dynamic dead-zone compensation coefficient and the sign function of each phase current to obtain the three-phase dead-zone compensation voltage signal includes: The sign of the three-phase high-frequency currents is determined to obtain the sign functions of the A-phase, B-phase, and C-phase currents; The dead time voltage is calculated based on the preset ideal dead time and PWM switching frequency to obtain the ideal dead time compensation voltage reference value. The dynamic dead zone compensation coefficient is multiplied by the ideal dead zone compensation voltage reference value to obtain the compensation voltage amplitude. The three-phase dead zone compensation voltage signal is obtained by performing sign matching and amplitude synthesis with the sign functions of the currents in phases A, B, and C, respectively, based on the compensation voltage amplitude.

5. The motor control method using a sensorless control system according to claim 1, characterized in that, The step of synchronously demodulating and PI controlling the high-frequency current input position phase-locked loop after dead-zone compensation to obtain the rotor position angle includes: The high-frequency current input position phase-locked loop after dead-zone compensation is synchronously multiplied and demodulated with the reference sine signal and the reference cosine signal to obtain the demodulated signal. The demodulated signal is subjected to low-pass filtering and quadrature component separation to obtain a first position detection component and a second position detection component; The rotor position error angle is obtained by performing an arctangent function operation based on the first position detection component and the second position detection component. The rotor position error angle is input into a variable gain PI controller for segmented gain adjustment and closed-loop control to obtain the rotor position angle.

6. The motor control method using a non-contact control sensor according to claim 5, characterized in that, The step of performing arctangent function calculation based on the first position detection component and the second position detection component to obtain the rotor position error angle includes: The first position detection component and the second position detection component are subjected to amplitude verification and division-by-zero protection preprocessing to obtain orthogonal position detection components within a safe range; The orthogonal position detection components within the safe range are input into the four-quadrant arctangent function to calculate the angle, thus obtaining the initial position error angle. Based on the sign and magnitude of the initial position error angle, quadrant correction and angle range adjustment are performed to obtain the corrected position error angle; The rotor position error angle is obtained by performing precision limiting and numerical stability filtering on the corrected position error angle.

7. The motor control method using a sensorless control according to claim 1, characterized in that, The step of establishing the q-axis starting current based on the rotor position angle to obtain the starting torque command includes: The absolute value of the position error is calculated based on the rotor position angle and compared with the radian threshold to obtain a position locking success confirmation signal. Based on the successful confirmation signal of position locking, a three-stage q-axis current establishment strategy is initiated and segmented current increment processing is performed to obtain a segmented q-axis current command that includes preload stage current, acceleration stage current and steady-state stage adjustable current. The d-axis current is set to zero and kept constant to obtain a d-axis zero current command signal. Based on load current change rate monitoring, and when the change rate exceeds the target current value, the current slope is adaptively adjusted to obtain a fast current slope command. Based on the fast current slope command, the phased q-axis current command and the d-axis zero current command are combined in coordinates and torque is calculated to obtain the starting torque command.

8. The motor control method using a non-contact control sensor according to claim 1, characterized in that, The process executes the starting torque command to drive the permanent magnet synchronous motor to rotate and monitors the hub speed in real time. When the speed reaches a preset target value, it switches from high-frequency injection mode to back EMF observation mode and outputs a dual-mode control switching signal, including: The starting torque command is input to the drive controller of the permanent magnet synchronous motor for PWM modulation and three-phase drive processing to obtain the three-phase drive voltage that drives the permanent magnet synchronous motor to rotate. Based on the rotor position angle, a multi-point difference algorithm is used for calculation and numerical differentiation to obtain the real-time detection value of the hub speed. The real-time detected value of the wheel hub speed is compared with the preset speed threshold and the time is confirmed to obtain a speed compliance confirmation signal; Based on the speed target confirmation signal, the back EMF observer is started and runs in parallel with the high-frequency injection position detector and the position error is compared to obtain the mode switching ready signal. Based on the mode switching readiness signal, the amplitude of the high-frequency injection signal is linearly reduced from 2V to 0V, and the back-cut threshold monitoring is started at the same time. The output is a dual-mode control switching signal that includes forward switching and back-cut protection.

9. A sensorless motor control system, characterized in that, A motor control method for performing the non-sensory control sensor as described in any one of claims 1-8, comprising: The injection module is used to inject pulse voltage signals into the estimated d-axis of the permanent magnet synchronous motor and extract the three-phase high-frequency current. The compensation module is used to perform adaptive dead-zone compensation based on the three-phase high-frequency current to obtain the high-frequency current after dead-zone compensation. The control module is used to synchronously demodulate and PI control the high-frequency current input position phase-locked loop after dead-zone compensation to obtain the rotor position angle. A module is established to generate the q-axis starting current based on the rotor position angle, thereby obtaining the starting torque command. The output module is used to execute the starting torque command to drive the permanent magnet synchronous motor to rotate and to detect the hub speed in real time. When the speed reaches the preset target value, it performs the switching process from high frequency injection mode to back electromotive force observation mode and outputs a dual-mode control switching signal.

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