Fault signal dual-mode output method of a magnetic latching relay

Abstract

The application discloses a kind of fault signal double-mode output methods of magnetic latching relay, specifically relates to relay control technical field, for solving the mode switching efficiency low problem caused by existing physical jumper mechanism;By receiving remote double-mode switching instruction, the current mechanical state of relay is detected;According to state, the drive voltage sequence corresponding to target mode is applied;Based on the dynamic margin of the residual magnetism intensity of iron core and target mode nominal value, drive enable or demagnetization trigger signal is generated;When responding to drive enable signal, the lag time difference of voltage rising edge and magnetic flux jump starting point is captured in real time, and the product integral of current differential and temperature rise is combined to generate eddy current loss characteristic quantity;According to the situation of lag time difference and eddy current loss characteristic quantity over limit, the rising slope and duty cycle of drive voltage sequence are adjusted;Finally, the matching of execution result and target mode identification is verified by controlling coil current response waveform, and remote reliable switching continuous holding mode and instantaneous pulse mode are realized.

Description

Technical Field

[0001] This invention relates to the field of relay control technology, and in particular to a method for dual-mode output of fault signals for a magnetic latching relay. Background Technology

[0002] In the field of power distribution network fault monitoring, magnetic latching relays are widely used in the alarm signal output of fault indicators due to their self-holding characteristics in the power-off state. In the existing technology, the dual-mode output function (continuous holding mode and instantaneous pulse mode) of magnetic latching relays is preset through physical jumper devices. Specifically, operators need to adjust the position of the hardware jumper cap on site to pre-fix the relay's operating mode. This method has been used in actual products (such as panel-type short-circuit fault indicators) for many years.

[0003] However, the physical jumper mechanism restricts mode switching operations to manual intervention on-site: when the output mode needs to be switched due to upgrades to distribution network automation terminals or changes in operation and maintenance strategies, the cabinet must be opened to adjust the position of the jumper cap. This operation poses safety risks in a live environment, and frequent disassembly and assembly of equipment reduces operation and maintenance efficiency. More fundamentally, there is a structural contradiction between the physical control channel of the magnetic latching relay and the dynamic functional requirements—the static configuration characteristics of the hardware jumper cannot respond to the needs of remote real-time control, which restricts the flexible evolution of the distribution automation system. Summary of the Invention

[0004] This invention addresses the technical problems existing in the prior art by providing a dual-mode output method for fault signals of a magnetic latching relay.

[0005] The technical solution of the present invention to solve the above-mentioned technical problems is as follows: This invention provides the following technical solution: A method for dual-mode output of fault signals for a magnetic latching relay includes: S1. Receive a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detect the current mechanical state of the magnetic latching relay; S2. Apply a drive voltage sequence corresponding to the target mode identifier to the control coil according to the current mechanical state; S3. Based on the dynamic margin judgment result of the iron core residual magnetism intensity and the nominal value corresponding to the target mode identifier, generate a drive enable signal or a demagnetization trigger signal. S4. In response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by the product of the current derivative and the temperature rise; in response to the demagnetization trigger signal, the drive enable signal is regenerated after automatic demagnetization is performed. S5. If the lag time difference exceeds the first threshold range of the target mode or the eddy current loss characterization exceeds the second threshold range, then adjust the rising slope and duty cycle of the driving voltage sequence. S6. Verify whether the execution result matches the target mode identifier by measuring the current response waveform generated by the adjusted drive voltage sequence in the control coil.

[0006] Furthermore, it receives a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detects the current mechanical state of the magnetic latching relay, including: Receive dual-mode switching commands transmitted by remote control terminals via power line carrier communication, and parse the target mode identifier in the dual-mode switching commands; The real-time position of the armature of the magnetic latching relay is detected by a position sensor; If the armature is in the reset position, the current mechanical state is determined to be the reset state; If the armature is in the motion hold position, the current mechanical state is determined to be the motion hold state.

[0007] Furthermore, a drive voltage sequence corresponding to the target mode identifier is applied to the control coil according to the current mechanical state, including: If the current mechanical state is reset, then the drive voltage sequence corresponding to the target mode identifier is directly applied to the control coil; If the current mechanical state is in the motion holding state, a reverse pulse current is first applied to the control coil to perform a reset operation; After the reset operation is completed, a drive voltage sequence corresponding to the target mode identifier is applied to the control coil.

[0008] Furthermore, based on the dynamic margin judgment result of the remanent magnetization of the iron core and the nominal value corresponding to the target mode identifier, a drive enable signal or a demagnetization trigger signal is generated, including: The remanence of the iron core is measured using a Hall effect sensor; Read the nominal value of remanence intensity corresponding to the target mode identifier from the pre-stored nominal value table; Calculate the absolute deviation rate between the remanence intensity and the nominal value of the remanence intensity; If the absolute deviation rate is less than or equal to the preset margin threshold, a drive enable signal is generated; If the absolute deviation rate is greater than the preset margin threshold, a demagnetization trigger signal is generated.

[0009] Furthermore, the pre-stored nominal value table is generated in the following way: Calibration experiments were conducted on the continuous hold mode and the instantaneous pulse mode respectively; In the calibration experiment, the remanence of the iron core is measured when the magnetic flux required to reach the target mode is measured. Store the remanence value corresponding to the continuous hold mode as the first nominal value; Store the remanence value corresponding to the instantaneous pulse mode as the second nominal value; A mapping table between target mode identifiers and nominal values ​​is established and written into the EEPROM memory to obtain a pre-stored nominal value table; the nominal values ​​include a first nominal value and a second nominal value.

[0010] Furthermore, in response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by integrating the product of the current derivative and the temperature rise, including: The magnetic flux change rate monitoring window is triggered at the start of the rising edge of the driving voltage. Magnetic flux change rate data are continuously collected using a Hall sensor; When the rate of change of magnetic flux exceeds the preset jump threshold, mark the current moment as the start time of the magnetic flux jump. The time difference between the start of the rising edge of the driving voltage and the start of the magnetic flux jump is calculated as the hysteresis time difference. The current differentiation circuit and temperature acquisition circuit are started synchronously during the application of the driving voltage sequence; The output signal of the current differential operation circuit and the output signal of the temperature acquisition circuit are input into the analog multiplier. The output signal of the analog multiplier is input into the integral circuit to generate the eddy current loss characterization quantity.

[0011] Furthermore, the output signal of the analog multiplier is input into the integrator circuit to generate a characteristic quantity of eddy current loss, which is achieved in the following way: The output voltage signal of the analog multiplier is connected to the integration circuit consisting of an operational amplifier and an integrating capacitor. Set the integration time window to cover the entire application period of the driving voltage sequence; The output voltage signal of the analog multiplier is continuously integrated over time using an integration circuit. At the end of the integration time window, the voltage across the integrating capacitor is read as a characterization of eddy current loss.

[0012] Furthermore, in response to the demagnetization trigger signal, automatic demagnetization is performed and a drive enable signal is regenerated, including: Apply a reverse pulse current of a preset duration to the control coil; The drive enable signal is regenerated after the reverse pulse current ends.

[0013] Furthermore, if the hysteresis time difference exceeds the first threshold range of the target mode or the eddy current loss characterization exceeds the second threshold range, the rising slope and duty cycle of the driving voltage sequence are adjusted, including: When the hysteresis time difference is greater than the upper limit of the first threshold of the target mode, increase the rising slope of the driving voltage sequence; When the hysteresis time difference is less than the lower limit of the first threshold of the target mode, reduce the rising slope of the driving voltage sequence; When the eddy current loss characterization is greater than the upper limit of the second threshold of the target mode, reduce the duty cycle of the driving voltage sequence; When the eddy current loss characterization is less than the second threshold lower limit of the target mode, increase the duty cycle of the driving voltage sequence.

[0014] Furthermore, by measuring the current response waveform generated by the regulated drive voltage sequence in the control coil, the execution result is verified to match the target mode identifier, including: The real-time current signal at both ends of the control coil is acquired to generate a current response waveform; If the target mode is identified as continuous hold mode, detect whether the current response waveform maintains a steady current higher than the pull-in threshold for a preset duration; if and only if the current response waveform is detected to maintain a steady current higher than the pull-in threshold for a preset duration, it is determined that the current response waveform characteristics meet the requirements of continuous hold mode. If the target mode is identified as instantaneous pulse mode, detect whether the current response waveform has a single overshoot spike within the preset pulse width and the spike amplitude exceeds the action threshold; if and only if the current response waveform has a single overshoot spike within the preset pulse width and the spike amplitude exceeds the action threshold, it is determined that the current response waveform characteristics meet the instantaneous pulse mode requirements. The current response waveform characteristics are deemed to meet the target mode identification requirements if and only if the current response waveform characteristics meet the instantaneous pulse mode requirements or the continuous hold mode requirements. When the current response waveform characteristics meet the target mode identifier requirements, the execution result is determined to be a successful match; When the current response waveform characteristics do not meet the target mode identifier requirements, the dual-mode switching instruction is regenerated and the drive flow for executing the target mode identifier is returned.

[0015] The beneficial effects of this invention are: 1. A dual-mode dynamic switching mechanism for magnetic latching relays was constructed. Through the coordinated control of remote command parsing and closed-loop feedback of core magnetic state, the dependence on physical jumpers was eliminated, and the static hardware configuration was transformed into dynamic control of electrical parameters. Based on the target mode identifier, a customized drive sequence was generated in real time. Combined with the online verification mechanism of core residual magnetic strength, seamless switching between continuous holding mode and instantaneous pulse mode was achieved. This broke through the physical constraints of the fixed function of traditional relays, enabling the distribution network fault indicator to have remote reconfigurability. Maintenance personnel can respond to system strategy changes without opening the cabinet on site, significantly reducing the risk of live operation and improving the flexibility and response speed of distribution network automation terminals.

[0016] 2. A multi-dimensional collaborative compensation system is introduced. Through a dual-parameter dynamic adjustment mechanism of lag time difference and eddy current loss characterization, the influence of residual core magnetism interference and eddy current effect on mode switching is avoided. Unlike the open-loop drive method of existing technologies, the magnetic flux jump characteristics are captured in real time during the execution of the drive voltage sequence. Combined with a temperature-sensing eddy current loss quantification model, the drive parameters are adaptively optimized to achieve precise matching between relay output characteristics and target mode. This solves the problem of malfunction caused by residual magnetic history state from the root, extends device life and ensures the determinism of fault signal output, and provides a highly reliable dual-mode output paradigm for distribution network protection equipment. Attached Figure Description

[0017] Figure 1 This is a flowchart of a method for dual-mode output of fault signals for a magnetic latching relay according to the present invention; Figure 2 This is a flowchart of the logic for adjusting the driving voltage sequence parameters of the present invention. Detailed Implementation

[0018] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0019] Example: Figure 1 This invention provides a dual-mode output method for fault signals of a magnetic latching relay, comprising: S1. Receive a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detect the current mechanical state of the magnetic latching relay; S2. Apply a drive voltage sequence corresponding to the target mode identifier to the control coil according to the current mechanical state; S3. Based on the dynamic margin judgment result of the iron core residual magnetism intensity and the nominal value corresponding to the target mode identifier, generate a drive enable signal or a demagnetization trigger signal. S4. In response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by the product of the current derivative and the temperature rise; in response to the demagnetization trigger signal, the drive enable signal is regenerated after automatic demagnetization is performed. S5. If the lag time difference exceeds the first threshold range of the target mode or the eddy current loss characterization exceeds the second threshold range, then adjust the rising slope and duty cycle of the driving voltage sequence. S6. Verify whether the execution result matches the target mode identifier by measuring the current response waveform generated by the adjusted drive voltage sequence in the control coil.

[0020] S1. Receive a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detect the current mechanical state of the magnetic latching relay. Specific implementation includes: The remote control terminal sends a dual-mode switching command to the magnetic latching relay via power line carrier communication. The specific process includes: the power line carrier communication module uses a sinusoidal carrier signal with a frequency of, for example, 132 kHz, to load the binary data stream of the command onto the carrier via amplitude shift keying modulation; the carrier signal is injected into the distribution network line through a coupling circuit containing a broadband transformer and a safety capacitor; the power line carrier receiving chip on the relay side separates the carrier signal through a bandpass filter and demodulates to restore the data stream; the 3rd and 4th bits in the data stream are extracted as the target mode identifier, where the bit combination "00" represents continuous hold mode and "01" represents instantaneous pulse mode; during parsing, cyclic redundancy check is performed, and if the check fails, a retransmission of the command is requested.

[0021] The position sensor uses a bipolar Hall effect sensor, installed directly above the armature shaft, for example, 1.5 mm away. The sensor is powered by 5V DC, and its output is connected to the non-inverting input of a voltage comparator. When the armature is in the reset position, the south pole of the magnet is less than 0.8 mm away from the sensor, and the sensor outputs a low level of, for example, 0.2V. When the armature is in the hold position, the north pole of the magnet is less than 0.8 mm away from the sensor, and the sensor outputs a high level of, for example, 4.8V. The voltage comparator compares the sensor output voltage with a 2.5V reference voltage and outputs a digital signal to the status determination module.

[0022] The state determination module receives the comparator output signal: if it is low, it determines that the armature is in the reset position and generates a reset state flag; if it is high, it determines that the armature is in the action hold position and generates an action hold state flag; the reset position is defined as the position where the air gap between the armature and the iron core is greater than 1.2 mm, and the action hold position is defined as the position where the air gap is less than 0.1 mm; the determination result is stored in the form of an encoding, for example, "01" represents the reset state and "10" represents the action hold state.

[0023] Temperature drift compensation of the Hall sensor is achieved through a negative temperature coefficient thermistor: the thermistor is co-located with the sensor, and its resistance change is converted into a compensation voltage by an operational amplifier and superimposed on the sensor power supply circuit; the compensation coefficient is, for example, -1.2 millivolts per degree Celsius, ensuring that the position detection error is less than ±0.05 mm in the range of -40 degrees Celsius to 85 degrees Celsius; the position determination is set with an anti-jitter timer, and the status is updated only when the position change lasts for more than 15 milliseconds.

[0024] The data frame of power line carrier communication includes a preamble, address field, instruction field, and check field; the address field is matched with the unique address of the relay; the coupling circuit uses a broadband transformer and a safety capacitor in series; the signal amplification at the receiving end uses a three-stage operational amplifier cascaded, with a total gain of, for example, 60 dB.

[0025] The calibration method for armature position detection includes: during assembly, measuring the reset position distance using a displacement sensor and adjusting the sensor mounting bracket to make the distance, for example, 1.5 mm; during operation and position calibration, inserting a standard gauge block between the armature and the iron core, and adjusting the magnet orientation to achieve an output of 4.8 volts when the gauge block thickness is 0.1 mm.

[0026] The status determination module is set to diagnose faults: if the sensor output exceeds the range of 0.1 volts to 4.9 volts, or the position change frequency exceeds 10 Hz, an alarm is triggered and the standby detection mode is switched; the standby mode determines the air gap size based on the peak value of the induced voltage by applying a 1-millisecond detection pulse to the reset coil.

[0027] Validation is performed during target pattern identifier parsing: if the pattern identifier is "10" or "11", it is considered illegal; if the count of illegal commands exceeds 3, the communication interface is locked; all operations are recorded in the event log, which includes timestamp, event type and status code.

[0028] The anti-interference design of the position sensor includes: the power supply line of the sensor uses twisted pair cable, and a ceramic capacitor is connected in parallel at the output end; a permalloy shield with a thickness of, for example, 0.3 mm is set on the outside; the signal line uses double-shielded cable, and the shield is grounded.

[0029] S2. Apply a drive voltage sequence corresponding to the target mode identifier to the control coil according to the current mechanical state. Specific implementation includes: The drive control circuit receives the mechanical state determination result and target mode identifier from step S1. When the mechanical state is reset, it directly reads the drive voltage sequence parameters corresponding to the target mode identifier from the drive parameter memory. The drive parameter memory is a ferroelectric memory. Address 0x1000 stores continuous hold mode parameters, including a rise rate of 2 volts per millisecond, a duty cycle of 100%, and a duration of 500 milliseconds. Address 0x2000 stores instantaneous pulse mode parameters, including a rise rate of 10 volts per millisecond, a duty cycle of 30%, and a pulse width of 10 milliseconds. After the parameters are read, the actual drive voltage waveform is generated by a digital-to-analog converter. The digital-to-analog converter has an update rate of 100 kHz and an output voltage range of 0 to 24 volts DC. The drive voltage is applied to the two ends of the control coil through a push-pull amplifier circuit. This circuit uses a complementary power transistor pair and has a maximum output current of 5 amps.

[0030] When the mechanical state is in the motion hold state, the reset operation is initiated first: the reset control circuit closes the reverse current switch, causing the energy storage capacitor to discharge to the control coil; the energy storage capacitor has a capacity of, for example, 100 microfarads, and a pre-charge voltage of, for example, -12V DC, forming a discharge circuit through the current sensing resistor; the discharge circuit is connected in series with a reverse blocking diode to prevent forward current from flowing back; the discharge duration is a preset fixed value, for example, 2 milliseconds, controlled by a timer circuit; the coil current is monitored in real time during the discharge, and the discharge circuit is immediately shut off when the current reaches the reverse peak threshold, for example, -0.5 times the rated current; after the reset operation is completed, the position sensor detects the armature position in real time, and when the air gap is greater than 1.2 mm and maintained for more than 1 millisecond, a reset completion signal is generated.

[0031] The reset completion signal triggers the application process of the drive voltage sequence: the drive parameter selection logic activates the corresponding memory address according to the target mode identifier; the continuous hold mode drive voltage sequence generation process is as follows: the digital-to-analog converter receives the rising slope parameter, generates a linear rising voltage waveform through the integrator circuit, and locks the output by the voltage comparator when the voltage reaches 24 volts; the instantaneous pulse mode drive voltage sequence generation process is as follows: the digital-to-analog converter outputs a square wave pulse with a pulse width of, for example, 10 milliseconds, and the leading edge of the pulse is accelerated to a rising rate of, for example, 10 volts per microsecond through the differentiating circuit; the coil temperature is monitored in real time during the application of the drive voltage, and when the temperature exceeds, for example, 85 degrees Celsius, derating protection is activated, and the output voltage is reduced by, for example, 0.5 volts per degree Celsius.

[0032] The key parameters for setting the reverse pulse current include: the discharge capacitor capacity is calculated by multiplying the current by the time and dividing by the voltage change, where the current value is, for example, 1.5 amps, the time value is, for example, 2 milliseconds, and the voltage change is, for example, 10 volts; the reverse blocking diode is selected as a fast recovery diode with a withstand voltage of, for example, 60 volts and a current of, for example, 5 amps; in the reset completion judgment condition, the air gap threshold of 1.2 mm is determined based on the iron core residual magnetic attraction curve.

[0033] The electrical matching design of the driving voltage sequence and the coil includes: controlling the DC resistance of the coil, for example, 4 ohms, and the inductance, for example, 60 millihenries; setting the output impedance of the push-pull amplifier circuit to, for example, 0.1 ohms; the accuracy calibration method for the voltage rise slope is: using a standard current probe to measure the rate of change of the coil current, and adjusting the feedback resistor of the integrator circuit so that the error between the actual slope and the set value is less than 3%; the duty cycle control is achieved through a programmable timer, with a timer clock frequency of, for example, 10 MHz.

[0034] If a reset completion signal is not detected within 3 milliseconds after the reverse pulse current is executed, a three-level retry strategy is triggered: the first retry increases the discharge time to, for example, 3 milliseconds; the second retry increases the discharge voltage to, for example, -15 volts; and the third retry fails and an alarm fault is reported. If an open circuit in the coil is detected during the application of the drive voltage, the output is immediately shut down and a fault code is stored.

[0035] The temperature coefficient of the coil resistance is, for example, 0.00393 degrees Celsius. The compensation circuit uses a thermistor to measure the ambient temperature, and generates a compensation voltage through an operational amplifier, which is then superimposed on the driving reference voltage. The compensation amount is calculated by multiplying the resistance reference value by the temperature coefficient, the temperature difference, and the set current. The resistance reference value is the coil resistance value at 25 degrees Celsius, and the temperature difference is the current temperature minus 25 degrees Celsius.

[0036] Optical isolation is used between the drive circuit and the logic control circuit; a metal oxide varistor is used in the coil drive circuit, with a clamping voltage of, for example, 36 volts; the control line is transmitted using twisted-pair shielded cable; the circuit board layout separates the power ground from the signal ground.

[0037] Field calibration method for drive parameters: Input test commands through external debugging interface, and the relay automatically performs parameter scanning; for continuous hold mode, apply drive voltage in 0.5V steps from 20V to 28V, detect and store the minimum voltage value that makes the armature fully engage; for instantaneous pulse mode, adjust the pulse width in 1ms steps from 5ms to 15ms, detect and store the shortest pulse width that makes the armature reliably operate.

[0038] S3. Based on the dynamic margin judgment result of the remanent magnetization of the iron core and the nominal value corresponding to the target mode identifier, generate a drive enable signal or a demagnetization trigger signal. The specific implementation includes: The Hall sensor is installed in the air gap of the iron core, with its sensitive axis aligned with the magnetic circuit axis of the iron core, maintaining a distance of, for example, 1.0 mm. The sensor is powered by 5 volts DC, and its output is connected to the input of an instrumentation amplifier. Before measurement, a demagnetization pretreatment is performed: an alternating magnetic field with decreasing amplitude is applied until the residual magnetic flux density is below, for example, 0.01 Tesla. After demagnetization, the sensor is allowed to stand for, for example, 10 milliseconds, and the output voltage value of the Hall sensor is read. This value is then converted into a remanent magnetization value using a calibration curve. The calibration curve is established by using a standard magnetic field generator to record the sensor output voltage in steps of 0.1 Tesla within the range of -1.0 to +1.0 Tesla, forming a linear relationship: the remanent magnetization is equal to the sensor voltage multiplied by a coefficient K plus an offset C, where K is, for example, 0.02 Tesla per millivolt, and C is, for example, -0.05 Tesla.

[0039] The pre-stored nominal value table is stored in the EEPROM memory. Example address allocation: continuous hold mode nominal values ​​are stored at addresses 0x3000 to 0x3003, and instantaneous pulse mode nominal values ​​are stored at addresses 0x3100 to 0x3103. During reading, the address is selected based on the target mode identifier: when the identifier is "00", the continuous mode address is accessed; when the identifier is "01", the pulse mode address is accessed. Data is stored in 32-bit floating-point format, for example, a continuous mode nominal value of 0.95 Tesla and a pulse mode nominal value of 0.45 Tesla. The reading process is implemented via the I²C bus, with a clock frequency of, for example, 100 kHz.

[0040] The absolute deviation rate is calculated using an analog circuit: the measured value is input to terminal A of the divider, and the nominal value is input to terminal B; the calculation result is the absolute value of the measured value minus the nominal value, which is then divided by the nominal value; the circuit uses an analog multiplier configured as a reciprocal circuit to perform division, and an instrumentation amplifier to perform absolute value calculation; the calculation response time is less than, for example, 20 microseconds; the divider output is connected to the non-inverting terminal of the subtractor, and the inverting terminal of the subtractor is connected to a reference voltage of 1.0 volts.

[0041] The preset margin threshold is set by an adjustable resistor network; when the target mode is continuous hold mode, the threshold is set to, for example, 5%; when it is instantaneous pulse mode, the threshold is set to, for example, 10%; the threshold selection is achieved by switching the resistor voltage division ratio through an analog switch, and the control signal comes from the target mode identifier register; the threshold voltage is input to the inverting input of the comparator, and the absolute deviation rate voltage is input to the non-inverting input.

[0042] Comparator output drive signal generation: When the absolute deviation rate voltage is less than or equal to the threshold voltage, the comparator outputs a high-level trigger drive enable signal; when the absolute deviation rate voltage is greater than the threshold voltage, the comparator outputs a low-level trigger demagnetization trigger signal; the signal generation uses a D flip-flop latch, and the clock is driven by, for example, a 10 kHz square wave; the drive enable signal is a continuous high-level pulse with the same width as the period of the drive voltage sequence; the demagnetization trigger signal is a single negative pulse, for example, 50 milliseconds.

[0043] The pre-stored nominal value table is generated before leaving the factory: The calibration experiment uses a magnetic flux testing platform, and the magnetic flux required for the target mode is applied to the iron core; the magnetic flux for the continuous hold mode is set to, for example, 1.2 Tesla, the excitation current is adjusted to saturate the iron core and maintained for, for example, 200 milliseconds, and then the current is removed, and the remanence intensity is measured immediately; the magnetic flux for the instantaneous pulse mode is set to, for example, 0.8 Tesla, and the remanence intensity is measured after applying a single pulse current for, for example, 10 milliseconds; each mode is measured repeatedly for, for example, 100 times, and the arithmetic mean is taken after discarding the maximum and minimum values ​​as the nominal value.

[0044] The nominal value is written to the EEPROM process as follows: the write command and data are sent to an external calibrator through the programming interface; the memory page erase time is, for example, 5 milliseconds, and the byte write time is, for example, 10 milliseconds; the write is immediately read back for verification, and the error is rewritten if it exceeds, for example, 0.5%.

[0045] When the Hall sensor output exceeds the range, switch to the backup measurement mode—calculate the residual magnetic intensity by detecting the electromotive force induced by the detection coil; when a verification error occurs when reading the nominal value, use the default value, such as 0.8 Tesla; the calculation circuit sets the output limit to ensure that the deviation rate does not exceed 100%; perform a self-test before each measurement: inject, for example, 1 mA current into the test coil to verify the sensor response.

[0046] The Hall sensor has a sensitivity temperature drift coefficient of, for example, -0.03% per degree Celsius. The compensation circuit uses a thermistor to form a bridge circuit, and the output compensation voltage is superimposed on the amplifier reference terminal. The thermistor is installed close to the sensor. The compensation formula is: the compensation voltage equals the temperature drift coefficient multiplied by the temperature difference multiplied by the reference voltage, where the temperature difference is the current temperature minus 25 degrees Celsius.

[0047] Measurement timing control: After demagnetization, a delay of 5 milliseconds is started; the Hall signal sampling and holding time is 1 millisecond; the entire measurement cycle is fixed at 10 milliseconds; the timing is generated by the crystal oscillator frequency divider circuit, and the main clock frequency is 16 MHz.

[0048] S4. In response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by integrating the product of the current derivative and the temperature rise. In response to the demagnetization trigger signal, the drive enable signal is regenerated after automatic demagnetization. The specific implementation includes: When the drive enable signal generated in step S3 is received, the drive voltage sequence application process is initiated; at the beginning of the rising edge of the drive voltage, the voltage comparator detects that the voltage exceeds, for example, a 0.5-volt threshold and triggers the magnetic flux change rate monitoring window; the duration of this window is set to, for example, 20% of the period of the drive voltage sequence, and the specific duration is determined by the period of the drive sequence; the window trigger signal starts a high-speed sampling clock with a clock frequency of, for example, 10 MHz; the window closing condition is the arrival of the drive sequence end signal or the expiration of the window duration.

[0049] The Hall sensor continuously collects magnetic flux change rate data at a sampling rate of, for example, 100,000 samples per second; the sensor output signal is amplified by an instrumentation amplifier and then input to a differentiating circuit, with the differentiating time constant set to, for example, 10 microseconds; when the differential output voltage exceeds a preset jump threshold, for example, 0.5 Tesla per millisecond, the comparator flips and marks the current moment as the start time of the magnetic flux jump; the jump threshold is set according to the characteristics of the iron core material and is adjustable by an adjustable resistor within a range of, for example, 0.1 to 1.0 Tesla per millisecond; the marking signal is held by a latch until the window ends.

[0050] Time difference measurement is achieved through a high-precision timer: the timer starts counting at the rising edge of the drive voltage and stops at the starting point of the magnetic flux jump; the timer reference clock frequency is, for example, 100 MHz, and the resolution is, for example, 10 nanoseconds; the timing result is stored in a register and output as the lag time difference; measurement error compensation is achieved using a calibration table: the timing deviation at different temperatures is measured in advance, the compensation value is stored in memory, and the application is queried in real time.

[0051] At the start of the application of the driving voltage sequence, the current differentiation circuit and the temperature acquisition circuit are started synchronously. The current differentiation is achieved through a sampling resistor and a differentiating circuit: a sampling resistor of, for example, 0.01 ohms is connected in series in the control coil circuit, and its voltage drop input is a differentiator composed of an operational amplifier and, for example, a 100 nanofarad capacitor; the differentiation time constant is set to, for example, 50 microseconds, and the output signal voltage range is, for example, from -5 volts to +5 volts; a limiting diode is set at the output terminal to prevent overvoltage.

[0052] The temperature acquisition circuit uses a negative temperature coefficient thermistor mounted on the surface of the coil frame; the thermistor and a precision resistor form a voltage divider circuit, and the voltage divider value is read by an analog-to-digital converter; the temperature sampling rate is, for example, 1 kHz, and the resolution is, for example, 0.1 degrees Celsius; the analog-to-digital conversion result is output as an analog voltage through a digital-to-analog converter, with a scaling factor of, for example, 10 millivolts per degree Celsius; the temperature probe uses a spring-loaded connection to ensure good contact.

[0053] The analog multiplier receives a differential current signal and a temperature signal: the differential current signal is input to the X input terminal of the multiplier, and the temperature signal is input to the Y input terminal; the multiplier gain coefficient is set to, for example, 0.1; the output signal is the product of the two input voltages multiplied by the gain coefficient, and the voltage range is, for example, from -10V to +10V; a current-limiting resistor is connected in series at the output terminal to prevent short circuit.

[0054] The multiplier output signal is connected to the integration circuit: the inverting input of the operational amplifier receives the signal through a resistor, for example, 1 megohm; an integrating capacitor, for example, 10 microfarads, is connected in parallel between the inverting input and the output; the non-inverting input is grounded; the integration time window is controlled by the drive voltage sequence enable signal: when the drive starts, the analog switch closes to start integration, and when the drive ends, the switch opens to stop integration; after integration, the voltage across the capacitor is read through the analog-to-digital converter, and this value is used as a measure of eddy current loss; after reading, the capacitor voltage is reset to zero through the discharge circuit.

[0055] Upon receiving the demagnetization trigger signal generated in step S3, the automatic demagnetization procedure is initiated: the reverse current switch is closed to connect the energy storage capacitor to the control coil; the pre-charge voltage of the energy storage capacitor is, for example, -12 volts, and the capacitance is, for example, 100 microfarads; a fast recovery diode is connected in series in the discharge circuit to prevent current backflow; the discharge duration is fixed by a timer to, for example, 2 milliseconds; after the discharge ends, there is a delay of, for example, 1 millisecond, during which the drive enable signal is regenerated by the signal generation circuit. The preset duration refers to the discharge duration; the delay after the discharge ends is the inherent response delay of the drive enable signal generation circuit and is not related to the preset duration.

[0056] Reverse pulse current parameter calibration method: Measure the actual current waveform using a standard current probe, and adjust the energy storage capacitor voltage so that the negative peak current reaches, for example, 1.2 times the rated value; verify the demagnetization effect by using a Hall sensor to detect the residual magnetic intensity after demagnetization, which should be less than, for example, 0.05 Tesla; if the standard is not met, increase the discharge voltage in steps, up to, for example, -18 volts.

[0057] Physical calibration of eddy current loss characterization parameters: Apply a standard driving sequence under constant temperature conditions, measure the coil temperature rise with a thermal imager, and establish the correspondence between the integrated voltage and the temperature rise value; the calibration data is stored in the memory for threshold setting in step S5; the calibration process is repeated at multiple temperature points to form a temperature compensation curve.

[0058] When the magnetic flux jump start point exceeds the monitoring window, the window is extended to a complete drive cycle; when the integral output voltage exceeds the range, automatic range switching is initiated, and a parallel shunt capacitor is used to reduce the gain; a three-level retry strategy is adopted for demagnetization failure handling, with the discharge time increased by, for example, 1 millisecond each time; all abnormal events are logged.

[0059] The analog signal channel uses shielded twisted-pair cable for transmission; a grounding protection ring is installed around the integrating amplifier; a filter capacitor is connected in parallel at the power input; a shielding layer is laid on the printed circuit board; and critical signal lines are grounded.

[0060] The opening and closing logic of the monitoring window is implemented by digital circuits: the window opening signal triggers a monostable multivibrator, and the output pulse width is equal to the preset window time; the window closing signal drives the timer to stop; the window time register can be configured via serial port, and the default value is 20% of the drive cycle.

[0061] Differential circuit calibration method: Input a standard ramp signal, measure the output slope, and adjust the feedback resistor so that the error between the actual differential coefficient and the design value is less than, for example, 3%; the calibration process is completed during the factory testing phase.

[0062] Linear compensation for temperature acquisition: Multiple calibration points are set within the range of 0 to 100 degrees Celsius, the actual partial pressure value is measured, the compensation coefficient is calculated and written into memory; nonlinear error is corrected by looking up a table during operation.

[0063] Immediately after demagnetization, the residual magnetic intensity is measured. If it is still greater than the threshold, the discharge voltage is automatically increased. After three consecutive failures, a fault code is reported and the operation is locked.

[0064] Step S3 dynamically generates drive or demagnetization signals by real-time detection of the matching degree between the remanent magnetization of the iron core and the target mode, solving the mode switching failure problem caused by residual magnetism in traditional electromagnetic systems. It introduces a magnetic state closed-loop verification mechanism to eliminate the influence of historical magnetization states on current operation. Step S4 dynamically compensates the drive parameters by capturing the voltage-magnetic flux lag time difference and eddy current loss characteristics, overcoming the response delay and abnormal temperature rise caused by the eddy current effect in the iron core. The two steps work together to form multi-physics collaborative control, significantly improving the reliability of dual-mode switching, reducing the false alarm rate, extending device life, and realizing the transformation from open-loop drive to state feedback control compared to existing static drive schemes.

[0065] Figure 2 A flowchart of the driving voltage sequence parameter adjustment logic of the present invention is given. S5: If the lag time difference exceeds the first threshold range of the target mode or the eddy current loss characterization value exceeds the second threshold range, the rising slope and duty cycle of the driving voltage sequence are adjusted. Specific implementation includes: The regulating circuit receives the hysteresis time difference and eddy current loss characterization data from step S4, and simultaneously acquires the target mode identifier. The target mode identifier is used to select the corresponding threshold parameters: when the target mode identifier is continuous hold mode, the upper and lower limits of the first threshold range and the upper and lower limits of the second threshold range are read from the parameter memory; when the target mode identifier is instantaneous pulse mode, the corresponding upper and lower limits of the first threshold range and the upper and lower limits of the second threshold range are read. The threshold parameters are set by conducting multiple sets of experiments under standard operating conditions, measuring the distribution of hysteresis time difference and eddy current loss characterization data during normal operation, and determining the threshold range based on the data distribution characteristics. For example, by statistically analyzing the normal data range, the 5th percentile of the probability distribution is taken as the lower limit and the 95th percentile as the upper limit to ensure that the threshold range covers the vast majority of normal operating conditions.

[0066] The time lag comparison circuit consists of a window comparator: the input signal is connected to the time lag difference value, the high end of the reference voltage is connected to the upper limit of the first threshold, and the low end of the reference voltage is connected to the lower limit of the first threshold; when the input signal is greater than the high end of the reference voltage, a high-level output is triggered to increase the rising slope signal; when the input signal is less than the low end of the reference voltage, a low-level output is triggered to decrease the rising slope signal; the response time of the window comparator is less than, for example, 10 microseconds; the reference voltage value is set by a digital potentiometer, and the potentiometer resistance value is automatically configured by the microcontroller according to the target mode identifier.

[0067] The rising slope adjustment is achieved by changing the integral time constant of the drive circuit: In the drive voltage generation circuit, the integral resistor network contains multiple parallel branches, each consisting of a precision resistor and a field-effect transistor switch; when it is necessary to increase the rising slope, the resistor branch with a smaller resistance value is turned on, for example, switching from 10 kΩ to 5 kΩ; when it is necessary to decrease the rising slope, the resistor branch with a larger resistance value is turned on, for example, switching from 10 kΩ to 20 kΩ; the switching action is completed within the adjustment window period after the end of the drive sequence, and the switching time is less than, for example, 1 millisecond; the resistor value is selected based on the calculation of the coil electrical parameters to ensure that the voltage change rate matches the target slope.

[0068] The comparison of eddy current loss characteristics is achieved through a dual-limit comparator: the input signal is connected to the eddy current loss characteristic value, the high end of the reference voltage is connected to the upper limit of the second threshold, and the low end of the reference voltage is connected to the lower limit of the second threshold; when the input signal is greater than the high end of the reference voltage, a high-level output is triggered to decrease the duty cycle; when the input signal is less than the low end of the reference voltage, a low-level output is triggered to increase the duty cycle; the comparison result is held by a latch until the end of the current drive cycle; the reference voltage source uses a low-temperature drift reference chip with a temperature coefficient of less than, for example, 10 ppm per degree Celsius.

[0069] Duty cycle adjustment is achieved by modifying the pulse width modulation parameters: In the drive voltage sequence generation circuit, a programmable timer generates a reference clock, and a counter sets the on-time register; when it is necessary to reduce the duty cycle, the setting value of the on-time register is decreased, for example, from 500 microseconds to 450 microseconds; when it is necessary to increase the duty cycle, the setting value of the on-time register is increased, for example, from 500 microseconds to 550 microseconds; the duty cycle adjustment step size is fixed at, for example, 1%, and the adjustment range is limited to a safe range based on the coil's thermal capacity; after each adjustment, the duty cycle value is recalculated and verified to ensure it does not exceed the safe boundary.

[0070] After each drive sequence ends, the comparator output signal triggers the parameter update logic; the update magnitude is dynamically calculated based on the degree of deviation: the relative percentage deviation between the current measured value and the threshold boundary is calculated, and the adjustment magnitude is determined according to the deviation ratio; for example, a deviation of 5% corresponds to a parameter adjustment of 1%, and a deviation of 10% corresponds to an adjustment of 2%; the adjusted parameters are temporarily stored in the shadow register and take effect when the next drive sequence begins; the adjustment process records the original parameter value, adjustment amount, and adjustment reason.

[0071] During the first drive sequence after the new parameters take effect, the hysteresis time difference and eddy current loss characteristics are re-acquired; if the measured values ​​still exceed the threshold range, a second adjustment is performed at, for example, 1.5 times the current adjustment range; if the threshold range is still not reached after three consecutive adjustments, a system-level fault diagnosis is triggered, the parameters are frozen and the abnormal event is reported; the fault event record includes timestamps, measurement value sequences and adjustment history.

[0072] The upper limit of the ramp rate is determined based on the coil insulation class, for example, no more than 30 volts per microsecond; the lower limit is set based on the minimum operating voltage, for example, no less than 0.5 volts per microsecond; the upper limit of the duty cycle is limited based on the continuous temperature rise of the coil, for example, no more than 90%; the lower limit is based on the minimum magnetization energy requirement of the iron core, for example, no less than 10%; the boundary parameters are stored in a read-only memory, and the boundary values ​​are compared in real time during the adjustment process.

[0073] An ambient temperature sensor monitors temperature changes in real time and corrects threshold parameters based on the temperature value. The compensation rule is: for every 1 degree Celsius increase in temperature, the upper limit of the first threshold increases by, for example, 0.1%, and the upper limit of the second threshold decreases by, for example, 0.05%. The compensation coefficient is obtained through high and low temperature calibration experiments: the system performance is tested in the temperature range of -40℃ to 85℃, and a fitting curve of the threshold changing with temperature is established.

[0074] During the adjustment process, the drive voltage waveform and coil current waveform are digitized by a high-speed analog-to-digital converter; the waveform data is stored in a buffer and can be output through a diagnostic interface; the monitoring system detects abnormal waveform characteristics, such as oscillating waveforms or flat-top waveforms, and automatically triggers the protection mechanism; all operation events are recorded in non-volatile memory, and the storage format includes timestamps, event types, and snapshots of key parameters.

[0075] The comparator input is equipped with a second-order low-pass filter, and the cutoff frequency is set to, for example, 1 kHz according to the signal characteristics; the reference voltage source adopts a dual voltage regulation circuit, with a Zener diode for primary voltage regulation and an operational amplifier buffer for secondary voltage regulation; the signal transmission line adopts a twisted pair shielded structure, and the shielding layer is grounded at a single point at the drive circuit end; a local grounding plane is set in the sensitive area of ​​the circuit board.

[0076] The system enters learning mode via an external calibration command; it automatically performs parameter scanning to test the performance of different parameter combinations under typical operating conditions; it updates threshold parameters and adjustment rules based on the test results, such as optimizing the adjustment step size or adjusting the threshold boundary; and the calibration data is encrypted and written to a protected storage area.

[0077] When parameter oscillation is detected (e.g., more than 10 adjustments within 1 minute), it automatically switches to conservative adjustment mode, reducing the adjustment range by 50%; when the power supply voltage fluctuates by more than ±10%, the adjustment function is suspended; all protection states are managed through a state machine to ensure that the action logic is conflict-free; fault recovery requires manual confirmation or system self-test pass.

[0078] S6. Verify whether the execution result matches the target mode identifier by measuring the current response waveform generated by the adjusted drive voltage sequence in the control coil. Specific implementation includes: The current response waveform acquisition circuit acquires the real-time current signal by controlling a sampling resistor connected in series in the control coil circuit. The resistance value of the sampling resistor is, for example, 0.005 ohms. The voltage drop across the sampling resistor is amplified by an instrumentation amplifier and then input to a high-speed analog-to-digital converter. The gain of the instrumentation amplifier is set to, for example, 100 times. The sampling rate of the analog-to-digital converter is set to, for example, 1 MHz. The converted digital signal is stored in a ring buffer with a depth of, for example, 1024 sampling points. Each sampling point contains a timestamp and current value data. The acquisition process is triggered at the start of the application of the driving voltage sequence and stops at the end of the sequence.

[0079] If the target mode is identified as continuous hold mode, the waveform analysis circuit performs steady-state current detection: First, the current response waveform is filtered by moving average, with a window width set to, for example, 100 microseconds; after filtering, the interval from 50 milliseconds after the start of the drive sequence to 50 milliseconds before the end is extracted as the analysis window; the average current value within this window is calculated as the steady-state current value; the steady-state current value is compared with the pull-in threshold comparator; the pull-in threshold is determined through calibration experiments, and the calibration method is as follows: under standard ambient temperature, the drive current is gradually increased until the armature reliably pulls in, and this critical current value is recorded as a reference, and then multiplied by a safety factor, for example, 1.1, as the pull-in threshold; it is detected whether the steady-state current remains above the pull-in threshold for the entire preset duration; the preset duration is set to, for example, 90% of the total duration of the drive sequence; if the condition is met, a matching success signal is output.

[0080] The current response waveform is deemed to meet the requirements of the continuous hold mode if and only if the current response waveform is detected to maintain a steady current above the pull-in threshold for a preset duration.

[0081] If the target mode is identified as instantaneous pulse mode, the waveform analysis circuit performs overshoot spike detection: The current response waveform is differentiated, with the differentiation time constant set to, for example, 1 microsecond; the moment when the differential output exceeds a preset rate of change threshold is detected, with the rate of change threshold set to, for example, 10 amperes per microsecond; the first point exceeding the rate of change threshold is recorded as the overshoot spike start point; the search continues until the differential value falls below the rate of change threshold, which is recorded as the spike end point; the spike pulse width and peak current are calculated; the spike pulse width is detected as being less than a preset pulse width threshold, with the preset pulse width threshold set to, for example, 50 microseconds; simultaneously, the peak current is detected as exceeding the action threshold; the action threshold is determined through a calibration experiment. The calibration method is as follows: a standard pulse drive is applied, and the pulse amplitude is gradually reduced until the action fails. The minimum peak current of a reliable action is recorded as a reference, and then multiplied by a safety factor, for example, 1.5, as the action threshold; a matching success signal is output when the conditions are met.

[0082] The current response waveform is deemed to meet the instantaneous pulse mode requirements if and only if a single overshoot spike is detected in the current response waveform within a preset pulse width and the spike amplitude exceeds the action threshold.

[0083] The matching judgment logic is implemented by a state machine: when the target mode is identified as continuous hold mode, the steady-state current detection path is activated; when the target mode is identified as instantaneous pulse mode, the overshoot spike detection path is activated; the detection result is input to the judgment circuit; a high level output indicates successful matching; the successful matching signal triggers the status indicator light to change and records the verification log.

[0084] The current response waveform characteristics are deemed to meet the target mode identification requirements if and only if the current response waveform characteristics meet the instantaneous pulse mode requirements or the continuous hold mode requirements.

[0085] When the current response waveform characteristics do not meet the target mode identifier requirements, the fault handling circuit is activated: first, a fault record is generated; the fault record is stored in non-volatile memory; then, the dual-mode switching command is regenerated; the regeneration process includes a command counter; the counter is incremented after each command is regenerated; the regenerated command is sent through the power line carrier communication module; after sending, the drive flow for executing the target mode identifier is returned; if the number of consecutive regenerations exceeds, for example, 3 times, a system-level fault lockout is triggered.

[0086] Dynamic adjustment method for pull-in threshold and actuation threshold: threshold calibration is performed periodically, with the calibration cycle set to, for example, 1000 operations; the calibration process includes applying a standard test sequence, measuring the actual operating current, and updating the threshold parameters; the threshold is stored in ferroelectric memory; the threshold adjustment range is limited to ±20% of the rated value; the temperature compensation circuit corrects the threshold in real time: for every 1 degree Celsius increase in temperature, the pull-in threshold decreases by, for example, 0.1%, and the actuation threshold decreases by, for example, 0.15%.

[0087] Quality control of waveform analysis: Self-calibration is performed before each acquisition: a standard test current, such as 1 ampere DC, is injected into the sampling circuit to verify the measurement accuracy; signal quality is monitored in real time during the acquisition process, and re-acquisition is triggered when the signal-to-noise ratio is lower than, for example, 40 dB; and a rationality check is performed before the analysis results are output.

[0088] Optimization strategy for regenerating instructions: Adjust the regeneration strategy according to the fault type: If it is a steady-state current deficiency fault, increase the driving voltage amplitude in the new instruction; if it is a peak amplitude deficiency fault, increase the driving voltage rise slope; the adjustment range is based on historical data analysis.

[0089] Verification result feedback mechanism: Successful matching signal triggers drive parameter optimization: calculates the deviation between actual waveform parameters and ideal values, and fine-tunes drive parameters; Failed matching signal triggers fault diagnosis: analyzes abnormal characteristics of current waveform and identifies potential fault modes.

[0090] Current waveform data is output to the diagnostic port via a serial peripheral interface; the monitoring system can set trigger conditions; the interface data transmission rate is set to, for example, 10 megabits per second. The sampling loop uses a four-wire connection to eliminate the influence of wire resistance; double-shielded cables are used for signal transmission; the analog-to-digital converter reference voltage source uses a low-temperature drift device. The waveform acquisition start signal and the drive sequence enable signal are strictly synchronized with a delay of less than 100 nanoseconds; the timestamp counter is obtained by dividing the drive system's main clock; key time parameters are implemented using a hardware timer.

[0091] All calculations involved in the embodiments are dimensionless numerical calculations, and the preset parameters and thresholds in the calculations are set by those skilled in the art according to the actual situation.

[0092] It should be noted that this invention can be deployed on the device itself to realize embedded applications, or it can run on a PC or other terminal with a user interface, thereby meeting various hardware environments and usage requirements.

[0093] The above embodiments can be implemented, in whole or in part, by software, hardware, firmware, or any other combination thereof. When implemented using software, the above embodiments can be implemented, in whole or in part, as a computer program product. The computer program product includes one or more computer instructions or computer programs. When the computer instructions or computer programs are loaded or executed on a computer, all or part of the processes or functions described in the embodiments of this application are generated. The computer can be a general-purpose computer, a special-purpose computer, a computer network, or other programmable device. The computer instructions can be stored in a computer-readable storage medium or transmitted from one computer-readable storage medium to another. For example, the computer instructions can be transmitted from one website, computer, server, or data center to another website, computer, server, or data center via wireless or wired transmission; wired transmission methods include optical fiber, twisted pair, coaxial cable, etc.; wireless transmission includes infrared, microwave, etc. The computer-readable storage medium can be any available medium that a computer can access or a data storage device such as a server or data center containing one or more sets of available media. The available medium can be a magnetic medium (e.g., floppy disk, hard disk, magnetic tape), an optical medium (e.g., DVD), or a semiconductor medium. A semiconductor medium can be a solid-state drive.

[0094] Those skilled in the art will understand that, for the sake of convenience and brevity, the specific working processes of the systems, devices, and modules described above can be referred to the corresponding processes in the foregoing method embodiments, and will not be repeated here.

[0095] In the several embodiments provided in this application, it should be understood that the disclosed systems, apparatuses, and methods can be implemented in other ways. For example, the apparatus embodiments described above are merely illustrative; for instance, the division of modules is only a logical functional division, and in actual implementation, there may be other division methods. For example, multiple modules or components may be combined or integrated into another system, or some features may be ignored or not executed. Furthermore, the coupling or direct coupling or communication connection shown or discussed may be through some interfaces; the indirect coupling or communication connection between apparatuses or modules may be electrical, mechanical, or other forms.

[0096] The modules described as separate components may or may not be physically separate. The components shown as modules may or may not be physical modules; they may be located in one place or distributed across multiple network modules. Some or all of the modules can be selected to achieve the purpose of this embodiment according to actual needs.

[0097] In addition, the functional modules in the various embodiments of this application can be integrated into one processing module, or each module can exist physically separately, or two or more modules can be integrated into one module.

[0098] If the aforementioned functions are implemented as software functional modules and sold or used as independent products, they can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of this application, in essence, or the part that contributes to the prior art, or a portion of the technical solution, can be embodied in the form of a software product. This computer software product is stored in a storage medium and includes several instructions to cause a computer device (which may be a personal computer, server, or network device, etc.) to execute all or part of the steps of the methods described in the various embodiments of this application. The aforementioned storage medium includes various media capable of storing program code, such as USB flash drives, portable hard drives, read-only memory (ROM), random access memory (RAM), magnetic disks, or optical disks.

[0099] The above description is merely a specific embodiment of this application, but the scope of protection of this application is not limited thereto. Any variations or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in this application should be included within the scope of protection of this application. Therefore, the scope of protection of this application should be determined by the scope of the claims.

[0100] In conclusion, the above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A method for dual-mode output of fault signals from a magnetic latching relay, characterized in that, include: S1. Receive a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detect the current mechanical state of the magnetic latching relay; S2. Apply a drive voltage sequence corresponding to the target mode identifier to the control coil according to the current mechanical state; S3. Based on the dynamic margin judgment result of the iron core residual magnetism intensity and the nominal value corresponding to the target mode identifier, generate a drive enable signal or a demagnetization trigger signal. S4. In response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by the product of the current derivative and the temperature rise; in response to the demagnetization trigger signal, the drive enable signal is regenerated after automatic demagnetization is performed. S5. If the lag time difference exceeds the first threshold range of the target mode or the eddy current loss characterization exceeds the second threshold range, then adjust the rising slope and duty cycle of the driving voltage sequence, including: When the hysteresis time difference is greater than the upper limit of the first threshold of the target mode, increase the rising slope of the driving voltage sequence; When the hysteresis time difference is less than the lower limit of the first threshold of the target mode, reduce the rising slope of the driving voltage sequence; When the eddy current loss characterization is greater than the upper limit of the second threshold of the target mode, reduce the duty cycle of the driving voltage sequence; When the eddy current loss characterization is less than the second threshold lower limit of the target mode, increase the duty cycle of the driving voltage sequence; S6. Verify whether the execution result matches the target mode identifier by measuring the current response waveform generated by the adjusted drive voltage sequence in the control coil.

2. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 1, characterized in that, Receive a dual-mode switching command containing a target mode identifier sent by a remote control terminal, and detect the current mechanical state of the magnetic latching relay, including: Receive dual-mode switching commands transmitted by remote control terminals via power line carrier communication, and parse the target mode identifier in the dual-mode switching commands; The real-time position of the armature of the magnetic latching relay is detected by a position sensor; If the armature is in the reset position, the current mechanical state is determined to be the reset state; If the armature is in the motion hold position, the current mechanical state is determined to be the motion hold state.

3. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 1, characterized in that, Apply a sequence of drive voltages corresponding to the target mode identifier to the control coil based on the current mechanical state, including: If the current mechanical state is reset, then the drive voltage sequence corresponding to the target mode identifier is directly applied to the control coil; If the current mechanical state is in the motion holding state, a reverse pulse current is first applied to the control coil to perform a reset operation; After the reset operation is completed, a drive voltage sequence corresponding to the target mode identifier is applied to the control coil.

4. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 1, characterized in that, Based on the dynamic margin judgment result of the remanent magnetization of the iron core and the nominal value corresponding to the target mode identifier, a drive enable signal or a demagnetization trigger signal is generated, including: The remanence of the iron core is measured using a Hall effect sensor; Read the nominal value of remanence intensity corresponding to the target mode identifier from the pre-stored nominal value table; Calculate the absolute deviation rate between the remanence intensity and the nominal value of the remanence intensity; If the absolute deviation rate is less than or equal to the preset margin threshold, a drive enable signal is generated; If the absolute deviation rate is greater than the preset margin threshold, a demagnetization trigger signal is generated.

5. The method for dual-mode output of fault signals for a magnetic latching relay according to claim 4, characterized in that, The pre-stored nominal value table is generated in the following way: Calibration experiments were conducted on the continuous hold mode and the instantaneous pulse mode respectively; In the calibration experiment, the remanence of the iron core is measured when the magnetic flux required to reach the target mode is measured. Store the remanence value corresponding to the continuous hold mode as the first nominal value; Store the remanence value corresponding to the instantaneous pulse mode as the second nominal value; A mapping table between target mode identifiers and nominal values ​​is established and written into the EEPROM memory to obtain a pre-stored nominal value table; the nominal values ​​include the first nominal value and the second nominal value.

6. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 1, characterized in that, In response to the drive enable signal, the hysteresis time difference between the rising edge of the drive voltage and the starting point of the magnetic flux jump is captured in real time when the drive voltage sequence is applied, and the eddy current loss characterization quantity is generated by integrating the product of the current derivative and the temperature rise, including: The magnetic flux change rate monitoring window is triggered at the start of the rising edge of the driving voltage. Magnetic flux change rate data are continuously collected using a Hall sensor; When the rate of change of magnetic flux exceeds the preset jump threshold, mark the current moment as the start time of the magnetic flux jump. The time difference between the start of the rising edge of the driving voltage and the start of the magnetic flux jump is calculated as the hysteresis time difference. The current differentiation circuit and temperature acquisition circuit are started synchronously during the application of the driving voltage sequence; The output signal of the current differential operation circuit and the output signal of the temperature acquisition circuit are input into the analog multiplier. The output signal of the analog multiplier is input into the integral circuit to generate the eddy current loss characterization quantity.

7. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 6, characterized in that, The output signal of the analog multiplier is input into the integrator circuit to generate an eddy current loss characteristic, which is achieved in the following way: The output voltage signal of the analog multiplier is connected to the integration circuit consisting of an operational amplifier and an integrating capacitor. Set the integration time window to cover the entire application period of the driving voltage sequence; The output voltage signal of the analog multiplier is continuously integrated over time using an integration circuit. At the end of the integration time window, the voltage across the integrating capacitor is read as a characterization of eddy current loss.

8. The method for dual-mode output of fault signals of a magnetic latching relay according to claim 1, characterized in that, In response to the demagnetization trigger signal, automatic demagnetization is performed, and then a new drive enable signal is generated, including: Apply a reverse pulse current of a preset duration to the control coil; The drive enable signal is regenerated after the reverse pulse current ends.

9. A method for dual-mode output of fault signals for a magnetic latching relay according to claim 1, characterized in that, The execution result is verified to match the target mode identifier by measuring the current response waveform generated by the adjusted drive voltage sequence in the control coil, including: The real-time current signal at both ends of the control coil is acquired to generate a current response waveform; If the target mode is identified as continuous hold mode, detect whether the current response waveform maintains a steady current higher than the pull-in threshold for a preset duration; if and only if the current response waveform is detected to maintain a steady current higher than the pull-in threshold for a preset duration, it is determined that the current response waveform characteristics meet the requirements of continuous hold mode. If the target mode is identified as instantaneous pulse mode, detect whether the current response waveform has a single overshoot spike within the preset pulse width and the spike amplitude exceeds the action threshold; if and only if the current response waveform has a single overshoot spike within the preset pulse width and the spike amplitude exceeds the action threshold, it is determined that the current response waveform characteristics meet the instantaneous pulse mode requirements. The current response waveform characteristics are deemed to meet the target mode identification requirements if and only if the current response waveform characteristics meet the instantaneous pulse mode requirements or the continuous hold mode requirements. When the current response waveform characteristics meet the target mode identifier requirements, the execution result is determined to be a successful match; When the current response waveform characteristics do not meet the target mode identifier requirements, the dual-mode switching instruction is regenerated and the drive flow for executing the target mode identifier is returned.

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