A stepper motor controller and control method for automobiles

Abstract

This invention relates to the field of motor control technology, specifically to a stepper motor controller and control method for automobiles. The method includes: real-time acquisition of bus voltage, calculation of voltage fluctuation compensation coefficient, and superposition of this coefficient into the given phase voltage command of the current inner loop; acquisition of phase current input back EMF using a sliding mode observer to estimate the actual electrical angle and estimated speed of the rotor; construction of a dual closed-loop control structure consisting of a position outer loop and a current inner loop; differential calculation of the given speed and estimated speed to obtain the speed deviation; determination of whether the speed deviation exceeds a preset threshold; if it does, triggering an adaptive step angle adjustment mechanism to dynamically reduce the microstepping value to increase the single-step electromagnetic torque; if the deviation reverts, restoring the initial microstepping value. This scheme eliminates the distortion effect of bus voltage fluctuations on the phase current waveform, maintains the rotor's synchronous tracking state under sudden load increases, and avoids step loss.

Description

Technical Field

[0001] This invention relates to the field of motor control technology, specifically to a stepper motor controller and control method for automobiles. Background Technology

[0002] Stepper motor control in automotive actuators typically employs an open-loop constant current chopper drive or a single current closed-loop control architecture. The operation of this conventional control architecture is as follows: the controller receives position commands from the host computer, generates a corresponding stepped sinusoidal current command signal based on a fixed microstepping value, obtains the actual value of the motor phase current through a current sampling circuit, performs differential calculation between the actual phase current value and the command signal, and inputs the differential result into the current regulator to generate a pulse width modulation signal, ultimately driving the full-bridge power circuit to output drive current to the motor windings. During this process, the controller relies on a fixed microstepping value to maintain the rotating magnetic field of the motor, and through the current closed loop, ensures that the actual phase current tracks the command current, thereby controlling the displacement and speed of the motor rotor.

[0003] Vehicle power supply networks experience bus voltage dips and surges under conditions such as start-stop system operation and instantaneous connection of high-power loads. The aforementioned existing technology uses a fixed microstep subdivision output drive signal. When the bus voltage drops, the fixed phase current command signal cannot output electromagnetic torque to overcome the current mechanical load under the constraint of the reduced bus voltage. The rotor lags behind the stator rotating magnetic field, causing the motor to lose synchronism when the load suddenly increases and is accompanied by power supply fluctuations. Summary of the Invention

[0004] The purpose of this invention is to provide a stepper motor controller and control method for automobiles, which can effectively solve the problems in the background art.

[0005] To achieve the above objectives, the technical solution adopted by the present invention is as follows:

[0006] A stepper motor controller and control method for automobiles include the following steps: real-time acquisition of the bus voltage of the stepper motor, calculation of the voltage fluctuation compensation coefficient, and superposition of the voltage fluctuation compensation coefficient into the given phase voltage command of the current inner loop;

[0007] The phase current of the stepper motor is collected in real time and input to the back EMF sliding mode observer. The actual electrical angle and estimated speed of the rotor are estimated through the back EMF sliding mode observer.

[0008] A dual closed-loop control structure consisting of an outer position loop and an inner current loop is constructed, and the speed deviation is obtained by differential calculation between the estimated speed and the given speed.

[0009] Determine whether the speed deviation exceeds a preset threshold. If the speed deviation exceeds the preset threshold, trigger the adaptive step angle adjustment mechanism to dynamically reduce the current microstep subdivision to increase the single-step electromagnetic torque.

[0010] If the speed deviation returns to within the preset threshold, the initial microstep subdivision is restored, and a drive signal is generated based on the adjusted microstep subdivision to control the operation of the stepper motor.

[0011] Preferably, the step of calculating the voltage fluctuation compensation coefficient and superimposing the voltage fluctuation compensation coefficient into the given phase voltage command of the current inner loop includes: obtaining the voltage deviation value between the current bus voltage and the rated bus voltage, and calculating the ratio of the voltage deviation value to the rated bus voltage to obtain the voltage fluctuation ratio;

[0012] The voltage fluctuation ratio is input into the feedforward compensation function to calculate the voltage fluctuation compensation coefficient;

[0013] Within each pulse width modulation cycle, the duty cycle of the base phase voltage output of the inner current loop is obtained, and the duty cycle of the base phase voltage is divided by the voltage fluctuation compensation coefficient to obtain the corrected target phase voltage duty cycle. A space vector pulse width modulation signal is generated based on the target phase voltage duty cycle.

[0014] Preferably, the step of inputting the phase current to the back EMF sliding mode observer and estimating the actual electrical angle and estimated speed of the rotor through the back EMF sliding mode observer includes: performing Clark transformation and Park transformation on the phase current to obtain the actual current components in the stationary two-phase coordinate system;

[0015] The actual current component is input into the sliding mode approach law function to calculate the observed current component containing the high-frequency chattering signal;

[0016] Calculate the difference between the actual current component and the observed current component, input the difference into a low-pass filter to filter out high-frequency signals, and extract the back electromotive force observation value;

[0017] The back electromotive force observation value is input into the phase-locked loop, and the actual electrical angle of the rotor and the estimated speed are calculated by the proportional-integral regulator of the phase-locked loop.

[0018] Preferably, the construction of the dual closed-loop control structure of the position outer loop and the current inner loop includes: calculating the position deviation by differentially calculating the actual electrical angle of the rotor and the given electrical angle, and inputting the position deviation into the position proportional regulator to calculate the given speed;

[0019] The speed deviation is calculated by differentiating the given speed from the estimated speed, and the speed deviation is input to the speed proportional-integral controller to calculate the given quadrature-axis current;

[0020] The current deviation is calculated by differentially calculating the given quadrature-axis current and the actual quadrature-axis current. The current deviation is then input into the current proportional-integral regulator to calculate the given phase voltage command, thus completing the series control of the position outer loop on the current inner loop.

[0021] Preferably, determining whether the speed deviation exceeds a preset threshold includes: obtaining the current operating condition parameters of the stepper motor, wherein the operating condition parameters include the estimated speed and the current microstep subdivision.

[0022] Based on the operating condition parameters, a pre-stored dynamic threshold mapping table is queried to obtain the target dynamic threshold corresponding to the current estimated speed and the current microstep subdivision number. The target dynamic threshold in the dynamic threshold mapping table decreases as the estimated speed increases.

[0023] The absolute value of the speed deviation is compared with the target dynamic threshold. If the absolute value of the speed deviation is greater than the target dynamic threshold, it is determined that the speed deviation exceeds the preset threshold.

[0024] Preferably, the dynamic reduction of the current microstep subdivision number to increase the single-step electromagnetic torque includes: when the adaptive step angle adjustment mechanism is triggered, obtaining the first phase current given amplitude corresponding to the current microstep subdivision number at the current moment, and the second phase current given amplitude corresponding to the reduced target microstep subdivision number;

[0025] During the transition interval from the current pulse width modulation cycle to the next pulse width modulation cycle, linear ramp interpolation is performed on the given amplitude of the first phase current and the given amplitude of the second phase current to obtain the given amplitude of the transition phase current.

[0026] The given amplitude of the transition phase current is used as the input reference for the inner current loop, so that the phase current amplitude changes continuously during the microstep subdivision switching process.

[0027] Preferably, after calculating the ratio of the voltage deviation value to the rated bus voltage to obtain the voltage fluctuation ratio, the method further includes: performing a first-order derivative calculation on the voltage deviation value to obtain the bus voltage change rate.

[0028] The voltage differential compensation amount is obtained by multiplying the bus voltage change rate by a preset voltage change rate weighting coefficient.

[0029] The voltage differential compensation is superimposed on the voltage fluctuation ratio to obtain the voltage fluctuation ratio with fused differential characteristics;

[0030] The voltage fluctuation ratio of the fused differential feature is input into the feedforward compensation function to calculate the voltage fluctuation compensation coefficient of the fused differential feature, so as to correct the duty cycle of the target phase voltage in advance when the bus voltage undergoes a step change.

[0031] Preferably, before inputting the difference into a low-pass filter to filter out high-frequency signals, the method further includes: extracting the estimated rotational speed, and dynamically adjusting the observation gain coefficient in the sliding mode approach law function according to the rotational speed range in which the estimated rotational speed is located, wherein when the estimated rotational speed is in the low-speed range, the observation gain coefficient is configured to a first value, and when the estimated rotational speed is in the high-speed range, the observation gain coefficient is configured to a second value, and the first value is greater than the second value;

[0032] Meanwhile, within the high-speed range, a back EMF amplitude compensation factor is calculated based on the estimated rotational speed, and the back EMF amplitude compensation factor is superimposed on the input of the low-pass filter.

[0033] Preferably, the operating condition parameters also include the winding temperature of the stepper motor and the bus voltage;

[0034] The step of querying a pre-stored dynamic threshold mapping table based on the operating condition parameters to obtain the target dynamic threshold corresponding to the current estimated speed and the current microstep subdivision number includes: taking the winding temperature, the bus voltage, the estimated speed and the current microstep subdivision number as input vectors, performing interpolation query in the four-dimensional dynamic threshold mapping table to obtain the initial dynamic threshold;

[0035] Obtain the step loss trigger frequency of the stepper motor in the historical operating cycle, and multiply the step loss trigger frequency by the penalty coefficient to obtain the threshold offset;

[0036] The threshold offset is superimposed on the initial dynamic threshold to obtain the final target dynamic threshold.

[0037] Preferably, while performing linear ramp interpolation calculation on the given amplitude of the first phase current and the given amplitude of the second phase current, the method further includes: obtaining the first electrical angle step resolution under the current microstep subdivision number and the second electrical angle step resolution under the target microstep subdivision number;

[0038] Calculate the phase angle difference between the first electrical angle step resolution and the second electrical angle step resolution;

[0039] Within the transition interval of the linear ramp interpolation calculation, the phase angle difference is synchronously superimposed onto the actual electrical angle of the rotor according to the interpolation ratio to obtain the compensated actual electrical angle.

[0040] The compensated actual electrical angle is used as the feedback input of the outer loop of the position to eliminate the spatial electrical angle mapping fault caused by the microstep subdivision switching.

[0041] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0042] 1. This invention calculates a voltage fluctuation compensation coefficient by collecting the bus voltage and superimposes it onto the given phase voltage command in the inner current loop. This allows the phase current command signal to adaptively compensate for bus voltage drops and surges, eliminating the distortion effect of power supply fluctuations on the phase current waveform. Simultaneously, a back-EMF sliding mode observer is introduced to obtain the estimated speed. The deviation between the estimated speed and the given speed is compared with a preset threshold. When the deviation exceeds the threshold, the microstepping increment is dynamically reduced to increase the single-step electromagnetic torque. This enables the motor to generate torque to overcome mechanical loads under conditions of power supply drops and sudden load increases, maintaining rotor synchronous tracking and preventing step loss.

[0043] 2. By querying a dynamic threshold mapping table based on operating condition parameters, the target dynamic threshold is obtained, enabling the out-of-step judgment condition to adapt to the electromechanical characteristics under different speeds and microstep subdivisions. During the microstep subdivision switching process, linear ramp interpolation is used to calculate the given amplitude of the transition phase current, and the phase angle difference is superimposed onto the actual electrical angle according to the interpolation ratio, eliminating the sudden change in current amplitude and spatial electrical angle mapping discontinuity caused by the microstep subdivision switching process. In voltage feedforward compensation, the bus voltage change rate is introduced to obtain the voltage differential compensation amount, and the phase voltage command is corrected in advance when a step change occurs in the bus voltage, changing the response timing of the feedforward compensation. Attached Figure Description

[0044] Figure 1 This is the overall main flowchart of the stepper motor control method for automobiles according to the present invention;

[0045] Figure 2 This is a flowchart illustrating the calculation of voltage fluctuation compensation coefficients and the generation of space vector pulse width modulation signals according to the present invention.

[0046] Figure 3 This is a flowchart illustrating the estimation of the actual electrical angle and rotational speed of the rotor using a back EMF sliding mode observer according to the present invention.

[0047] Figure 4 This is a logic flowchart of the dual closed-loop control structure of the position outer loop and the current inner loop of the present invention;

[0048] Figure 5 This is a flowchart of the mechanism for determining speed deviation and triggering adaptive step angle adjustment based on a dynamic threshold mapping table according to the present invention.

[0049] Figure 6This is a flowchart illustrating the linear ramp interpolation and spatial electrical angle compensation during the microstep subdivision switching process of the present invention. Detailed Implementation

[0050] The technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. Based on the embodiments of this application, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of this application.

[0051] Please refer to Figure 1 This embodiment provides a stepper motor controller and control method for automobiles, which is applied to a two-phase hybrid stepper motor in automobile actuators. The automobile actuators include, but are not limited to, electronic throttle actuators, electronic gear shift actuators, window lift actuators, seat adjustment actuators, and other on-board stepper motor drive systems.

[0052] In one embodiment, the control process of the stepper motor is as follows:

[0053] Within each pulse width modulation (PWM) cycle, the bus voltage of the stepper motor is synchronously acquired. A voltage fluctuation compensation coefficient is calculated based on the acquired bus voltage, and this coefficient is then superimposed on the given phase voltage command of the inner current loop to achieve feedforward compensation for bus voltage fluctuations. Specifically, the bus voltage acquisition time is synchronized with the midpoint of the PWM cycle to avoid the influence of electromagnetic interference generated by the switching action of power devices on the sampling results. The acquired bus voltage is digitally filtered before being used to calculate the voltage fluctuation compensation coefficient. The digital filtering process employs a first-order inertial filtering algorithm, with the filtering time constant configured based on the PWM cycle to ensure a balance between the smoothness of the sampling results and the response speed. The voltage fluctuation compensation coefficient is calculated based on the ratio of the real-time acquired bus voltage to the rated bus voltage. The resulting coefficient is used to correct the duty cycle of the phase voltage output from the inner current loop, enabling the phase voltage amplitude output to the motor windings to adaptively adjust to follow the fluctuations in the bus voltage, ensuring that the tracking accuracy of the phase current is not affected by bus voltage drops or surges.

[0054] Synchronized with the acquisition of the bus voltage, the phase currents of two-phase windings of the stepper motor are acquired in each pulse width modulation cycle. The acquired phase currents are input to a back-EMF sliding mode observer, which estimates the actual electrical angle and rotational speed of the rotor. Specifically, the acquisition of the two-phase winding phase currents and the bus voltage are sampled synchronously. After undergoing the same digital filtering process as the bus voltage, the acquired phase currents are first subjected to a Clarke transform to convert the current components in the three-phase stationary coordinate system into current components in the stationary two-phase coordinate system. The third-phase current of the two-phase hybrid stepper motor can be calculated based on the constraint that the sum of the two-phase currents is zero. Part of the stationary two-phase current components after the Clarke transform are input to the input of the back-EMF sliding mode observer, while the other part, after a Park transform, is converted into direct-axis and quadrature-axis current components in a synchronous rotating coordinate system, serving as the feedback input for the inner current loop. The back EMF sliding mode observer is based on the mathematical model of the stepper motor in a stationary two-phase coordinate system. Through the sliding mode variable structure control algorithm, the back EMF component of the motor winding is observed. After the observed back EMF component is filtered, the rotor position information and speed information are extracted, and finally the actual electrical angle and estimated speed of the rotor are output.

[0055] A dual-closed-loop control structure consisting of an outer position loop and an inner current loop is constructed, configured in a cascaded manner. The output of the outer position loop serves as the given input for the speed control loop, and the output of the speed control loop serves as the given input for the inner current loop. The output of the inner current loop serves as the final phase voltage control command. Specifically, the estimated speed output from the back-EMF sliding mode observer is differentially calculated with the given speed from the speed control loop to obtain the speed deviation. This speed deviation characterizes the degree of deviation between the actual and desired operating states of the rotor, serving as the core basis for determining the risk of loss of synchronization. The input of the outer position loop is the given electrical angle converted from the given position command sent by the host computer, and the feedback input of the outer position loop is the actual electrical angle of the rotor output from the back-EMF sliding mode observer. Based on the difference between the given and actual electrical angles, the outer position loop outputs a given speed through proportional adjustment calculation, which serves as the given input for the speed control loop. Based on the speed deviation between the given and estimated speeds, the speed control loop outputs a given quadrature-axis current through proportional-integral adjustment calculation, which serves as the given input for the inner current loop. The inner current loop outputs a given quadrature-axis voltage and a given direct-axis voltage based on the difference between the given quadrature-axis current and the actual quadrature-axis current, and the difference between the given direct-axis current and the actual direct-axis current, respectively, through proportional-integral adjustment calculations. After inverse Park transformation, the given quadrature-axis voltage and the given direct-axis voltage are converted into a given phase voltage command in a stationary two-phase coordinate system. The given phase voltage command is the reference command corrected by the voltage fluctuation compensation coefficient.

[0056] The calculated speed deviation is compared with a preset threshold to determine if it exceeds the threshold. If the speed deviation exceeds the preset threshold, the motor is deemed to have a risk of losing steps, triggering an adaptive step angle adjustment mechanism to dynamically reduce the current microstepping value, thereby increasing the single-step electromagnetic torque and improving the motor's resistance to load disturbances and voltage fluctuations. If the speed deviation returns to within the preset threshold, the risk of losing steps is eliminated, and the motor is restored to the initial microstepping value, ensuring the smooth operation and position control accuracy of the motor. Based on the adjusted microstepping value and the actual electrical angle of the rotor, a corresponding stepped sinusoidal phase current reference signal is generated. This phase current reference signal serves as the input reference for the current inner loop and is ultimately used to generate a drive signal through space vector pulse width modulation. This signal controls the power drive circuit of the stepper motor to output the corresponding drive current, driving the stepper motor to run. Specifically, the adjustment of the microstepping value is performed based on a preset microstepping value mapping relationship. Each adjustment is performed by gradually decreasing or increasing the preset step size to avoid sudden changes in the motor's operating state caused by abrupt changes in the microstepping value. The single-step electromagnetic torque is negatively correlated with the microstep subdivision. The lower the microstep subdivision, the larger the single-step electrical angle, the stronger the coupling effect between the stator rotating magnetic field and the rotor permanent magnet, and the higher the maximum single-step electromagnetic torque that can be output. This can overcome the problem of insufficient torque caused by larger mechanical load disturbances and power supply fluctuations.

[0057] In this embodiment, the basic operating parameters and initial configuration of the stepper motor are shown in the table below:

[0058] Table 1 Basic Operating Parameters and Initial Configuration of Automotive Stepper Motor

[0059]

[0060] This table shows the basic operating parameters and initial configuration used in the stepper motor control process in this embodiment. All control logic calculations are performed based on these configuration parameters. The initial microstep subdivision is the default configuration for steady-state operation of the motor. The preset speed deviation threshold is the initial benchmark value for judging the risk of motor step loss. All regulator parameters are tuned based on the electromechanical characteristics of the motor under rated operating conditions to ensure the stability and response speed of the control loop.

[0061] In this embodiment, real-time acquisition and feedforward compensation of the bus voltage eliminates the impact of fluctuations in the vehicle power supply network on the phase current output, ensuring the stability of the motor's output electromagnetic torque. A back-EMF sliding mode observer enables sensorless estimation of rotor position and speed, eliminating the need for additional position sensors, reducing the hardware cost and failure rate of the vehicle-mounted actuator, and improving the system's environmental adaptability. A dual-closed-loop cascaded control structure of the position outer loop and current inner loop achieves precise control of the motor's position, speed, and current, improving the system's dynamic response performance and steady-state control accuracy. An adaptive step angle adjustment mechanism based on speed deviation dynamically increases the output torque when the motor is at risk of losing steps, avoiding step loss under conditions of sudden load increases and power supply drops, thus ensuring the operational reliability of the vehicle-mounted actuator.

[0062] In a preferred embodiment, reference Figure 2 Based on the bus voltage acquisition process of the aforementioned embodiment, the calculation of voltage fluctuation compensation coefficient and the correction process of phase voltage command are further refined.

[0063] Within each pulse width modulation cycle, the real-time acquired actual bus voltage value and the rated bus voltage of the stepper motor are obtained. The voltage deviation between the current bus voltage and the rated bus voltage is calculated, whereby the voltage deviation is the difference between the actual bus voltage value and the rated bus voltage. Based on the calculated voltage deviation, the ratio of the voltage deviation to the rated bus voltage is calculated to obtain the voltage fluctuation ratio, which characterizes the degree of deviation of the bus voltage from its rated value. The calculated voltage fluctuation ratio is input into a feedforward compensation function to calculate the voltage fluctuation compensation coefficient. The feedforward compensation function is a linear proportional function, and the output voltage fluctuation compensation coefficient has a linear relationship with the voltage fluctuation ratio, expressed as:

[0064]

[0065] In the formula, This is the voltage fluctuation compensation coefficient. This is the actual value of the bus voltage collected in real time. This is the rated bus voltage of the stepper motor.

[0066] Within each pulse width modulation (PWM) cycle, the duty cycle of the base phase voltage output from the inner current loop is acquired. This base phase voltage duty cycle is the uncompensated phase voltage duty cycle calculated by the inner current loop based on the current deviation, corresponding to the phase voltage output requirement under the rated bus voltage. The base phase voltage duty cycle is divided by the voltage fluctuation compensation coefficient to obtain the corrected target phase voltage duty cycle. The space vector pulse width modulation (SVM) signal is then generated based on this target phase voltage duty cycle. Specifically, the generation process of the SVM signal includes four stages: sector determination, vector action time calculation, overmodulation processing, and carrier comparison. The corrected target phase voltage duty cycle is directly used as the input reference for vector action time calculation, enabling the duty cycle of the final output PWM signal to adaptively adjust to follow the fluctuations in the bus voltage. When the bus voltage drops, the voltage fluctuation compensation coefficient is less than 1, and the corrected target phase voltage duty cycle is greater than the base phase voltage duty cycle, increasing the conduction time of the power drive circuit and compensating for the reduction in phase voltage amplitude caused by the voltage drop. When the bus voltage surges, the voltage fluctuation compensation coefficient is greater than 1, and the corrected target phase voltage duty cycle is less than the base phase voltage duty cycle, reducing the conduction time of the power drive circuit, suppressing the increase in phase voltage amplitude caused by the voltage surge, and ultimately ensuring the stability of the phase voltage amplitude output to the motor windings, eliminating the distortion effect of bus voltage fluctuations on the phase current waveform.

[0067] Further, after calculating the ratio of the voltage deviation value to the rated bus voltage to obtain the voltage fluctuation ratio, the first derivative of the voltage deviation value is performed to obtain the bus voltage change rate, which is used to characterize the dynamic change trend of the bus voltage. The calculated bus voltage change rate is multiplied by a preset voltage change rate weighting coefficient to obtain the voltage differential compensation amount. The voltage change rate weighting coefficient is tuned based on the pulse width modulation period and the system response bandwidth to ensure a balance between the proactive adjustment effect of differential compensation and system stability. The expression for the voltage differential compensation amount is:

[0068]

[0069] In the formula, This is the voltage differential compensation amount. The preset voltage change rate weighting coefficient, This is the first derivative of the bus voltage with respect to time, i.e., the rate of change of the bus voltage.

[0070] The calculated voltage differential compensation is superimposed on the aforementioned voltage fluctuation ratio to obtain the voltage fluctuation ratio with integrated differential characteristics. This integrated differential characteristic voltage fluctuation ratio simultaneously includes both static deviation information and dynamic trend information of the bus voltage. The integrated differential characteristic voltage fluctuation ratio is input into the aforementioned feedforward compensation function to calculate the integrated differential characteristic voltage fluctuation compensation coefficient. Based on this coefficient, the duty cycle of the base phase voltage is corrected to obtain the final target phase voltage duty cycle. When the bus voltage undergoes a step change, the absolute value of the bus voltage change rate is large, and the corresponding voltage differential compensation can reflect the voltage change trend in advance. This allows the corrected target phase voltage duty cycle to adapt to the upcoming voltage fluctuation in advance, changing the response timing of the feedforward compensation and avoiding transient distortion of phase current and torque fluctuation caused by voltage step changes.

[0071] In this embodiment, the calculation results of the compensation coefficient and the duty cycle correction effect under different bus voltage fluctuation conditions are shown in the following table:

[0072] Table 2. Correspondence between bus voltage fluctuation conditions and compensation coefficients

[0073]

[0074] This table shows the calculation results of the voltage fluctuation compensation coefficient and the duty cycle correction effect under different bus voltage fluctuation conditions. Condition 1 represents the rated voltage steady-state condition; conditions 2-4 represent static voltage dips and surges; and conditions 5-6 represent dynamic voltage step changes. Voltage change rate weighting coefficient. Configured to 0.05ms / V, when the voltage undergoes a step change, the differential compensation is superimposed on the voltage fluctuation ratio, so that the fused compensation coefficient can respond to the voltage change trend in advance. The corrected duty cycle can adapt to the upcoming voltage fluctuation in advance, avoiding phase current distortion caused by sudden voltage changes.

[0075] In this embodiment, feedforward compensation based on voltage fluctuation ratio is used to achieve accurate compensation for static fluctuations in bus voltage, ensuring the tracking accuracy of phase current under steady-state conditions. By introducing voltage differential compensation, proactive adjustment of dynamic changes in bus voltage is achieved. When a step change occurs in voltage, the phase voltage duty cycle can be corrected in advance, eliminating current distortion and torque fluctuations caused by transient voltage fluctuations, and further improving the stability and reliability of motor operation under complex on-board power supply conditions.

[0076] In another preferred embodiment, reference Figure 3 and Figure 4 Based on the phase current acquisition process of the aforementioned embodiments, the estimation process of the back EMF sliding mode observer and the implementation process of the dual closed-loop control structure are further refined.

[0077] The acquired two-phase winding phase currents are subjected to Clarke and Parker transformations to obtain the actual current components in the stationary two-phase coordinate system and the direct-axis and quadrature-axis actual current components in the synchronous rotating coordinate system. Specifically, the Clarke transformation is performed based on the mapping relationship between the two-phase stationary coordinate system and the three-phase stationary coordinate system, converting the acquired phase currents in the two-phase natural coordinate system into... The actual current component in a stationary two-phase coordinate system, the stationary two-phase coordinate system The shaft coincides with the axis of the motor's A-phase winding. Axis ahead The axis is 90 degrees electrical angle. The Parker transformation calculation is performed based on the actual electrical angle of the rotor. The actual current component in the stationary two-phase coordinate system is converted to The direct-axis current component and the quadrature-axis current component in a synchronously rotating coordinate system, the The synchronous rotating coordinate system rotates synchronously with the magnetic field of the rotor's permanent magnet. The d-axis coincides with the direct axis of the rotor's permanent magnet, and the q-axis leads the d-axis by 90 electrical degrees.

[0078] The actual current components in the stationary two-phase coordinate system obtained by the Clarke transform are input into the back EMF sliding mode observer, which is based on the stepper motor in... The state equations for a stationary two-phase coordinate system are constructed as follows:

[0079]

[0080] In the formula, , They are respectively Stator current components in the coordinate system , They are respectively Stator voltage components in coordinate system , These are the back electromotive force components in the α-β coordinate system, respectively. The phase resistance of the stator winding. This refers to the phase inductance of the stator winding.

[0081] Based on sliding mode variable structure control theory, a sliding surface is constructed. The sliding surface is the difference between the observed current component and the actual current component, i.e., the sliding surface function is: ,in To observe the current vector, The actual current vector is used. An exponential reaching law is employed as the reaching law function of the sliding mode observer. The actual current component is input into the sliding mode reaching law function to calculate the observed current component containing the high-frequency chattering signal. The expression for the sliding mode reaching law function is as follows:

[0082]

[0083] In the formula, , They are respectively Observation of current components in coordinate system The sliding mode observation gain coefficient, This is a sign function that outputs 1 when the input value is greater than 0, -1 when it is less than 0, and 0 when it is equal to 0.

[0084] The difference between the actual current component and the observed current component is calculated; this difference is the output value of the sliding mode surface function. This difference is then input into a low-pass filter to filter out the high-frequency chattering signal caused by the sliding mode control and extract the observed back electromotive force. The low-pass filter is a first-order low-pass filter, and its transfer function is as follows:

[0085]

[0086] In the formula, Let be the transfer function of the low-pass filter. This is the cutoff angular frequency of the low-pass filter. For the Laplace operator.

[0087] The filtered back EMF observation value is input into the phase-locked loop (PLL). The PLL uses a proportional-integral (PI) regulator as a loop filter. The PLL's PLL uses the PPI regulator to adjust the position error signal contained in the back EMF observation value, and calculates the rotor position error. The estimated electrical angular velocity is then integrated to obtain the actual electrical angle and estimated rotational speed of the rotor. The calculation process of the phase-locked loop is as follows:

[0088]

[0089] In the formula, This is the angle error value. , These are the filtered back electromotive force observations. To estimate the rotor's electrical angular velocity, This refers to the proportional coefficient of the phase-locked loop proportional controller. The integral coefficient of the phase-locked loop integral controller. This represents the estimated actual electrical angle of the rotor. The estimated speed is the mechanical speed obtained by dividing the estimated electrical angular velocity by the number of pole pairs of the motor, and is used as the speed feedback input in the dual closed-loop control structure.

[0090] Furthermore, a dual-closed-loop control structure of an outer position loop and an inner current loop is constructed. This dual-closed-loop control structure is implemented using a three-stage cascaded configuration of a position loop, a speed loop, and a current loop. The outer position loop is the outermost loop, the inner current loop is the innermost loop, and the speed loop is located between the outer position loop and the inner current loop. Specifically, the actual rotor electrical angle output by the phase-locked loop is differentially calculated with the given electrical angle converted from the given position command sent by the host computer to obtain the position deviation. This position deviation is then input into the position proportional regulator, and the given speed is calculated through proportional adjustment. The calculation expression of the position proportional regulator is as follows:

[0091]

[0092] In the formula, For a given rotational speed, This is the proportional coefficient of the position proportional controller. Given an electrical angle, To estimate the actual electrical angle of the rotor, This is the positional deviation.

[0093] The calculated given speed is differentially calculated with the estimated speed output by the phase-locked loop to obtain the speed deviation. This speed deviation is then input into the speed proportional-integral (PI) controller, which calculates the given quadrature-axis current through proportional-integral (PI) regulation. The operational expression of the speed PI controller is as follows:

[0094]

[0095] In the formula, Given a quadrature-axis current, The proportional coefficient of the speed proportional-integral controller. The integral coefficient of the speed proportional-integral controller. This refers to the speed deviation. The direct-axis current is preset to 0 to achieve maximum torque-to-current ratio control and improve the motor's torque output efficiency.

[0096] The calculated given quadrature-axis current is differentially calculated with the actual quadrature-axis current obtained by Parker transformation to obtain the quadrature-axis current deviation. Similarly, the given direct-axis current is differentially calculated with the actual direct-axis current obtained by Parker transformation to obtain the direct-axis current deviation. These deviations are then input into the corresponding proportional-integral (PI) regulators. Through PI regulation, the given quadrature-axis voltage and the given direct-axis voltage are calculated respectively. These two voltages together constitute the given phase voltage command. The operational expression for the PI regulator is as follows:

[0097]

[0098] In the formula, The given quadrature-axis voltage is the component of the given phase voltage command in the quadrature-axis direction. This refers to the proportional coefficient of the current proportional-integral controller. The integral coefficient of the current proportional-integral controller. This is the actual quadrature-axis current. This represents the quadrature-axis current deviation. The voltage calculation expression for the direct-axis direction is the same as that for the quadrature-axis direction, and will not be repeated here. The calculated given quadrature-axis voltage and given direct-axis voltage are converted to... after inverse Parker transformation. The given phase voltage component in the stationary two-phase coordinate system is used for the subsequent generation of the space vector pulse width modulation signal, thus completing the series control of the position outer loop to the current inner loop.

[0099] Furthermore, before inputting the difference between the actual current component and the observed current component into the low-pass filter, the estimated rotational speed output from the phase-locked loop is extracted. Based on the rotational speed range in which the estimated speed falls, the observation gain coefficient in the sliding mode reaching law function is dynamically adjusted. Specifically, three rotational speed ranges—low speed, medium speed, and high speed—are pre-defined. When the estimated speed is in the low-speed range, the observation gain coefficient is configured with a first value; when the estimated speed is in the high-speed range, the observation gain coefficient is configured with a second value, where the first value is greater than the second value. When the estimated speed is in the medium-speed range, the observation gain coefficient is calculated by linear interpolation between the first and second values ​​based on the estimated speed, achieving a smooth transition of the observation gain coefficient. The dynamic adjustment expression for the observation gain coefficient is as follows:

[0100]

[0101] In the formula, This refers to the dynamically adjusted sliding mode observation gain coefficient. The first value corresponds to the observation gain in the low-speed range. The second value corresponds to the observation gain in the high-speed range, and , These are the speed thresholds for the low-speed and medium-speed ranges. These are the speed thresholds for the medium-speed and high-speed ranges.

[0102] Simultaneously, when the estimated rotational speed is in the high-speed range, a back-EMF amplitude compensation factor is calculated based on the estimated rotational speed. This factor is used to compensate for the back-EMF amplitude attenuation caused by the low-pass filter in the high-frequency range, and its value increases with the increase of the estimated rotational speed. The calculated back-EMF amplitude compensation factor is superimposed on the input of the low-pass filter to pre-compensate the amplitude of the difference signal output by the sliding mode surface function, thus offsetting the amplitude attenuation caused by the low-pass filter and ensuring the amplitude accuracy of the back-EMF observation under high-speed conditions.

[0103] In this embodiment, the parameter configurations of the sliding mode observer in different speed ranges are shown in the following table:

[0104] Table 3. Sliding mode observer parameters and corresponding speed range configuration table

[0105]

[0106] This table shows the corresponding parameter configurations for the sliding mode observer in different speed ranges, where the low-speed range corresponds to a mechanical speed of 0. At 6 r / min, a higher sliding mode observation gain coefficient is configured to improve the sensitivity of back EMF observation and the accuracy of position estimation at low speeds. In the high-speed range, corresponding to mechanical speeds of 30 r / min and above, a lower sliding mode observation gain coefficient is configured to reduce high-frequency chattering caused by sliding mode control. At the same time, the cutoff frequency of the low-pass filter is increased to reduce the phase lag of back EMF observation. The back EMF amplitude compensation factor is used to correct the amplitude attenuation caused by the filter at high speeds. In the medium-speed range, linear interpolation is used to achieve a smooth transition of parameters, ensuring stable operation and estimation accuracy of the observer across the entire speed range.

[0107] In this embodiment, a back-EMF sliding mode observer based on sliding mode variable structure control theory achieves accurate sensorless estimation of rotor position and speed across the entire speed range, eliminating the need for additional position sensors and reducing system cost and hardware complexity. By dynamically adjusting the sliding mode observation gain coefficient and filter parameters, the system adapts to the motor operating characteristics of different speed ranges. High gain is used in the low-speed range to improve observation sensitivity, while low gain is used in the high-speed range to reduce chattering. An amplitude compensation factor ensures observation accuracy. A dual-closed-loop control structure with a three-stage cascaded series connection of position, speed, and current enables precise control of the motor's operating state, improving the system's dynamic response performance and steady-state control accuracy, and ensuring stable motor operation across the entire operating range.

[0108] In yet another preferred embodiment, reference is made to Figure 5 and Figure 6 Based on the speed deviation calculation process of the aforementioned embodiment, the threshold judgment process of speed deviation and the smooth switching process of adaptive step angle adjustment are further refined.

[0109] The current operating parameters of the stepper motor are obtained, including the estimated speed and the current microstepping value. Further, the operating parameters also include the winding temperature and bus voltage of the stepper motor. Based on the obtained operating parameters, a pre-stored dynamic threshold mapping table is consulted to obtain the target dynamic threshold corresponding to the current operating parameters. This dynamic threshold mapping table is pre-obtained through motor bench calibration tests and stored in the controller's non-volatile memory. The target dynamic threshold in the dynamic threshold mapping table decreases with increasing estimated speed, increases with increasing microstepping value, increases with increasing winding temperature, and increases with decreasing bus voltage. Specifically, the dynamic threshold mapping table is a four-dimensional mapping table, using estimated speed, current microstepping value, winding temperature, and bus voltage as four input dimensions, and an initial dynamic threshold as the output dimension. The obtained winding temperature, bus voltage, estimated speed, and current microstepping value are used as input vectors, and a linear interpolation query is performed in the four-dimensional dynamic threshold mapping table to obtain the initial dynamic threshold. The linear interpolation query expression is as follows:

[0110]

[0111] In the formula, As the initial dynamic threshold, It is a four-dimensional linear interpolation function. To estimate the rotational speed, This represents the current number of microstep subdivisions. This refers to the winding temperature of the stepper motor. This is the actual value of the bus voltage.

[0112] Further, the stepper motor's stepout trigger frequency within a historical operating cycle is obtained. This stepout trigger frequency is the ratio of the number of times the motor triggers stepout protection to the total operating time within a preset historical statistical period. The stepout trigger frequency is multiplied by a preset penalty coefficient to obtain a threshold offset. This penalty coefficient is a pre-tuned positive coefficient; the higher the stepout trigger frequency, the larger the threshold offset. The calculated threshold offset is then added to the aforementioned initial dynamic threshold to obtain the final target dynamic threshold. The calculation expression for the final target dynamic threshold is as follows:

[0113]

[0114] In the formula, The final target dynamic threshold, The preset penalty coefficient, This refers to the stepper motor's step loss trigger frequency during its historical operating cycles.

[0115] The absolute value of the calculated speed deviation is compared with the final target dynamic threshold. If the absolute value of the speed deviation is greater than the target dynamic threshold, it is determined that the speed deviation exceeds the preset threshold and the motor is at risk of losing steps, triggering the adaptive step angle adjustment mechanism. If the absolute value of the speed deviation is less than or equal to the target dynamic threshold, it is determined that the speed deviation is within the preset threshold and the motor is at no risk of losing steps, maintaining the current microstep subdivision or restoring to the initial microstep subdivision.

[0116] When the adaptive step angle adjustment mechanism is triggered, the current microstep subdivision is dynamically reduced to increase the single-step electromagnetic torque. Specifically, when the adaptive step angle adjustment mechanism is triggered, the first phase current given amplitude corresponding to the current microstep subdivision and the second phase current given amplitude corresponding to the reduced target microstep subdivision are obtained. Both the first and second phase current given amplitudes are pre-stored in the microstep subdivision parameter mapping table. The phase current given amplitude increases as the microstep subdivision decreases to match the increase in single-step electromagnetic torque. Within the transition interval from the current pulse width modulation period to the next pulse width modulation period, linear ramp interpolation is performed on the first and second phase current given amplitudes to obtain the transition phase current given amplitude at each moment within the transition interval. The calculation expression for the linear ramp interpolation is as follows:

[0117]

[0118] In the formula, To assign an amplitude to the transition phase current within the transition interval. The amplitude of the first phase current corresponding to the current microstep subdivision is given. The amplitude of the second-phase current corresponding to the target microstep subdivision is given. The duration of the transition period. This is the start time of the transition interval. For the current moment, and The duration of the transition interval is configured as an integer multiple of the pulse width modulation period to ensure that the interpolation calculation process is synchronized with the pulse width modulation period. The calculated amplitude of the transition phase current is used as the input reference for the inner current loop, so that the phase current amplitude changes linearly and continuously during the microstep subdivision switching process, avoiding torque impact and operating noise caused by sudden changes in phase current amplitude.

[0119] Furthermore, while performing linear ramp interpolation calculations on the given amplitudes of the first and second phase currents, the first electrical angle step resolution under the current microstepping level and the second electrical angle step resolution under the target microstepping level are obtained. The electrical angle step resolution is the electrical angle increment corresponding to a single microstep and is inversely proportional to the microstepping level. The phase angle difference between the first and second electrical angle step resolutions is calculated. This phase angle difference is the difference in electrical angle increment within a single control cycle before and after the microstepping level switching. Within the transition interval of the linear ramp interpolation calculation, the phase angle difference is synchronously superimposed onto the actual electrical angle of the rotor according to the interpolation ratio to obtain the compensated actual electrical angle. The synchronous compensation calculation expression for the phase angle difference is as follows:

[0120]

[0121] In the formula, The actual electrical angle after compensation. This is the original actual electrical angle output by the sliding mode observer. This is the phase angle difference between the first electrical angle step resolution and the second electrical angle step resolution; the other parameters are defined in the aforementioned linear interpolation formula.

[0122] The compensated actual electrical angle is used as the feedback input for the outer position loop, ensuring a continuous and smooth change in the position feedback signal during microstep increment switching. This eliminates spatial electrical angle mapping discontinuities caused by microstep increment switching and avoids control loop disturbances and torque fluctuations due to abrupt changes in the position feedback signal. When the speed deviation returns to within the target dynamic threshold, the microstep increment is gradually increased using the same linear interpolation and phase compensation method as described above for reducing the microstep increment, eventually restoring it to the initial microstep increment. This ensures the smoothness of the motor's steady-state operation and the accuracy of position control.

[0123] In this embodiment, the motor operating parameters corresponding to different microstep subdivisions are shown in the table below:

[0124] Table 4. Mapping Table of Microstep Subdivisions and Corresponding Operating Parameters

[0125]

[0126] This table lists the electromechanical characteristic parameters of stepper motors corresponding to different microstepping subdivisions. The microstepping subdivision is inversely proportional to the electrical angle per step; a lower subdivision results in a larger electrical angle per step, a higher maximum output electromagnetic torque per step, and the ability to handle more extreme load and power supply fluctuations. A higher subdivision results in higher electrical angle stepping resolution, better motor smoothness, and higher position control accuracy, making it suitable for steady-state high-precision control conditions. In this embodiment, the adaptive step angle adjustment mechanism is based on this mapping table, dynamically selecting the corresponding microstepping subdivision according to the motor's operating state, achieving an optimal balance between anti-step loss capability and operational smoothness.

[0127] In this embodiment, a four-dimensional dynamic threshold mapping table based on multi-dimensional operating condition parameters is used to achieve dynamic adaptive adjustment of the out-of-step risk judgment threshold. This allows the out-of-step judgment conditions to adapt to the electromechanical characteristics of the motor under different speeds, microstep increments, winding temperatures, and bus voltages, improving the accuracy of out-of-step risk judgment and avoiding false triggering and missed triggering. By introducing a penalty mechanism based on historical out-of-step trigger frequencies, threshold adaptive correction is performed for operating conditions that frequently trigger out-of-step protection, further improving the system's operational reliability. Through current linear ramp interpolation and phase angle synchronous compensation during microstep increment switching, a shock-free smooth switching of microstep increments is achieved, eliminating sudden changes in current amplitude and spatial electrical angle mapping discontinuities during the switching process. This avoids torque impacts, operating noise, and control loop disturbances during the switching process, improving the motor's anti-out-of-step capability while ensuring the motor's operational stability and control accuracy.

Claims

1. A control method for a stepper motor in automobiles, characterized in that, Includes the following steps: The bus voltage of the stepper motor is acquired in real time, the voltage fluctuation compensation coefficient is calculated, and the voltage fluctuation compensation coefficient is superimposed on the given phase voltage command of the current inner loop. The phase current of the stepper motor is collected in real time and input to the back EMF sliding mode observer. The actual electrical angle and estimated speed of the rotor are estimated through the back EMF sliding mode observer. A dual closed-loop control structure consisting of an outer position loop and an inner current loop is constructed, and the speed deviation is obtained by differential calculation between the estimated speed and the given speed. Determine whether the speed deviation exceeds a preset threshold. If the speed deviation exceeds the preset threshold, trigger the adaptive step angle adjustment mechanism to dynamically reduce the current microstep subdivision to increase the single-step electromagnetic torque. If the speed deviation returns to within the preset threshold, the initial microstep subdivision is restored, and a drive signal is generated based on the adjusted microstep subdivision to control the operation of the stepper motor. The dual closed-loop control structure of the outer position loop and the inner current loop includes: calculating the position deviation by differentially calculating the actual electrical angle of the rotor and the given electrical angle, and inputting the position deviation into the position proportional regulator to calculate the given speed; The speed deviation is calculated by differentiating the given speed from the estimated speed, and the speed deviation is input to the speed proportional-integral controller to calculate the given quadrature-axis current; The current deviation is calculated by differential calculation between the given quadrature-axis current and the actual quadrature-axis current. The current deviation is then input into the current proportional-integral regulator to calculate the given phase voltage command, thus completing the series control of the position outer loop on the current inner loop. Determining whether the speed deviation exceeds a preset threshold includes: obtaining the current operating condition parameters of the stepper motor, wherein the operating condition parameters include the estimated speed and the current microstep subdivision. Based on the operating condition parameters, a pre-stored dynamic threshold mapping table is queried to obtain the target dynamic threshold corresponding to the current estimated speed and the current microstep subdivision number. The target dynamic threshold in the dynamic threshold mapping table decreases as the estimated speed increases. The absolute value of the speed deviation is compared with the target dynamic threshold. If the absolute value of the speed deviation is greater than the target dynamic threshold, it is determined that the speed deviation exceeds the preset threshold. The operating parameters also include the winding temperature of the stepper motor and the bus voltage; The step of querying a pre-stored dynamic threshold mapping table based on the operating condition parameters to obtain the target dynamic threshold corresponding to the current estimated speed and the current microstep subdivision number includes: taking the winding temperature, the bus voltage, the estimated speed and the current microstep subdivision number as input vectors, performing interpolation query in the four-dimensional dynamic threshold mapping table to obtain the initial dynamic threshold; Obtain the step loss trigger frequency of the stepper motor in the historical operating cycle, and multiply the step loss trigger frequency by the penalty coefficient to obtain the threshold offset; The threshold offset is superimposed on the initial dynamic threshold to obtain the final target dynamic threshold.

2. The control method for a stepper motor for automobiles according to claim 1, characterized in that, The step of calculating the voltage fluctuation compensation coefficient and superimposing the voltage fluctuation compensation coefficient into the given phase voltage command of the current inner loop includes: obtaining the voltage deviation value between the current bus voltage and the rated bus voltage, and calculating the ratio of the voltage deviation value to the rated bus voltage to obtain the voltage fluctuation ratio; The voltage fluctuation ratio is input into the feedforward compensation function to calculate the voltage fluctuation compensation coefficient; Within each pulse width modulation cycle, the duty cycle of the base phase voltage output of the inner current loop is obtained, and the duty cycle of the base phase voltage is divided by the voltage fluctuation compensation coefficient to obtain the corrected target phase voltage duty cycle. A space vector pulse width modulation signal is generated based on the target phase voltage duty cycle.

3. The control method for a stepper motor for automobiles according to claim 1, characterized in that, The step of inputting the phase current to the back EMF sliding mode observer and estimating the actual electrical angle and estimated speed of the rotor through the back EMF sliding mode observer includes: performing Clark transformation and Park transformation on the phase current to obtain the actual current components in the stationary two-phase coordinate system. The actual current component is input into the sliding mode approach law function to calculate the observed current component containing the high-frequency chattering signal; Calculate the difference between the actual current component and the observed current component, input the difference into a low-pass filter to filter out high-frequency signals, and extract the back electromotive force observation value; The back electromotive force observation value is input into the phase-locked loop, and the actual electrical angle of the rotor and the estimated speed are calculated by the proportional-integral regulator of the phase-locked loop.

4. The control method for a stepper motor for automobiles according to claim 1, characterized in that, The method of dynamically reducing the current microstep subdivision number to increase the single-step electromagnetic torque includes: when the adaptive step angle adjustment mechanism is triggered, obtaining the first phase current given amplitude corresponding to the current microstep subdivision number at the current moment, and the second phase current given amplitude corresponding to the reduced target microstep subdivision number. During the transition interval from the current pulse width modulation cycle to the next pulse width modulation cycle, linear ramp interpolation is performed on the given amplitude of the first phase current and the given amplitude of the second phase current to obtain the given amplitude of the transition phase current. The given amplitude of the transition phase current is used as the input reference for the inner current loop.

5. The control method for a stepper motor for automobiles according to claim 2, characterized in that, After calculating the ratio of the voltage deviation value to the rated bus voltage to obtain the voltage fluctuation ratio, the method further includes: performing a first-order derivative calculation on the voltage deviation value to obtain the bus voltage change rate. The voltage differential compensation amount is obtained by multiplying the bus voltage change rate by a preset voltage change rate weighting coefficient. The voltage differential compensation is superimposed on the voltage fluctuation ratio to obtain the voltage fluctuation ratio with fused differential characteristics; The voltage fluctuation ratio of the fused differential feature is input into the feedforward compensation function to calculate the voltage fluctuation compensation coefficient of the fused differential feature, so as to correct the duty cycle of the target phase voltage in advance when the bus voltage undergoes a step change.

6. The control method for a stepper motor for automobiles according to claim 3, characterized in that, Before inputting the difference into a low-pass filter to filter out high-frequency signals, the method further includes: extracting the estimated rotational speed and dynamically adjusting the observation gain coefficient in the sliding mode approach law function according to the rotational speed range in which the estimated rotational speed is located. When the estimated rotational speed is in the low-speed range, the observation gain coefficient is configured to a first value, and when the estimated rotational speed is in the high-speed range, the observation gain coefficient is configured to a second value, wherein the first value is greater than the second value. Meanwhile, within the high-speed range, a back EMF amplitude compensation factor is calculated based on the estimated rotational speed, and the back EMF amplitude compensation factor is superimposed on the input of the low-pass filter.

7. The control method for a stepper motor for automobiles according to claim 4, characterized in that, While performing linear ramp interpolation calculation on the given amplitude of the first phase current and the given amplitude of the second phase current, the method also includes: obtaining the first electrical angle step resolution under the current microstep subdivision number and the second electrical angle step resolution under the target microstep subdivision number. Calculate the phase angle difference between the first electrical angle step resolution and the second electrical angle step resolution; Within the transition interval of the linear ramp interpolation calculation, the phase angle difference is synchronously superimposed onto the actual electrical angle of the rotor according to the interpolation ratio to obtain the compensated actual electrical angle. The compensated actual electrical angle is used as the feedback input of the outer position loop.

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