Wide voltage multi-mode control method and system for three-phase dual active bridge dc converter

Abstract

A wide-voltage multi-mode control method and system for a three-phase dual active bridge DC-DC converter includes: obtaining a reference external shift ratio and voltage gain; calculating a gain adaptive factor and a power adaptive factor through two preset nonlinear functions; calculating a gain characteristic quantity based on the relationship between the voltage gain and the gain adaptive factor, and calculating a power characteristic quantity based on the relationship between the voltage gain, the power adaptive factor, and the reference external shift ratio; determining the operating condition based on the gain characteristic quantity and the power characteristic quantity, and setting different operating modes; setting different first phase shift angles, second phase shift angles, and third phase shift angles for different operating modes; and selecting either a PS control mode or a cyclic rolling parallel phase shift mode according to the operating mode to convert the corresponding first phase shift angle, second phase shift angle, and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC-DC converter. This invention achieves a higher power transmission upper limit and soft switching at high power levels.

Description

Technical Field

[0001] This invention belongs to the field of DC-DC converter control technology, and more specifically, relates to a wide-voltage multi-mode control method and system for a three-phase dual active bridge DC-DC converter. Background Technology

[0002] Electric vehicles, as an effective way to consume new energy sources, are gradually becoming a mainstay in the transportation sector with strong government support. However, problems such as long charging times, limited battery range, and a shortage of charging stations restrict the further promotion and popularization of electric vehicles, especially in remote areas where the problem is more pronounced, greatly reducing consumer enthusiasm for purchasing them. Figure 1 Vehicle-to-vehicle energy sharing is a feasible solution to alleviate range anxiety for car owners. If an electric vehicle experiences insufficient battery power while driving, the owner can use a portable energy sharing device to seek quick energy replenishment from other car owners to improve battery range.

[0003] Dual active bridge (DAB) DC-DC converters are widely used in energy exchange due to their advantages such as electrical isolation, bidirectional energy flow, and ease of soft switching. From a topology perspective, a three-phase dual active bridge (3p-DAB) DC-DC converter can be viewed as three single-phase dual active bridge (1p-DAB) DC-DC converters connected in parallel with phase shifts. For the same rated power, the 3p-DAB offers better magnetic material utilization and a significantly smaller DC-link capacitor, resulting in higher power density and transmission capacity. Therefore, to achieve portable, miniaturized, and fast-charging energy exchange devices, the 3p-DAB is often the preferred converter.

[0004] Currently, single-phase shift (PS) control is still the dominant control method for 3p-DAB converters, as it is simple, reliable, and easy to implement. In addition, novel control methods have been proposed. Method 1 proposes a numerical algorithm for obtaining the minimum conduction loss modulation scheme using duty cycle control (DCC) with three degrees of freedom across the entire load range. Method 2 proposes an asymmetric phase shift (APS) modulation scheme, introducing variable phase shifts between the converter phase legs to achieve the minimum root mean square current (RMS). Method 3 proposes a synchronous PWM control method that enables two phase arms in a 3p-DAB to operate in parallel, thus effectively transforming it into a 1p-DAB.

[0005] In reality, the charging interface voltage levels of electric vehicles range from 400V to 700V, from microcars to large vehicles. Therefore, the voltage gain for energy exchange is not fixed, exhibiting a variable voltage adjustment range of 0.5-2. Traditional PS control, when the voltage gain is not 1, especially under light load conditions, increases switching losses due to the narrow operating range of zero-voltage switching (ZVS). Furthermore, the PS control's single degree of freedom results in a limited operating mode, failing to optimize minimum current stress operation based on the actual circuit conditions, thus failing to reduce conduction losses. These transmission losses are highly detrimental to batteries with limited energy. Of the proposed novel methods, Method 1 has a complex derivation process; once parameters change, the numerical tables need to be recalculated, resulting in a large computational load and lacking universality, making further practical application difficult. Method 2 uses synchronous parallel operation between converter phase legs, leading to different current flows for each switch, resulting in uneven losses and excessive current stress. Method 3 results in an average current in the independent operating phase that is twice that of the parallel operating phase. This is detrimental to device selection and thermal design, and lowers the power transmission limit, making it difficult to apply to high-power transmission scenarios. Furthermore, methods 1-3 only address boosting or bucking scenarios, thus unsuitable for energy sharing in electric vehicles. Additionally, methods 1 and 2 involve switching between multiple control states but lack technical solutions. If the system is subjected to external disturbances, the dynamic response time will increase due to mode oscillations, potentially leading to system collapse in severe cases. Summary of the Invention

[0006] To address the shortcomings of existing technologies, this invention provides a wide-voltage multi-mode control method and system for a three-phase dual active bridge DC-DC converter.

[0007] The present invention adopts the following technical solution.

[0008] The first aspect of this invention proposes a wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter, wherein the three-phase dual active bridge DC-DC converter consists of a three-phase full-bridge inverter and a three-phase full-bridge rectifier; both the three-phase full-bridge inverter and the three-phase full-bridge rectifier include six output switching transistors, specifically: Output reference voltage U ref and output sampling voltage U 2_m The difference input PI control algorithm obtains the reference shift compared to; Based on the transformer turns ratio and the input sampling voltage U 1_m and output sampling voltage U 2_m Calculate the voltage gain; Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U refThe gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively. The gain characteristic is calculated based on the relationship between voltage gain and gain adaptive factor, and the power characteristic is calculated based on the relationship between voltage gain, power adaptive factor and reference shift ratio. The current operating condition is determined based on the gain characteristic, power characteristic and preset operating condition judgment conditions, and different operating modes are set for different operating conditions. Different first phase shift angle, second phase shift angle and third phase shift angle are set for different operating modes. Depending on the operating mode, the PS control mode or the cyclic rolling parallel phase shift mode is selected to convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter.

[0009] Preferably, the step of adjusting the transformer turns ratio and the input sampling voltage U... 1_m and output sampling voltage U 2_m Calculate the voltage gain as follows: The set transformer turns ratio multiplied by the output sampling voltage U 2_m Divide by the input sampling voltage U 1_m The voltage gain is obtained.

[0010] Preferably, the calculation of the gain adaptive factor specifically involves: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; Multiply the difference between the natural base e and 1 by the difference between 1 and the voltage tracking error, then add 1 to calculate the ln function of the sum; subtract the ln function result from 1 to obtain the gain adaptive factor.

[0011] Preferably, the power adaptive factor is specifically: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; The negative value of the voltage tracking error is used as the exponent, and the natural base e is used as the base. An exponential operation is performed, and the result of the exponential operation is multiplied by a set optimization adjustment constant to obtain the power adaptive factor.

[0012] Preferably, the gain characteristic is calculated based on the relationship between the voltage gain and the gain adaptive factor, specifically as follows: When the voltage gain is less than the difference between 1 and the gain adaptive factor, the gain characteristic is 0; when the voltage gain is greater than or equal to the difference between 1 and the gain adaptive factor and less than the sum of 1 and the gain adaptive factor, the gain characteristic is 1; when the voltage gain is greater than or equal to the sum of 1 and the gain adaptive factor, the gain characteristic is 2.

[0013] Preferably, the power characteristic quantity is calculated based on the relationship between the voltage gain, the power adaptive factor, and the reference shift ratio; When the voltage gain is less than 1, if the difference between the reference shift ratio and the voltage gain is greater than or equal to 0 and less than 0.5 times 1, then the power characteristic is 0. If the difference between the reference shift ratio and the voltage gain is greater than or equal to 0.5 times 1 and less than the difference between 1 divided by 2 and the power adaptive factor, then the power characteristic is 1. If the difference between 1 divided by 2 and the power adaptive factor is greater than or equal to 1, then the power characteristic is 2. When the voltage gain is greater than or equal to 1, if the difference between the reference shift ratio and the voltage gain is greater than or equal to 0 and less than 0.5 times 1, the power characteristic is 3. If the difference between the reference shift ratio and the voltage gain is greater than or equal to 0.5 times 1 and less than the difference between 1 divided by 2 and the power adaptive factor, the power characteristic is 4. If the difference between 1 divided by 2 and the power adaptive factor is greater than or equal to 1, the power characteristic is 2.

[0014] Preferably, the current operating condition is determined based on the gain characteristic quantity, the power characteristic quantity, and preset operating condition judgment conditions, specifically as follows: The operating conditions include buck low-power transmission, buck medium-power transmission, boost low-power transmission, boost medium-power transmission, voltage gain of 1, and high-power transmission. When the gain characteristic is 0 and the power characteristic is 0, it operates in buck low power transmission mode and selects the first working mode. When the gain characteristic is 0 and the power characteristic is 1, it operates in buck power transmission mode and selects the second working mode. When the gain characteristic is 2 and the power characteristic is 3, it operates in the boost low power transmission condition and selects the third working mode. When the gain characteristic is 2 and the power characteristic is 4, it operates in the boost power transmission mode and selects the fourth working mode. When the gain characteristic is 1 or the power characteristic is 2, it operates in the voltage gain of 1 or high power transmission condition, and selects the fifth working mode.

[0015] Preferably, for different operating modes, different first phase shift angles, second phase shift angles, and third phase shift angles are set, specifically as follows: For the first operating mode, the first shift ratio is twice the voltage gain multiplied by the reference shift ratio and then divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second shift ratio, it is twice the reference shift ratio divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second operating mode, the first shift ratio is: multiply the difference between 1 and the voltage gain by the difference between 1 and 2 times the reference external shift ratio, then divide by the voltage gain, and subtract the division result from 1 to obtain the first shift ratio. The second shift ratio is 1. For the third operating mode, the first shift ratio is twice the voltage gain divided by the difference between the voltage gain and 1, and then multiplied by the reference shift ratio; the second shift ratio is twice the difference between the voltage gain and 1, and then multiplied by the reference shift ratio. For the fourth operating mode, the first shift ratio is 1; the second shift ratio is the difference between 2 times the voltage gain and 1, multiplied by the reference external shift ratio, plus 2, and then minus the voltage gain. For the first, second, third, and fourth working modes, the third shift ratio is equal to the reference shift ratio plus 0.5 times the corresponding first shift ratio minus 0.5 times the corresponding second shift ratio; For the fifth operating mode, the first shift ratio and the second shift ratio are both 0, and the third shift ratio is the difference between the reference external shift ratio and 0.5, multiplied by 0.5 and the difference between 0.5 and the boundary power constant, then divided by 0.5, plus the boundary power constant; the boundary power constant is a set value. For all operating modes, the phase angles of the first, second, third, fourth, and fifth shifts are equal to the ratio of the first, second, third, fourth, and fifth shifts multiplied by the first set period, respectively.

[0016] Preferably, the selection of PS control mode or cyclic rolling parallel phase shift mode converts the corresponding first phase shift angle, second phase shift angle, and third phase shift angle into modulation waves corresponding to the output switching transistors of the three-phase dual active bridge DC converter, specifically as follows: Four source drive signals are generated. All four source drive signals are PWM wave signals with the same period (first set period), equal amplitude, and duty cycle of 0.5. The second source drive signal lags the first source drive signal by a first phase shift angle; the fourth source drive signal lags the first source drive signal by a second phase shift angle; and the third source drive signal lags the first source drive signal by a third phase shift angle. When in the fifth working mode, select the cyclic rolling parallel phase shift control mode; otherwise, select the PS control mode. Based on the selected cyclic rolling parallel phase-shifting control mode or PS control mode, the four source drive signals are processed with different set time delays to generate the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier. In a three-phase full-bridge inverter and a three-phase full-bridge rectifier, the drive signal of the upper output switch of each bridge arm is equal to the corresponding bridge arm drive signal, and the drive signal of the lower output switch of each bridge arm is equal to the inverted signal of the corresponding bridge arm drive signal. Preferably, if the PS control mode is selected, the three bridge arm drive signals of the three-phase full-bridge inverter are G1, G1 with a delay of 120° and G1 with a delay of 240°, respectively. The three bridge arm drive signals of the three-phase full-bridge rectifier are G3, G3 with a delay of 120°, and G3 with a delay of 240°. G1 and G3 are the first and third source drive signals, respectively.

[0017] Preferably, if the selected mode is the cyclic rolling parallel phase-shifting control mode, the period of the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier becomes six times the first set period; each first set period is one mode. First mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G2 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G4 respectively. Second mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G3, G3, and G4. Third mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G4. Fourth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G1 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G3 respectively. Fifth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G4, G4, and G3. Sixth mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G3. G1, G2, G3, and G4 are the first source drive signal, the second source drive signal, the third source drive signal, and the fourth source drive signal, respectively.

[0018] The second aspect of this invention proposes a wide-voltage multi-mode control system for a three-phase dual active bridge DC-DC converter based on the method described in the first aspect of this invention, comprising a PI controller, a multi-objective joint control unit, a dual-mode reconfigurable modulated wave generator, and a two-dimensional adaptive dynamic threshold condition identification unit, specifically: PI controller: used to determine the output reference voltage U ref and output sampling voltage U 2_m The difference is compared to the reference value shifted outward; Two-dimensional adaptive dynamic threshold condition identification unit: used to identify the transformer turns ratio and input sampling voltage U. 1_m and output sampling voltage U 2_m Calculate the voltage gain; output the reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively; the gain characteristic is calculated based on the relationship between voltage gain and gain adaptive factor; the power characteristic is calculated based on the relationship between voltage gain, power adaptive factor and reference shift ratio; the current working condition is determined based on the gain characteristic and power characteristic and preset working condition judgment conditions, and different working modes are set for different working conditions. Multi-objective joint control unit: used to calculate the first phase shift angle, second phase shift angle and third phase shift angle under different operating modes; Dual-mode reconfigurable modulated wave generator: used to select either PS control mode or cyclic rolling parallel phase shift mode according to the operating mode, and convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulated wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter.

[0019] The beneficial effects of the present invention are as follows: compared with the prior art, (1) the present invention can expand the soft switching range and the minimum current stress optimization space under low and medium power, and has a high power transmission upper limit under high power. Moreover, when applied to a wide voltage gain adjustment range, the logic derivation process can be simplified by using an offline lookup table, reducing the computational complexity, and making it more practical and universal in implementation.

[0020] (2) This invention proposes a dual-mode reconfigurable modulation based on a combination of cyclic rolling parallel phase shift mode and traditional PS control mode, which can make the utilization rate of the output switching tube consistent under all operating conditions, thereby uniform current stress and thermal effects, improving the reliability of switching devices and reducing the difficulty of converter thermal design.

[0021] (3) This invention proposes a method based on voltage gain d and reference shift ratio D f Performing two-dimensional adaptive dynamic threshold condition identification can reduce mode oscillations, accelerate dynamic response speed, and improve the robustness of the control system.

[0022] (4) This invention proposes a wide voltage multi-mode control system for a three-phase dual active bridge DC-DC converter for the field of vehicle-to-vehicle energy exchange. The system has a wide voltage regulation capability, a wide soft switching range and minimum current stress optimization under low and medium power, a higher power transmission limit and soft switching under high power, and fast dynamic response speed, making it more practical. Attached Figure Description

[0023] Figure 1 This is a schematic diagram of vehicle-to-vehicle energy exchange. Figure 2 This is a block diagram of the overall system structure. Figure 3 This is a flowchart of the method of the present invention; Figure 4 Here is the control flowchart for a dual-mode reconfigurable modulated wave generator; Figure 5 This is a schematic diagram of four source drive signals; Figure 6 The simulation results of three-phase leakage inductance current and voltage under different operating modes are shown in the figure. Figure 6 (a) is a simulation result of the three-phase leakage inductance current and voltage under the first operating mode; Figure 6 (b) is a simulation result diagram of the three-phase leakage inductance current and voltage under the second operating mode; Figure 6 (c) is a simulation result diagram of the three-phase leakage inductance current and voltage under the fifth operating mode; Figure 7 The result diagram is a simulation of the dynamic working process; Figure 7 (a) is a simulation result of the output voltage dynamic operation; Figure 7 (b) is a simulation result diagram of the dynamic operation of the output current; Figure 8 A schematic diagram of an experimental platform based on HIL; Figure 9 A diagram illustrating the functional division and connection of DSP and FPGA; Figure 10 The figure shows the experimental results of closed-loop control. Figure 10 (a) is a diagram showing the experimental results of closed-loop control during startup; Figure 10 (b) is a graph showing the experimental results of boost regulation closed-loop control; Figure 10 (c) is a graph showing the experimental results of the load adjustment closed-loop control. Detailed Implementation

[0024] To make the objectives, technical solutions, and advantages of this invention clearer, the technical solutions of this invention will be clearly and completely described below with reference to the accompanying drawings of the embodiments of this invention. The embodiments described in this application are merely some embodiments of this invention, and not all embodiments. Based on the spirit of this invention, all other embodiments obtained by those skilled in the art without creative effort are within the protection scope of this invention.

[0025] like Figure 3 As shown, Embodiment 1 of the present invention proposes a wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter. The three-phase dual active bridge DC-DC converter consists of a three-phase full-bridge inverter and a three-phase full-bridge rectifier. Both the three-phase full-bridge inverter and the three-phase full-bridge rectifier include six output switching transistors. The method is characterized by: Output reference voltage U ref and output sampling voltage U 2_m The difference input PI control algorithm obtains the reference shift compared to; Based on the transformer turns ratio and the input sampling voltage U 1_m and output sampling voltage U 2_m Calculate the voltage gain; Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively. The gain characteristic is calculated based on the relationship between voltage gain and gain adaptive factor, and the power characteristic is calculated based on the relationship between voltage gain, power adaptive factor and reference shift ratio. The current operating condition is determined based on the gain characteristic, power characteristic and preset operating condition judgment conditions, and different operating modes are set for different operating conditions. Different first phase shift angle, second phase shift angle and third phase shift angle are set for different operating modes. Depending on the operating mode, the PS control mode or the cyclic rolling parallel phase shift mode is selected to convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter.

[0026] It should be noted that the three-phase dual active bridge DC-DC converter, as the main circuit system, has a 3p-DAB topology, and is powered by a voltage of... The input source and voltage are The output source, labeled Q 11 -Q 26 It consists of 12 output switching transistors, energy transfer inductor L, a three-phase high-frequency transformer with a turns ratio of N, input capacitor C1, and output capacitor C2.

[0027] In this preferred embodiment, the step of basing the transformer turns ratio and the input sampling voltage U... 1_m and output sampling voltage U 2_m Calculate the voltage gain as follows: The set transformer turns ratio multiplied by the output sampling voltage U 2_m Divide by the input sampling voltage U 1_m Obtain voltage gain The formula is:

[0028] In this preferred embodiment, the calculation of the gain adaptive factor specifically involves: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; Multiply the difference between the natural base e and 1 by the difference between 1 and the voltage tracking error, then add 1 to calculate the ln function of the sum; subtract the ln function result from 1 to obtain the gain adaptive factor. The formula is:

[0029] In this preferred embodiment, the power adaptive factor is specifically: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; The negative value of the voltage tracking error is used as the exponent, and the natural base e is used as the base. An exponential operation is then performed, and the result is multiplied by a set optimization adjustment constant. Obtain the power adaptive factor The formula is:

[0030] It should be noted that the optimization adjustment constant The smaller the value, the wider the optimization range, but the lower the power transmission capability. It can be adjusted according to the actual project, and it is recommended to take 0.05-0.1.

[0031] In this preferred embodiment, the gain characteristic quantity is calculated based on the relationship between the voltage gain and the gain adaptive factor, specifically as follows: When the voltage gain is less than the difference between 1 and the gain adaptive factor, the gain characteristic is 0; when the voltage gain is greater than or equal to the difference between 1 and the gain adaptive factor and less than the sum of 1 and the gain adaptive factor, the gain characteristic is 1; when the voltage gain is greater than or equal to the sum of 1 and the gain adaptive factor, the gain characteristic is 2.

[0032] Specifically, define the gain characteristic quantity Gain. When d < 1 - m, Gain = 0; when 1 - m ≤ d ≤ 1 + m, Gain = 1; when 1 + m < d, Gain = 2; Preferably in this embodiment, the power characteristic quantity is calculated according to the magnitude relationship among the voltage gain, the power adaptation factor, and the reference external shift ratio; When the voltage gain is less than 1, if the reference external shift ratio is greater than or equal to 0 and less than 0.5 times the difference between 1 and the voltage gain, the power characteristic quantity is 0. If the reference external shift ratio is greater than or equal to 0.5 times the difference between 1 and the voltage gain and less than 1 divided by the difference between 2 and the power adaptation factor, the power characteristic quantity is 1. If it is greater than or equal to 1 divided by the difference between 2 and the power adaptation factor, the power characteristic quantity is 2; When the voltage gain is greater than or equal to 1, if the reference external shift ratio is greater than or equal to 0 and less than 0.5 times the difference between 1 and the voltage gain, the power characteristic quantity is 3. If the reference external shift ratio is greater than or equal to 0.5 times the difference between 1 and the voltage gain and less than 1 divided by the difference between 2 and the power adaptation factor, the power characteristic quantity is 4. If it is greater than or equal to 1 divided by the difference between 2 and the power adaptation factor, the power characteristic quantity is 2.

[0033] Specifically, define the power characteristic quantity Power. When d < 1, when D f| ∈[0, (1 - d) / 2), Power = 0; when D f| ∈[(1 - d) / 2, 1 / 2 - n), Power = 1; when D f| ∈[1 / 2 - n, 1), Power = 2. When d ≥ 1, when D f| ∈[0, (d - 1) / 2d), Power =3; when D f| ∈[(d - 1) / 2d, 1 / 2 - n), Power = 4; when D f| ∈[1 / 2 - n, 1), Power = 2.

[0034] [ Preferably in this embodiment, the current working condition is judged according to the gain characteristic quantity, the power characteristic quantity, and the preset working condition judgment condition. Specifically: According to the optimization target requirements of different operating states of the circuit, the working mode can be queried according to the characteristic quantities of Gain and Power, and the following rules are established: The working conditions include the step-down low-power transmission working condition, the step-down medium-power transmission working condition, the boost low-power transmission working condition, the boost medium-power transmission working condition, the working condition with a voltage gain of 1, and the high-power transmission working condition; When the gain characteristic quantity is 0 and the power characteristic quantity is 0, it operates in the step-down low-power transmission working condition and selects the first working mode; When the gain characteristic is 0 and the power characteristic is 1, it operates in buck power transmission mode and selects the second working mode. When the gain characteristic is 2 and the power characteristic is 3, it operates in the boost low power transmission condition and selects the third working mode. When the gain characteristic is 2 and the power characteristic is 4, it operates in the boost power transmission mode and selects the fourth working mode. When the gain characteristic is 1 or the power characteristic is 2, it operates in the voltage gain of 1 or high power transmission condition, and selects the fifth working mode.

[0035] Specifically, the working mode characteristic quantity Mode is defined. When Gain=1 or Power=2, Mode=5; when Gain=0 and Power=0, Mode=1; when Gain=0 and Power=1, Mode=2; when Gain=2 and Power=3, Mode=3; when Gain=2 and Power=4, Mode=4. The value of the working mode characteristic quantity corresponds to which working mode.

[0036] It should be noted that, based on the voltage gain d and the reference shift compared to D f The dual-dimensional adaptive dynamic threshold condition identification can eliminate mode oscillations, accelerate dynamic response speed, and improve the robustness of the control system. The first dimension is the voltage gain d, which is set as a gain boundary identification based on an adaptive dynamic variable threshold. Its characteristic is that it can automatically adjust the gain condition boundary according to the relationship between the circuit output voltage amplitude and the target value, so as to achieve a fast response in the transient transition phase. The second dimension is the reference shift ratio D. f It is set as a power boundary identification based on adaptive dynamic variable threshold. Its feature is that it can automatically adjust the power operating condition boundary according to the relationship between the circuit output voltage amplitude and the target value. Its function is to eliminate mode oscillations and improve system reliability when the system is in critical steady state operation.

[0037] In this preferred embodiment, different first phase shift angles, second phase shift angles, and third phase shift angles are set for different operating modes, specifically as follows: For the first operating mode, the first shift ratio is twice the voltage gain multiplied by the reference shift ratio and then divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second shift ratio, it is twice the reference shift ratio divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second operating mode, the first shift ratio is: multiply the difference between 1 and the voltage gain by the difference between 1 and 2 times the reference external shift ratio, then divide by the voltage gain, and subtract the division result from 1 to obtain the first shift ratio. The second shift ratio is 1. For the third operating mode, the first shift ratio is twice the voltage gain divided by the difference between the voltage gain and 1, and then multiplied by the reference shift ratio; the second shift ratio is twice the difference between the voltage gain and 1, and then multiplied by the reference shift ratio. For the fourth operating mode, the first shift ratio is 1; the second shift ratio is the difference between 2 times the voltage gain and 1, multiplied by the reference external shift ratio, plus 2, and then minus the voltage gain. For the first, second, third, and fourth working modes, the third shift ratio is equal to the reference shift ratio plus 0.5 times the corresponding first shift ratio minus 0.5 times the corresponding second shift ratio; For the fifth operating mode, the first shift ratio and the second shift ratio are both 0, and the third shift ratio is the difference between the reference external shift ratio and 0.5, multiplied by 0.5 and the difference between 0.5 and the boundary power constant, then divided by 0.5, plus the boundary power constant; the boundary power constant is a set value. The details are shown in Table 1 below: Table 1. Comparison of different working modes

[0038] in, For reference, shift outwards; It is the boundary power constant; , , The first, second, and third shifts are compared respectively.

[0039] Mode 1: This mode operates under buck low-power transmission conditions. In this mode, the optimized control scheme satisfies the following relationship, which can achieve minimum current stress optimization and ZVS.

[0040]

[0041] Mode 2: This mode operates in buck power transmission mode. In this mode, the optimized control scheme satisfies the following relationship, which can achieve minimum current stress optimization and ZVS.

[0042]

[0043] Mode 3: This mode operates under boost low-power transmission conditions. In this mode, the optimized control scheme satisfies the following relationship, which can achieve minimum current stress optimization and ZVS.

[0044]

[0045] Mode 4: This mode operates in the power transmission condition during boosting. At this time, the optimized control scheme satisfies the following relationship, which can achieve minimum current stress optimization and ZVS.

[0046]

[0047] Mode 5: This mode operates under voltage gain of 1 or high-power transmission conditions. In this mode, the optimized control scheme satisfies the following relationship, enabling soft switching and achieving the upper limit of high-power transmission. Where the boundary power constant is D... edge = .

[0048] For all operating modes, the phase angles of the first, second, third, fourth, and fifth shifts are equal to the ratio of the first, second, third, fourth, and fifth shifts multiplied by the first set period, respectively.

[0049] It should be noted that the above-mentioned approach achieves joint optimization of target soft switching, minimum current stress, and maximum transmission power. This scheme expands the soft switching range and minimum current stress optimization space at low to medium power levels, while possessing a high power transmission upper limit at high power levels. Furthermore, the established control scheme includes both buck and boost scenarios, is applicable to a wide voltage gain adjustment range, and simplifies the logic derivation process by utilizing offline lookup tables, reducing computational complexity and making it more practical and universally applicable.

[0050] In this preferred embodiment, such as Figure 4 As shown, the selection of PS control mode or cyclic rolling parallel phase shift mode converts the corresponding first phase shift angle, second phase shift angle, and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter, specifically: like Figure 5 As shown, four source drive signals are generated. All four source drive signals are PWM wave signals with the same period (high level amplitude is 1) and duty cycle of 0.5, all with the same period of the first set period. The second source drive signal lags the first source drive signal by the first phase shift angle; the fourth source drive signal lags the first source drive signal by the second phase shift angle; and the third source drive signal lags the first source drive signal by the third phase shift angle. When in the fifth working mode, select the cyclic rolling parallel phase shift control mode; otherwise, select the PS control mode. Based on the selected cyclic rolling parallel phase-shifting control mode or PS control mode, the four source drive signals are processed with different set time delays to generate the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier. In a three-phase full-bridge inverter and a three-phase full-bridge rectifier, the drive signal of the upper output switch of each bridge arm is equal to the corresponding bridge arm drive signal, and the drive signal of the lower output switch of each bridge arm is equal to the inverted signal of the corresponding bridge arm drive signal. In this preferred embodiment, if the PS control mode is selected, the three bridge arm drive signals of the three-phase full-bridge inverter are G1, G1 with a delay of 120°, and G1 with a delay of 240°, respectively. The three bridge arm drive signals of the three-phase full-bridge rectifier are G3, G3 with a delay of 120°, and G3 with a delay of 240°. G1 and G3 are the first source drive signal and the third source drive signal, respectively. Right now , 120° delay Time delay 240° , 120° delay The time delay is 240°, among which, , , These are the three bridge arm drive signals of the three-phase full-bridge inverter; , , These are the drive signals for the three bridge arms of the three-phase full-bridge rectifier.

[0051] In this preferred embodiment, if the selected mode is the cyclic rolling parallel phase-shifting control mode, the period of the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier becomes six times the first set period; each first set period is a mode. First mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G2 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G4 respectively. Second mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G3, G3, and G4. Third mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G4. Fourth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G1 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G3 respectively. Fifth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G4, G4, and G3. Sixth mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G3. G1, G2, G3, and G4 are the first source drive signal, the second source drive signal, the third source drive signal, and the fourth source drive signal, respectively.

[0052] Specifically, the first cycle is set to a switching cycle of T. The working cycle of the cyclic rolling parallel phase-shifting control mode is 6T to achieve one cycle. Each switching cycle is a different mode, as detailed below: Mode 1: , , , .

[0053] Mode 2: , , , .

[0054] Mode 3: , , , .

[0055] Mode 4: , , , .

[0056] Mode 5: , , , .

[0057] Modal 6: , , , In a three-phase full-bridge inverter and a three-phase full-bridge rectifier, the drive signal for the upper output switch of each bridge arm is equal to the corresponding bridge arm drive signal, and the drive signal for the lower output switch of each bridge arm is equal to the inverted signal of the corresponding bridge arm drive signal. Specifically: Generate output switch Q 11 -Q 26 drive signal g 11 -g 26 It satisfies g 11 =S A g 12 =-S A g 13 =S B g 14 =-S B g 15 =S C g 16 =-S C g 21 =S a g 22 =-S a g 23 =Sb g 24 =-S b g 25 =S c g 26 =-S c ; Among them, g 11 g 12 g 13 g 14 g 15 g 16 These are the six output switching transistors Q of the three-phase full-bridge inverter. 11 Q 12 Q 13 Q 14 Q 15 Q 16 drive signal; g 21 g 22 g 23 g 24 g 25 g 26 These are the six output switching transistors Q of the three-phase full-bridge rectifier. 21 Q 22 Q 23 Q 24 Q 25 Q 26 The driving signal; it should be noted that the negative sign in the above formula indicates the inverted signal.

[0058] It should be noted that the dual-mode reconfigurable modulation based on the combination of cyclic rolling parallel phase-shifting mode and traditional PS control mode can ensure that the utilization rate of the output switching transistors is consistent under all operating conditions, thereby uniformly dissipating current stress and thermal effects, improving the reliability of switching devices, and reducing the difficulty of converter thermal design.

[0059] Embodiment 2 of the present invention proposes a wide-voltage multi-mode control system for a three-phase dual active bridge DC-DC converter based on the method described in Embodiment 1 of the present invention, including a PI controller, a multi-objective joint control unit, a dual-mode reconfigurable modulated wave generator, and a two-dimensional adaptive dynamic threshold condition identification unit, specifically as follows: PI controller: used to determine the output reference voltage U ref and output sampling voltage U 2_m The difference is compared with the reference shift and output to the two-dimensional adaptive dynamic threshold condition identification unit and the multi-objective joint control optimization unit; Two-dimensional adaptive dynamic threshold condition identification unit: used to identify the transformer turns ratio and input sampling voltage U. 1_m and output sampling voltage U 2_m Calculate the voltage gain; output the reference voltage Uref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively; the gain characteristic is calculated based on the relationship between voltage gain and gain adaptive factor; the power characteristic is calculated based on the relationship between voltage gain, power adaptive factor and reference shift ratio; the current working condition is determined based on the gain characteristic and power characteristic and preset working condition judgment conditions, and different working modes are set for different working conditions. The signal connection is as follows: the input signal of this module is the output reference voltage U. ref Input sampling voltage U 1_m Output sampling voltage U 2_m Compared to D, the reference shift is more precise. f The output signal voltage gain d and mode characteristic Mode are transmitted to the multi-objective joint control optimization unit, and the mode characteristic Mode is transmitted to the dual-mode reconfigurable modulated wave generator.

[0060] Multi-objective joint control unit: used to calculate the first phase shift angle, second phase shift angle and third phase shift angle under different operating modes; The signal connection relationship is as follows: The input signal of this module is the reference external shift ratio D. f The mode feature quantity Mode is used to obtain the first, second, and third phase shift angles by querying the optimization control table and then inputting them into the dual-mode reconfigurable generator.

[0061] Dual-mode reconfigurable modulated wave generator: used to select either PS control mode or cyclic rolling parallel phase shift mode according to the operating mode, and convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulated wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter. Signal connection relationship: The input signal of this module is the mode characteristic quantity Mode and the first, second and third phase shift angles. After passing through the reconfigurable modulation wave generator, it generates the output switch Q of the main circuit system. 11 -Q 26 drive signal g 11 -g 26 .

[0062] Embodiment 3 of the present invention is based on the system described in Embodiment 2 (i.e., reference). Figure 2(System Overall Structure Diagram) A simulation control system based on Simulink is built. In this simulation example, the PI controller can be replaced with a model predictive controller or a fuzzy control-based PI controller to further optimize the dynamic response. Furthermore, if the requirements for the system's dynamic performance are not high in actual operation, the two-dimensional adaptive dynamic threshold condition identification unit can be replaced by a fixed threshold mode partitioning method. The control scheme provided by the multi-objective joint optimization control unit aims at the joint optimization of soft switching, minimum current stress, and maximum transmission power. In actual operation, other parameters such as minimum reactive power and minimum heat loss can be further optimized.

[0063] When d=0.5, D f =0.15, when the load output power is 1500W, the system is operating in Mode 1. The three-phase leakage inductance current and voltage are as follows: Figure 6 (a). When d=0.5, D f =0.40, when the load output power is 6000W, the system is operating in Mode 2. The three-phase leakage inductance current and voltage are as follows: Figure 6 (b). When d=0.5, D f =0.85, three-phase leakage inductance current and voltage as follows Figure 6 (c) It was verified that the system built according to this technical method can adjust its working mode according to different working conditions to meet the target optimization requirements.

[0064] Simulation results of converter startup, boost regulation, and load regulation dynamic operation are as follows: Figure 7 As shown in (a), the simulation results of the three-phase leakage inductance current under boost regulation and load regulation are as follows. Figure 7 As shown in (b). According to the simulation results, after the converter starts up at time 0s, the output voltage can quickly track the reference voltage and reach a steady state. At time 0.1s, the reference voltage is adjusted from 450V to 500V, and the output voltage can quickly track it. At time 0.2s, the load resistance decreases from 20Ω to 10Ω, and the load power increases from 12.5kW to 25kW. The output voltage can stabilize at 500V after a brief disturbance. According to the simulation results, According to the simulation results, there are brief mode oscillations during the transition period of boost regulation and load regulation. However, the system can quickly stabilize, which verifies that the technical solution still has good dynamic response performance during transient processes.

[0065] Embodiment 3 of the present invention is based on the system described in Embodiment 2 and uses the StarSim HIL semi-physical system to build a portable energy exchange device; To alleviate range anxiety and charging inconvenience for users, vehicle-to-vehicle energy sharing has gained widespread attention as a flexible, distributed energy replenishment method. This model allows electric vehicles to request emergency power from other vehicles while driving via portable energy sharing devices, thereby quickly extending their range and enhancing travel flexibility, especially valuable in areas with weak charging infrastructure coverage. However, in practical applications, different vehicle models have significantly different charging interface voltage levels due to differences in battery system design and platform, ranging from 400V to 700V or even wider. This means that when performing vehicle-to-vehicle energy sharing, voltage mismatch often exists between the output and input ends, and direct energy transfer may lead to inefficiency, equipment damage, or even safety hazards.

[0066] Therefore, to achieve safe, efficient, and universally applicable vehicle-to-vehicle energy exchange, energy transfer devices need to possess wide-range, high-precision voltage regulation capabilities. Typical voltage gain should cover an adjustable range of 0.5 to 2 to accommodate voltage differences between different vehicles. Existing common charging or energy exchange devices are mostly designed for fixed voltage levels, with limited adjustment ranges, making it difficult to meet the flexible and efficient energy exchange needs under complex field conditions. Therefore, there is an urgent need to develop a portable energy exchange device with wide voltage regulation capabilities, high efficiency, and high reliability to support safe and convenient energy exchange between electric vehicles, further enhancing electric vehicle users' confidence in their travel and contributing to the green and low-carbon transformation of the transportation sector.

[0067] Against this backdrop, this embodiment provides a case study of a portable energy exchange device that applies the technical methods of this invention, based on the StarSim HIL hardware-in-the-loop experimental platform.

[0068] Considering circuit design costs and control algorithm development efficiency, a schematic diagram of an experimental platform based on StarSim HIL is shown below. Figure 8 As shown. The platform's hardware mainly consists of a HIL simulator for storing the main circuit model information, a sampling circuit, and a controller composed of a DSP and an FPGA. Additionally, a host computer and an oscilloscope are required for real-time observation and modification of circuit parameters. The main circuit model parameters are imported into the HIL simulator via the host computer. The sampling information is connected to the sampling circuit via the HIL's I / O interface, and after filtering by the sampling circuit, it is transmitted to the control system.

[0069] The DSP primarily implements digital filtering, PI control, two-dimensional adaptive dynamic threshold condition identification, and information interaction. The FPGA primarily implements memory, a multi-objective joint optimization control unit, a dual-mode reconfigurable modulated wave generator, and information interaction. The functional division and connection between the DSP and FPGA are as follows: Figure 9As shown. The DSP is connected to the sampling circuit, performs digital filtering on the sampled signal, calculates the algorithm, and then outputs the control information D. f The data is sent to the memory in the FPGA using 16-bit parallel communication. The FPGA continuously scans D in small time steps. f Then, by consulting the corresponding optimization control table for D1, D2, and D3, 12 drive signals g are ultimately generated. 11 -g 26 And then send it to the HIL simulator. In addition, the FPGA will continuously scan the input / output voltages and D in the memory with large time steps. f The data is then transmitted to a host computer to monitor the circuit's operating status in real time and to control it.

[0070] The closed-loop control experimental results based on the StarSim HIL hardware-in-the-loop experimental platform are as follows: Figure 10 As shown, during converter startup, boost regulation, and load regulation, the circuit exhibits brief mode oscillations. A well-designed controller can respond quickly. Analysis of the A-phase leakage inductance current waveform during the transient period reveals that Mode 5, with its higher transmission power, can quickly bring the output voltage to track the reference voltage; therefore, the converter primarily operates in Mode 5 during the transient state. Based on closed-loop experimental results, this invention demonstrates that the output voltage remains controllable after brief oscillations under both input and output disturbances, verifying the feasibility of this technical method.

[0071] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and not to limit it. Although the present invention has been described in detail with reference to the above embodiments, those skilled in the art should understand that modifications or equivalent substitutions can still be made to the specific implementation of the present invention. Any modifications or equivalent substitutions that do not depart from the spirit and scope of the present invention should be covered within the protection scope of the claims of the present invention.

Claims

1. A wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter, wherein the three-phase dual active bridge DC-DC converter consists of a three-phase full-bridge inverter and a three-phase full-bridge rectifier; both the three-phase full-bridge inverter and the three-phase full-bridge rectifier include six output switching transistors, characterized in that: Output reference voltage U ref and output sampling voltage U 2_m The difference input PI control algorithm obtains the reference shift compared to; Based on the transformer turns ratio and the input sampling voltage U 1_m and output sampling voltage U 2_m Calculate the voltage gain; Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively. The gain characteristic is calculated based on the relationship between voltage gain and gain adaptation factor, and the power characteristic is calculated based on the relationship between voltage gain, power adaptation factor and reference shift ratio. The current operating condition is determined based on the gain characteristic, power characteristic and preset operating condition judgment conditions. Different operating conditions are set with different operating modes; different operating modes are set with different first phase shift angle, second phase shift angle and third phase shift angle. Depending on the operating mode, the PS control mode or the cyclic rolling parallel phase shift mode is selected to convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter.

2. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 1, characterized in that: The above is based on the transformer turns ratio and the input sampling voltage U. 1_m and output sampling voltage U 2_m Calculate the voltage gain as follows: The set transformer turns ratio multiplied by the output sampling voltage U 2_m Divide by the input sampling voltage U 1_m The voltage gain is obtained.

3. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 1, characterized in that: The calculation of the adaptive gain factor is specifically as follows: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; Multiply the difference between the natural base e and 1 by the difference between 1 and the voltage tracking error, then add 1 to calculate the ln function of the sum; subtract the ln function result from 1 to obtain the gain adaptive factor.

4. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 1, characterized in that: The power adaptive factor is specifically: Output reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The absolute value of the proportion is used as the voltage tracking error; The negative value of the voltage tracking error is used as the exponent, and the natural base e is used as the base. An exponential operation is performed, and the result of the exponential operation is multiplied by a set optimization adjustment constant to obtain the power adaptive factor.

5. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 1, characterized in that: The gain characteristic is calculated based on the relationship between the voltage gain and the gain adaptive factor, specifically: When the voltage gain is less than the difference between 1 and the gain adaptive factor, the gain characteristic is 0; when the voltage gain is greater than or equal to the difference between 1 and the gain adaptive factor and less than the sum of 1 and the gain adaptive factor, the gain characteristic is 1; when the voltage gain is greater than or equal to the sum of 1 and the gain adaptive factor, the gain characteristic is 2.

6. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 5, characterized in that: The power characteristic is calculated based on the relationship between voltage gain, power adaptive factor, and reference shift ratio. When the voltage gain is less than 1, if the difference between the reference shift ratio and the voltage gain is greater than or equal to 0 and less than 0.5 times 1, then the power characteristic is 0. If the difference between the reference shift ratio and the voltage gain is greater than or equal to 0.5 times 1 and less than the difference between 1 divided by 2 and the power adaptive factor, then the power characteristic is 1. If the difference between 1 divided by 2 and the power adaptive factor is greater than or equal to 1, then the power characteristic is 2. When the voltage gain is greater than or equal to 1, if the difference between the reference shift ratio and the voltage gain is greater than or equal to 0 and less than 0.5 times 1, the power characteristic is 3. If the difference between the reference shift ratio and the voltage gain is greater than or equal to 0.5 times 1 and less than the difference between 1 divided by 2 and the power adaptive factor, the power characteristic is 4. If the difference between 1 divided by 2 and the power adaptive factor is greater than or equal to 1, the power characteristic is 2.

7. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 6, characterized in that: The current operating condition is determined based on the gain characteristic, power characteristic, and preset operating condition judgment conditions, specifically as follows: The operating conditions include buck low-power transmission, buck medium-power transmission, boost low-power transmission, boost medium-power transmission, voltage gain of 1, and high-power transmission. When the gain characteristic is 0 and the power characteristic is 0, it operates in buck low power transmission mode and selects the first working mode. When the gain characteristic is 0 and the power characteristic is 1, it operates in buck power transmission mode and selects the second working mode. When the gain characteristic is 2 and the power characteristic is 3, it operates in the boost low power transmission condition and selects the third working mode. When the gain characteristic is 2 and the power characteristic is 4, it operates in the boost power transmission mode and selects the fourth working mode. When the gain characteristic is 1 or the power characteristic is 2, it operates in the voltage gain of 1 or high power transmission condition, and selects the fifth working mode.

8. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 1, characterized in that: For different operating modes, different first phase shift angles, second phase shift angles, and third phase shift angles are set, specifically as follows: For the first operating mode, the first shift ratio is twice the voltage gain multiplied by the reference shift ratio and then divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second shift ratio, it is twice the reference shift ratio divided by 1, which is the difference between the voltage gain and the first shift ratio. For the second operating mode, the first shift ratio is: multiply the difference between 1 and the voltage gain by the difference between 1 and 2 times the reference external shift ratio, then divide by the voltage gain, and subtract the division result from 1 to obtain the first shift ratio. The second shift ratio is 1. For the third operating mode, the first shift ratio is twice the voltage gain divided by the difference between the voltage gain and 1, and then multiplied by the reference shift ratio; the second shift ratio is twice the difference between the voltage gain and 1, and then multiplied by the reference shift ratio. For the fourth operating mode, the first shift ratio is 1; the second shift ratio is the difference between 2 times the voltage gain and 1, multiplied by the reference external shift ratio, plus 2, and then minus the voltage gain. For the first, second, third, and fourth working modes, the third shift ratio is equal to the reference shift ratio plus 0.5 times the corresponding first shift ratio minus 0.5 times the corresponding second shift ratio; For the fifth operating mode, the first shift ratio and the second shift ratio are both 0, and the third shift ratio is the difference between the reference external shift ratio and 0.5, multiplied by 0.5 and the difference between 0.5 and the boundary power constant, then divided by 0.5, plus the boundary power constant; the boundary power constant is a set value. For all operating modes, the phase angles of the first, second, third, fourth, and fifth shifts are equal to the ratio of the first, second, third, fourth, and fifth shifts multiplied by the first set period, respectively.

9. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 8, characterized in that: The selection of PS control mode or cyclic rolling parallel phase shift mode converts the corresponding first phase shift angle, second phase shift angle, and third phase shift angle into the modulation wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter. Specifically: Four source drive signals are generated. All four source drive signals are PWM wave signals with the same period (first set period), equal amplitude, and duty cycle of 0.

5. The second source drive signal lags the first source drive signal by a first phase shift angle; the fourth source drive signal lags the first source drive signal by a second phase shift angle; and the third source drive signal lags the first source drive signal by a third phase shift angle. When in the fifth working mode, select the cyclic rolling parallel phase shift control mode; otherwise, select the PS control mode. Based on the selected cyclic rolling parallel phase-shifting control mode or PS control mode, the four source drive signals are processed with different set time delays to generate the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier. In a three-phase full-bridge inverter and a three-phase full-bridge rectifier, the drive signal of the upper output switch of each bridge arm is equal to the corresponding bridge arm drive signal, and the drive signal of the lower output switch of each bridge arm is equal to the inverted signal of the corresponding bridge arm drive signal.

10. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 9, characterized in that: If the PS control mode is selected, the three bridge arm drive signals of the three-phase full-bridge inverter are G1, G1 with a delay of 120° and G1 with a delay of 240° respectively. The three bridge arm drive signals of the three-phase full-bridge rectifier are G3, G3 with a delay of 120°, and G3 with a delay of 240°. G1 and G3 are the first and third source drive signals, respectively.

11. The wide-voltage multi-mode control method for a three-phase dual active bridge DC-DC converter according to claim 9, characterized in that: If the cyclic rolling parallel phase-shifting control mode is selected, the period of the three arm drive signals of the three-phase full-bridge inverter and the three arm drive signals of the three-phase full-bridge rectifier becomes six times the first set period; each first set period is one mode. First mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G2 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G4 respectively. Second mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G3, G3, and G4. Third mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G2; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G4. Fourth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G1, and G1 respectively; the three arm drive signals of the three-phase full-bridge rectifier are G4, G3, and G3 respectively. Fifth mode: The three arm drive signals of the three-phase full-bridge inverter are G2, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G4, G4, and G3. Sixth mode: The three arm drive signals of the three-phase full-bridge inverter are G1, G2, and G1; the three arm drive signals of the three-phase full-bridge rectifier are G3, G4, and G3. G1, G2, G3, and G4 are the first source drive signal, the second source drive signal, the third source drive signal, and the fourth source drive signal, respectively.

12. A three-phase dual active bridge DC-DC converter wide-voltage multi-mode control system based on the method of any one of claims 1-11, comprising a PI controller, a multi-objective joint control unit, a dual-mode reconfigurable modulated wave generator, and a two-dimensional adaptive dynamic threshold condition identification unit, characterized in that: PI controller: used to determine the output reference voltage U ref and output sampling voltage U 2_m The difference is compared to the reference value shifted outward; Two-dimensional adaptive dynamic threshold condition identification unit: used to identify the transformer turns ratio and input sampling voltage U. 1_m and output sampling voltage U 2_m Calculate the voltage gain; output the reference voltage U ref and output sampling voltage U 2_m The difference accounts for the output reference voltage U ref The gain adaptive factor and power adaptive factor are calculated by inputting two preset nonlinear functions respectively. The gain characteristic is calculated based on the relationship between voltage gain and gain adaptation factor, and the power characteristic is calculated based on the relationship between voltage gain, power adaptation factor and reference shift ratio. The current operating condition is determined based on the gain characteristic, power characteristic and preset operating condition judgment conditions, and different operating modes are set for different operating conditions. Multi-objective joint control unit: used to calculate the first phase shift angle, second phase shift angle and third phase shift angle under different operating modes; Dual-mode reconfigurable modulated wave generator: used to select either PS control mode or cyclic rolling parallel phase shift mode according to the operating mode, and convert the corresponding first phase shift angle, second phase shift angle and third phase shift angle into the modulated wave corresponding to the output switching transistor of the three-phase dual active bridge DC converter.

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