Fast smooth switching control method of LLC resonant converter and LLC resonant converter

Abstract

This invention relates to the field of electronic power conversion technology, specifically to a fast and smooth switching control method for an LLC resonant converter and an LLC resonant converter. The switching control method includes: receiving a mode switching command; detecting the dead time of the switching transistors of the full-bridge inverter in the primary circuit; within the detected dead time window, simultaneously changing the conduction state of the reconfiguration switching unit to switch between the series mode and the parallel mode of the secondary winding; after the switching is completed, controlling the switching transistors of the full-bridge inverter in the primary circuit to drive them from a preset highest switching frequency to the target frequency at a preset deceleration rate to perform a soft start; during the soft start process, detecting the current or output voltage flowing through the resonant cavity of the LLC resonant converter; when the detected value exceeds a preset threshold, limiting the output voltage. This invention maintains continuous power supply to the primary side during the switching process, achieving a fast and smooth transition without interrupting charging.

Description

Technical Field

[0001] This invention relates to the field of electronic power conversion technology, specifically to a fast and smooth switching control method for an LLC resonant converter and an LLC resonant converter. Background Technology

[0002] With the increasing popularity of electric vehicles, charging pile modules need to adapt to the charging requirements of different battery voltages (such as 200V-1000V). LLC resonant converters are widely used due to their high efficiency and low EMI, but their output voltage range is limited. When a wide gain output is required, the switching frequency must be far from the resonant frequency, which leads to increased circulating current, decreased efficiency, and even the inability to achieve soft switching.

[0003] To extend the output voltage range of LLC converters, existing technologies mainly employ the following solutions: Two-stage structure: front-end PFC plus back-end Buck / Boost, but this increases the number of components and cost, and reduces overall efficiency.

[0004] Variable frequency + variable duty cycle control: The gain range is extended by variable frequency and phase shift control, but the effect is limited and soft switching is difficult to guarantee over a wide range.

[0005] Secondary winding switching: By changing the connection method of the transformer's secondary winding through switching devices, the equivalent turns ratio is changed, thereby doubling the voltage range. This scheme is considered one of the most promising wide-range solutions because it can significantly expand the voltage range with only a few additional switches.

[0006] However, existing secondary winding switching schemes still have the following common problems: During the switching process, the primary-side switching transistor needs to be stopped, which causes the output voltage to drop and charging to be interrupted, failing to meet the strict requirement of "uninterrupted power supply" during dynamic voltage regulation of electric vehicles; the switching speed is slow (usually requiring hundreds of milliseconds, especially when using relays), which cannot meet the requirements for rapid dynamic response; voltage and current overshoots are prone to occur during the switching process, affecting the reliability of the switching transistor and rectifier devices; the lack of coordinated control between the primary and secondary sides before and after the switching results in excessive output voltage fluctuations, requiring a long time to stabilize again.

[0007] Therefore, there is an urgent need for a converter topology and its control method that can achieve wide voltage range output by switching the secondary winding, while maintaining continuous power supply to the primary side during the switching process, achieving fast and smooth transition, and uninterrupted charging. Summary of the Invention

[0008] The purpose of this invention is to overcome the shortcomings of the prior art and provide a fast and smooth switching control method and an LLC resonant converter, which can achieve a wide voltage range output by switching the secondary winding, and can maintain continuous power supply to the primary side during the switching process, achieving fast and smooth transition and uninterrupted charging.

[0009] To achieve the above objectives / to solve the above technical problems, the present invention is implemented using the following technical solution: In a first aspect, the present invention provides a fast and smooth switching control method for an LLC resonant converter. The LLC resonant converter includes: a primary circuit, a transformer, a reconfiguration switching unit, and a full-bridge rectifier. The transformer includes two secondary windings. The number of full-bridge rectifiers is the same as the number of secondary windings. Each secondary winding is connected to one full-bridge rectifier. The DC output terminals of the full-bridge rectifiers are connected in parallel to the output bus. The same-name terminals and different-name terminals of the two secondary windings are connected through the reconfiguration switching unit. The switching control method includes the following steps: Receive mode switching command; The dead time of the full-bridge inverter switching transistor in the primary circuit is detected. Within the detected dead time window, the conduction state of the reconfiguration switching unit is changed simultaneously to switch between the series mode and the parallel mode of the secondary winding. After the switching is completed, the switching transistors of the full-bridge inverter in the primary circuit are controlled to start from the preset highest switching frequency and drive towards the target frequency at a preset deceleration rate to perform soft start. During the soft-start process, the current or output voltage flowing through the resonant cavity of the LLC resonant converter is detected. When the detected value exceeds a preset threshold, at least one of the following operations is performed to limit the output voltage: pausing the decrease of the switching frequency; increasing the switching frequency; or skipping at least one drive pulse.

[0010] In conjunction with the first aspect, optionally, the preset maximum switching frequency is more than three times the operating frequency before switching.

[0011] In conjunction with the first aspect, optionally, the preset deceleration rate is a fixed frequency step size or a fixed frequency change slope.

[0012] In conjunction with the first aspect, optionally, the duration of the soft start is 1ms to 5ms.

[0013] In conjunction with the first aspect, optionally, after receiving the mode switching instruction step, the method may further include: detecting the current output voltage; if the output voltage is within a preset overlap region, then disabling the execution of the mode switching instruction.

[0014] In conjunction with the first aspect, optionally, the output voltage overlap region is 495V-505V.

[0015] In conjunction with the first aspect, optionally, the soft-start process includes: setting the switching frequency to the highest switching frequency at the beginning of the first complete switching cycle after switching; and then reducing the switching frequency by a fixed step every predetermined time interval until the target frequency is reached, wherein the predetermined time interval is 20us to 200us.

[0016] In a second aspect, the present invention provides an LLC resonant converter, employing the fast and smooth switching control method for the LLC resonant converter as described in any one of the first aspects; the LLC resonant converter includes: The primary-side circuit includes an input DC bus capacitor, a full-bridge inverter, and a resonant cavity connected in sequence. The resonant cavity includes a resonant inductor and a resonant capacitor connected in series. A transformer has a first secondary winding and a second secondary winding with equal turns; The first full-bridge rectifier has its AC input terminal connected to both ends of the first secondary winding, and its DC output terminal connected to the output bus. The second full-bridge rectifier has its AC input terminal connected to both ends of the second secondary winding, and its DC output terminal with positive and negative terminals connected to the same output bus. At least one filter capacitor connected in parallel between the positive and negative terminals of the output bus; The reconfiguration switch unit includes a first reconfiguration switch and a second reconfiguration switch; one end of the first reconfiguration switch is connected to the opposite-named end of the first secondary winding, the other end of the first reconfiguration switch is connected to one end of the second reconfiguration switch, and the other end of the second reconfiguration switch is connected to the same-named end of the second secondary winding.

[0017] In conjunction with the second aspect, optionally, the reconfiguration switch is a MOSFET or an IGBT.

[0018] In conjunction with the second aspect, optionally, a discharge circuit is also included, wherein the discharge circuit is composed of a discharge switch and a discharge resistor connected in series, and the discharge circuit is connected in parallel between the positive and negative terminals of the output bus.

[0019] Compared with the prior art, the beneficial effects achieved by the present invention are as follows: This invention detects the dead time of the primary-side full-bridge inverter switching transistors and completes the switching of the secondary-side reconfiguration switching unit within this extremely short time window. During this process, the primary-side PWM counter does not stop and the drive signal is continuously generated, ensuring that the current flowing through the resonant cavity and the output bus is uninterrupted. It is specifically applied to scenarios such as electric vehicle charging where instantaneous power outages are not allowed. This invention employs a control strategy that combines dead-zone switching with high-frequency soft-start after switching. First, switching occurs within the dead zone, utilizing the natural condition that the primary and secondary currents / voltages are zero to achieve zero-voltage turn-on and zero-current turn-off, fundamentally eliminating switching losses and high-frequency voltage spikes at the moment of switching. After switching, this invention uses a preset switching frequency for soft-start, making the resonant cavity exhibit extremely high impedance and limiting instantaneous changes in current and voltage. Subsequently, it is driven with a preset deceleration rate, causing the output voltage to exhibit a smooth rise curve, effectively suppressing large jumps caused by changes in the equivalent turns ratio, and achieving a smooth transition without overshoot or ringing. Attached Figure Description

[0020] Figure 1 This is a circuit topology diagram of an LLC resonant converter according to an embodiment of the present invention; Figure 2 This is a timing waveform diagram of the switching from parallel mode to series mode in an embodiment of the present invention; Figure 3 This is a timing waveform diagram of the dead time window in an embodiment of the present invention. Detailed Implementation

[0021] The technical solution of the present invention will be described in detail below with reference to the accompanying drawings and specific embodiments. It should be understood that the embodiments and specific features in the embodiments are detailed descriptions of the technical solution of the present application, rather than limitations thereof. In the absence of conflict, the embodiments and technical features in the embodiments can be combined with each other.

[0022] In the description of this invention, the terms "first" and "second" are used only to distinguish technical features and should not be construed as indicating or implying relative importance, or implicitly indicating the number of technical features indicated, or implicitly indicating the order of the technical features indicated.

[0023] In the description of this invention, references to terms such as "one embodiment," "some embodiments," "illustrative embodiment," "example," "specific example," or "some examples," etc., indicate that a specific feature, structure, material, or characteristic described in connection with that embodiment or example is included in at least one embodiment or example of the invention. In this specification, illustrative expressions of the above terms do not necessarily refer to the same embodiment or example. Furthermore, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or more embodiments or examples.

[0024] Example 1:

[0025] This invention provides an LLC resonant converter, the LLC resonant converter comprising: The primary circuit includes an input DC bus capacitor, a full-bridge inverter, and a resonant cavity connected in sequence. The resonant cavity includes a resonant inductor and a resonant capacitor connected in series.

[0026] A transformer has a first secondary winding and a second secondary winding with equal turns.

[0027] The first full-bridge rectifier has its AC input terminal connected to both ends of the first secondary winding, and its DC output terminals connected to the output bus.

[0028] The second full-bridge rectifier has its AC input terminal connected to both ends of the second secondary winding, and its DC output terminals connected to the same output bus.

[0029] At least one filter capacitor is connected in parallel between the positive and negative terminals of the output bus.

[0030] The reconfiguration switch unit includes a first reconfiguration switch and a second reconfiguration switch; one end of the first reconfiguration switch is connected to the opposite-named end of the first secondary winding, the other end of the first reconfiguration switch is connected to one end of the second reconfiguration switch, and the other end of the second reconfiguration switch is connected to the same-named end of the second secondary winding.

[0031] The reconfigurable switch is a MOSFET or an IGBT.

[0032] It also includes a discharge circuit, which consists of a discharge switch and a discharge resistor connected in series, and the discharge circuit is connected in parallel between the positive and negative terminals of the output bus.

[0033] like Figure 1 The diagram shows a schematic embodiment of an LLC resonant converter circuit topology. The primary-side circuit includes an input DC bus capacitor C1. The full-bridge inverter includes MOSFETs Q1, Q2, Q3, and Q4. The positive terminal of the input DC bus capacitor C1 is connected to the drains of MOSFETs Q1 and Q2, and the negative terminal is connected to the sources of MOSFETs Q3 and Q4. The source of MOSFET Q1 is connected to the drain of MOSFET Q3, and the source of MOSFET Q2 is connected to the drain of MOSFET Q4. The resonant cavity includes a resonant inductor LR1 and a resonant capacitor CR1. One end of the resonant inductor LR1 is connected to one end of the resonant capacitor CR1, and the other end of the resonant inductor LR1 is connected to the source of MOSFET Q2. The other end of the resonant capacitor CR1 is connected to one end of the primary winding T1P of the transformer, and the other end of the primary winding T1P is connected to the source of MOSFET Q1.

[0034] Transformer T1 has a first secondary winding T1S1 and a second secondary winding T1S2 with equal number of turns.

[0035] The first full-bridge rectifier includes diodes D11, D12, D13, and D14. The cathode of diode D11 is connected to the cathode of diode D12 and to the positive terminal of the output bus. The anode of diode D13 is connected to the anode of diode D14 and to the negative terminal of the output bus. The cathode of diode D13 is connected to the anode of diode D11. The cathode of diode D14 is connected to the anode of diode D12. The same-name terminal of the secondary winding T1S1 is connected to the anode of diode D11, and the opposite-name terminal of the secondary winding T1S1 is connected to the anode of diode D12.

[0036] Similarly, the second full-bridge rectifier includes diodes D21, D22, D23, and D24. The cathode of diode D21 is connected to the cathode of diode D22 and to the positive terminal of the output bus. The anode of diode D23 is connected to the anode of diode D24 and to the negative terminal of the output bus. The cathode of diode D23 is connected to the anode of diode D21. The cathode of diode D24 is connected to the anode of diode D22. The same-name terminal of the secondary winding T1S2 is connected to the anode of diode D22, and the opposite-name terminal of the secondary winding T1S2 is connected to the anode of diode D21.

[0037] Filter capacitors C2, C3, and C4 are connected in parallel between the positive and negative terminals of the output bus, respectively.

[0038] The reconfiguration switch unit includes a first reconfiguration switch Q5 and a second reconfiguration switch Q6. The drain of the first reconfiguration switch Q5 is connected to the opposite-named terminal of the first secondary winding T1S1, the source of the first reconfiguration switch Q5 is connected to the source of the first reconfiguration switch Q6, and the drain of the first reconfiguration switch Q6 is connected to the same-named terminal of the second secondary winding T1S2.

[0039] In this embodiment Figure 1 The discharge circuit is not shown in the diagram. The discharge circuit is existing technology and will not be described in detail here.

[0040] like Figure 1As shown, MOSFETs Q5 and Q6 are connected between the opposite-named terminal of the first secondary winding T1S1 and the same-named terminal of the second secondary winding T1S2. The same-named terminal of the first secondary winding T1S1 is connected to the common point of diodes D11 and D13, the same-named terminal of the second secondary winding T1S2 is connected to the common point of diodes D12 and D14, the same-named terminal of the second secondary winding T1S2 is connected to the common point of diodes D22 and D24, and the opposite-named terminal of the second secondary winding T1S2 is connected to the common point of diodes D21 and D23. This connection method, combined with a specific rectifier bridge connection, ensures that in series mode, current flows only through the two diagonally opposite diodes (D11, D13, D21, D23) in each full bridge. In this case, the voltage drop path of each diode and the series path of the windings are superimposed in the same direction, eliminating redundant circulating current paths and minimizing conduction losses.

[0041] Taking the series mode as an example, with both MOSFETs Q5 and Q6 turned on, the current path for the first half-cycle is: the same-name terminal of the first secondary winding T1S1 — diode D11 — positive terminal of the output bus — negative terminal of the output bus — diode D23 — the opposite-name terminal of the second secondary winding T1S2 — the same-name terminal of the second secondary winding T1S2 — MOSFET Q6 — MOSFET Q5 — the opposite-name terminal of the first secondary winding T1S1 — the same-name terminal of the first secondary winding T1S1, completing the cycle; the current path for the second half-cycle is: the opposite-name terminal of the second secondary winding T1S2 — diode D21 — positive terminal of the output bus — negative terminal of the output bus — diode D13 — inside the first secondary winding T1S1 from the same-name terminal to the opposite-name terminal — MOSFET Q5 — MOSFET Q6 — the same-name terminal of the second secondary winding T1S2, completing the cycle.

[0042] In this application, MOSFETs Q5 and Q6 are connected in reverse series between the opposite terminal of one winding and the same terminal of the other winding. This structure constitutes a bidirectional controllable switch that can withstand the positive and negative half-cycles of the secondary AC voltage, while preventing current from flowing through the body diode when the reconfiguration switching unit is not turned on.

[0043] In parallel mode, the two secondary windings independently supply power to the output bus through their respective rectifier bridges. In this application, after MOSFETs Q5 and Q6 are turned off, the only electrical connection point between the two windings is completely severed, eliminating any circulating current path and resulting in higher isolation after turn-off.

[0044] Example 2:

[0045] This invention provides a fast and smooth switching control method for an LLC resonant converter, wherein the LLC resonant converter includes: a primary circuit, a transformer, a reconfiguration switching unit, and a full-bridge rectifier. The transformer includes two secondary windings, and the number of full-bridge rectifiers is the same as the number of secondary windings. Each secondary winding is connected to one full-bridge rectifier. The DC output terminals of the full-bridge rectifiers are connected in parallel to the output bus. The same-name terminals and different-name terminals of the two secondary windings are connected through the reconfiguration switching unit. In some specific examples, the circuit topology of the LLC resonant converter is as follows: Figure 1 As shown.

[0046] The switching control method includes the following steps: Receive mode switching command; The dead time of the full-bridge inverter switching transistor in the primary circuit is detected. Within the detected dead time window, the conduction state of the reconfiguration switching unit is changed simultaneously to switch between the series mode and the parallel mode of the secondary winding. After the switching is completed, the switching transistors of the full-bridge inverter in the primary circuit are controlled to start from the preset highest switching frequency and drive towards the target frequency at a preset deceleration rate to perform soft start. During the soft-start process, the current or output voltage flowing through the resonant cavity of the LLC resonant converter is detected. When the detected value exceeds a preset threshold, at least one of the following operations is performed to limit the output voltage: pausing the decrease of the switching frequency; increasing the switching frequency; or skipping at least one drive pulse.

[0047] Figure 2 The detailed timing waveform diagrams for the switch from parallel mode to series mode in this embodiment are shown. For example... Figure 3 As shown, the drive signals of the primary-side switches Q1~Q4 enter a low level during the dead time from t1 to t2; within this low-level dead time window, the drive signals of the secondary-side reconfiguration switches Q5 and Q6 are simultaneously pulled high at time t1, completing the series reconfiguration of the secondary winding. After the switching is completed, as shown... Figure 2 As shown, the drive signals of the primary-side switching transistors Q1~Q4 are restored in the normal timing sequence during the period from t1 to t3. The output bus voltage Vout performs soft-start ramp-up, and its ramp-up slope is controlled by the drive strategy of decreasing from the preset highest switching frequency to the target frequency, thereby realizing a smooth transition and uninterrupted power supply during the switching process.

[0048] Specifically, the receiving mode switching command includes the controller receiving the target output voltage value from the charging pile main controller via the communication interface. When the target output voltage value does not match the voltage range corresponding to the current mode, a mode switching command is generated, for example: The current mode is parallel, which is applicable to a voltage range of 200V-500V. If the target value is 700V, it is necessary to switch to series mode. The current mode is series connection, which is applicable to a voltage range of 500V-1000V. If the target value is 400V, it is necessary to switch to parallel connection mode.

[0049] To avoid frequent mode switching, a voltage overlap region is set. In one illustrative embodiment, the voltage overlap region is 495V to 505V. If the current output voltage is within the overlap region, the switching command is ignored, and a report is sent to the host computer stating "The output voltage is in the hysteresis region, so we will not switch for now".

[0050] In some examples, to further smooth the transition, a pre-adjustment step can be performed before switching: for a parallel-to-series switch, the controller adjusts the output voltage to a value slightly higher than the maximum value set for the voltage overlap region, such as 510V, through frequency control, so that the operating point of the resonant cavity is slightly higher than the overlap region, so that the soft start after switching is smoother; similarly, for a series-to-parallel switch, the output voltage can be adjusted to a value slightly lower than the minimum value set for the voltage overlap region, such as 490V; this step is not necessary, but it can optimize the switching effect.

[0051] The detection of the dead time of the full-bridge inverter switching transistors in the primary-side circuit specifically includes: the controller needs to accurately capture the dead time of the primary-side full-bridge inverter switching transistors. During this time period, since the two switches are connected in series between the opposite-named and same-named terminals, the voltage across MOSFET Q5 is equal to the voltage across the opposite-named terminal of the first secondary winding T1S1, and the voltage across MOSFET Q6 is equal to the voltage across the same-named terminal of the second secondary winding T1S2. Since the voltages of both the first secondary winding T1S1 and the second secondary winding T1S2 are zero and in phase, the source-drain voltages of both MOSFETs Q5 and Q6 are zero, achieving zero-voltage turn-on. Simultaneously, the winding current is zero, achieving zero-current turn-off. This "double-zero" switching eliminates all switching losses and spikes. The dead time detection method can employ a hardware synchronization method based on a PWM counter or a software judgment method based on zero-crossing voltage detection. Both methods are existing technologies and will not be elaborated upon here. This embodiment uses a hardware synchronization method based on a PWM counter, which is accurate and does not require additional external components.

[0052] Within the detected dead time window, simultaneously changing the conduction state of the reconfiguration switch unit specifically includes: within the dead time window, such as... Figure 1 As shown, for the switch from parallel to series connection, the gate drive signals of MOSFETs Q5 and Q6 are switched from low level to high level; for the switch from series to parallel connection, the gate drive signals of MOSFETs Q5 and Q6 are switched from high level to low level; the two switches must operate strictly simultaneously.

[0053] by Figure 1The topology is explained as follows: during the dead time, primary-side switches Q1 to Q4 are all in the off state; therefore, the switching of the reconfigured switches is not affected by the primary-side current. After the switching is completed, the primary-side PWM resumes according to the normal timing sequence, that is, after the dead time ends, the next switching cycle begins.

[0054] It should be noted that this invention does not stop the generation of the primary-side PWM, that is, the PWM counter does not stop and the enable signal is not canceled. It only uses the inherent dead time to complete the switching, which is fundamentally different from the prior art of "completely stopping the primary-side operation and then restarting".

[0055] After the switching operation is completed, the connection method of the secondary winding has changed, that is, from parallel to series or from series to parallel. Due to the change in turns ratio, if the original switching frequency is kept unchanged, the output voltage will jump significantly (for example, from 500V to 1000V). To suppress the jump, the controller immediately executes the following soft-start procedure: The maximum switching frequency is preset to 3 to 5 times the operating frequency before switching to ensure that the gain is much less than 1 in the early stage of soft start, thereby limiting the rise of output voltage. The target frequency is calculated based on the target output voltage value and LLC gain curve in the new mode. The calculation method is existing technology and will not be elaborated on here. When driving towards the target frequency at a preset deceleration rate, a step-decreasing method is used, that is, the switching frequency is reduced by a fixed step at regular intervals until the target frequency is reached. For example: the preset maximum switching frequency is 500kHz, the target frequency is 150kHz, the soft start time is 3ms, and the preset duration is 50us, then the number of steps is 3ms / 50us=60, and the fixed step size is (500kHz-150kHz) / 60=5.833kHz / step; In another embodiment, the preset deceleration rate can also be a fixed frequency change slope. The controller uses a linear frequency sweep to continuously decrease the switching frequency over time at a fixed rate until the target frequency is reached. For example, the controller calculates the difference between the target frequency and the highest switching frequency, divides it by a preset total soft-start duration, and obtains a fixed slope value. The controller's PWM module calculates and updates the frequency register value based on this slope in each control cycle, thus causing the frequency to decrease along a continuous, smooth slope. Compared to a fixed-step, stepped decrease, a linear frequency sweep makes the change in resonant cavity gain smoother, further optimizing the soft-start waveform of the output voltage.

[0056] At the start of the first complete switching cycle after switching, the controller immediately sets the period register value of the PWM module to the value corresponding to the highest switching frequency; at this time, due to the extremely high frequency and the large impedance of the resonant cavity, the output voltage hardly rises.

[0057] Subsequently, at predetermined time intervals, the controller reduces the frequency by a fixed step size, gradually approaching the target frequency. During the reduction process, the controller continuously monitors the output bus voltage and the primary resonant current. The output bus voltage can be obtained through resistor voltage division sampling, and the primary resonant current can be obtained through resonant capacitor series current transformer or Hall sensor sampling. The sampling methods are all existing technologies and will not be elaborated on here.

[0058] If the primary-side resonant current exceeds a preset threshold during soft-start, for example, 1.2 times the rated peak current, the controller will perform any one or a combination of the following operations: Pause switching frequency reduction: Maintain the current frequency and prevent it from decreasing until the current drops below 0.9 times the preset threshold.

[0059] Increase the switching frequency: Jump the frequency up by one step, for example, twice the original set step, to quickly reduce the gain and suppress the current.

[0060] Skip drive pulses: In extreme overcurrent conditions, the controller can temporarily block 1 to 3 PWM pulses, causing the resonant cavity current to decay rapidly. This method is activated only when the current exceeds 1.5 times the preset threshold, serving as hard protection.

[0061] At the same time, if the output bus voltage rises too quickly, the controller will also actively increase the frequency to limit voltage overshoot.

[0062] The soft-start process ends when the frequency decreases to within ±2kHz of the target frequency and the output voltage reaches ±1% of the target output voltage value. The controller then returns control to the normal voltage closed-loop PI regulator, which outputs a frequency correction based on the voltage error. At this point, the entire switching process is complete.

[0063] The method in this embodiment is executed by a digital controller (such as a TMS320F28335 DSP or an STM32G4 series MCU), which integrates a PWM module, an ADC sampling module, a comparator, and a logic control unit.

[0064] The above description is only a preferred embodiment of the present invention. It should be noted that for those skilled in the art, several improvements and modifications can be made without departing from the technical principles of the present invention, and these improvements and modifications should also be considered within the scope of protection of the present invention.

Claims

1. A fast and smooth switching control method for an LLC resonant converter, characterized in that, The LLC resonant converter includes: a primary circuit, a transformer, a reconfiguration switching unit, and a full-bridge rectifier. The transformer includes two secondary windings. The number of full-bridge rectifiers is the same as the number of secondary windings. Each secondary winding is connected to one full-bridge rectifier. The DC output terminals of the full-bridge rectifiers are connected in parallel to the output bus. The same-name and different-name terminals of the two secondary windings are connected through the reconfiguration switching unit. The switching control method includes the following steps: Receive mode switching command; The dead time of the full-bridge inverter switching transistor in the primary circuit is detected. Within the detected dead time window, the conduction state of the reconfiguration switching unit is changed simultaneously to switch between the series mode and the parallel mode of the secondary winding. After the switching is completed, the switching transistors of the full-bridge inverter in the primary circuit are controlled to start from the preset highest switching frequency and drive towards the target frequency at a preset deceleration rate to perform soft start. During the soft-start process, the current or output voltage flowing through the resonant cavity of the LLC resonant converter is detected. When the detected value exceeds a preset threshold, at least one of the following operations is performed to limit the output voltage: pausing the decrease of the switching frequency; increasing the switching frequency; or skipping at least one drive pulse.

2. The fast and smooth switching control method for an LLC resonant converter according to claim 1, characterized in that: The preset maximum switching frequency is 3 to 5 times the operating frequency before switching.

3. The fast and smooth switching control method for an LLC resonant converter according to claim 1, characterized in that: The preset deceleration rate is a fixed frequency step size or a fixed frequency change slope.

4. The fast and smooth switching control method for an LLC resonant converter according to claim 1, characterized in that: The duration of the soft start is 1ms to 5ms.

5. The fast and smooth switching control method for an LLC resonant converter according to claim 1, characterized in that: The process after receiving the mode switching command step also includes: detecting the current output voltage; if the output voltage is within a preset overlap area, then the mode switching command is disabled.

6. The fast and smooth switching control method for an LLC resonant converter according to claim 5, characterized in that: The output voltage overlap range is 495V-505V.

7. The fast and smooth switching control method for an LLC resonant converter according to claim 1, characterized in that: The soft-start process includes: at the beginning of the first complete switching cycle after switching, setting the switching frequency to the highest switching frequency; and then reducing the switching frequency by a fixed step every predetermined time interval until the target frequency is reached.

8. An LLC resonant converter, characterized in that, A fast and smooth switching control method for implementing the LLC resonant converter as described in any one of claims 1 to 7; the LLC resonant converter includes: The primary-side circuit includes an input DC bus capacitor, a full-bridge inverter, and a resonant cavity connected in sequence. The resonant cavity includes a resonant inductor and a resonant capacitor connected in series. A transformer has a first secondary winding and a second secondary winding with equal turns; The first full-bridge rectifier has its AC input terminal connected to both ends of the first secondary winding, and its DC output terminal connected to the output bus. The second full-bridge rectifier has its AC input terminal connected to both ends of the second secondary winding, and its DC output terminal with positive and negative terminals connected to the same output bus. At least one filter capacitor connected in parallel between the positive and negative terminals of the output bus; The reconfiguration switch unit includes a first reconfiguration switch and a second reconfiguration switch; one end of the first reconfiguration switch is connected to the opposite-named end of the first secondary winding, the other end of the first reconfiguration switch is connected to one end of the second reconfiguration switch, and the other end of the second reconfiguration switch is connected to the same-named end of the second secondary winding.

9. The LLC resonant converter according to claim 8, characterized in that: The reconfiguration switch is a MOSFET or an IGBT.

10. The LLC resonant converter according to claim 8, characterized in that: It also includes a discharge circuit, which consists of a discharge switch and a discharge resistor connected in series, and the discharge circuit is connected in parallel between the positive and negative terminals of the output bus.

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