Method of modulating and demodulating wireless signals for high mobility communications, and apparatus implementing the method

Abstract

Radio transmission signals are modulated by converting a serial sequence of symbols in the time domain into a parallel symbol stream, converting the parallel symbol stream to the frequency domain, mapping the resulting frequency domain signal onto subcarriers, converting the mapped signal back to the time domain, and converting the parallel time domain signal back into a serial signal. This serial signal undergoes chirping and is transmitted to the receiver after adding a cyclic prefix (CP). The receiver removes the CP from the signal and, after dechirping and unprefixing, converts the signal back into a parallel signal. The parallel signal is converted back to the frequency domain for channel estimation, equalization, and subcarrier mapping of detected symbols. The mapped symbols are transformed back to the time domain before being converted into a serial stream of symbols.

Description

Technical Field

[0001] This invention relates to mobile wireless communication, and more particularly to wireless communication based on 6th generation (6G) standards and above.

[0002] symbol

[0003] Scalar values ​​are represented in this paper by italicized lowercase or uppercase letters, such as x or N, while vectors and matrices in the frequency domain are represented by bold lowercase and uppercase letters, such as x and X, respectively. The transpose and Hermitian transpose of a matrix are represented by superscript letters (·). T and(·) H The expression `diag{a}` represents a square diagonal matrix with elements `a` in its main diagonal. It is the Kronecker product. It is a tracing operation, and Indicates the desired action. Background Technology

[0004] Sometimes referred to as transmission waveform or simply waveform modulation, it has been considered an essential component of every generation of communications, enabling increased capacity and data rates, improved spectral and power efficiency, new applications, and more. The 6G waveform candidates currently under discussion can be categorized into three types: multi-carrier, single-carrier, and multi-carrier chirped.

[0005] Mobile wireless communications based on current 4G (4G, LTE), 5G, and future 6G communication standards typically use OFDM as a common modulation technique for downlink connections (i.e., connections from the base station (BS) to the user equipment (UE)). 6G specifically anticipates the use of different types of OFDM or entirely different modulation techniques, where different modulation techniques may be used for both uplink (i.e., from the mobile UE to the BS) and downlink communication connections. Among the possible downlink modulation techniques in 6G, analogous radio frequency division multiplexing (AFDM) and orthogonal time-frequency space (OTFS) are promising candidates, while Discrete Fourier transform extended OFDM (DFT-s-OFDM) is a promising modulation technique for uplink communication connections due to its low peak-to-average power ratio (PAPR). However, currently envisioned modulation techniques may not be suitable for the high mobility communication in 6G wireless networks, which will be an important use case.

[0006] Multicarrier waveforms include Orthogonal Time-Frequency Space (OTFS), Discrete Zak Transform-based OTFS, Orthogonal Delayed Doppler Frequency Division Multiplexing (ODDM), Interleaved Frequency Division Multiplexing (IFDM), and Constant Envelope OFDM (CE-OFDM). However, similar to OFDM, OTFS, ODDM, and multicarrier-based IFDM exhibit high PAPR, making them unsuitable for low-power devices. While FM-OFDM can reduce PAPR by modulating the OFDM signal with a frequency modulator, its spectral efficiency is low due to a large number of ineffective subcarriers. Moreover, the performance advantage of FM-OFDM over OTFS has not been observed. CE-OFDM can also achieve low PAPR by modulating the OFDM signal with a phase modulator. However, the performance of CE-OFDM is degraded in high-mobility communications.

[0007] Single-carrier categories include Orthogonal Time Multiplexing (OTSM), however, the PAPR of OTSM is as high as that of OTFS.

[0008] Orthogonal chirped division multiplexing (OCDM) and analogous radio frequency division multiplexing (AFDM) belong to the multi-carrier chirped category. AFDM is an enhanced version of OCDM catering to high-mobility communications. However, both are based on multiple carriers and exhibit high PAPR (Packet Replication Rate).

[0009] In terms of computational complexity, that is, the number of operations that must be performed to complete the modulation of the symbols to be transmitted, there is a wide variation among the conventional modulation techniques discussed above. Figure 1 This diagram illustrates a comparative analysis of the core operations required to obtain the transmitted signal from the selection of the aforementioned conventional modulation schemes. The core operation of OFDM is the M-point Inverse Fast Fourier Transform (IFFT), with a complexity of Mlog₂M, where M is the number of subcarriers used. Note that the number of subcarriers M increases with the channel bandwidth. The core operation of AFDM also adds the diagonal matrix multiplication upstream of this required M-point IFFT to another diagonal matrix multiplication downstream of the M-point IFFT, resulting in a complexity of Mlog₂M + 2M. OTFS requires the Inverse Symplectic Fast Fourier Transform (ISFFT), and due to the 2D frame structure, it requires M... OTFS Point IFFT results in Mlog2M + Mlog2M OTFS The overall complexity is as follows. The core operations of DFT-s-OFDM include an M-point DFT, followed by subcarrier mapping and an N-point IFFT, with an overall complexity of Mlog₂M + Nlog₂N. Finally, the core operations of cyclic shift chirping with DFT-s-OFDM include all the core operations of DFT-s-OFDM plus an FDSS filter for chirping. The complexity of the filter is not precisely known, but it undoubtedly increases the overall complexity.

[0010] OFDM provides acceptable reliability for wireless communication at the cost of high PAPR, which can lead to distortion and spectral spread when signal power peaks cause the transmitter to operate in a non-linear region. Designing RF amplifiers for high peak values ​​is expensive and reduces the average efficiency of the RF stage. While AFDM and OTFS offer higher reliability than OFDM, the disadvantages of high PAPR still exist. DFT-s-OFDM and cyclic shift chirp with DFT-s-OFDM provide similar reliability to AFDM and OTFS at lower PAPR. However, the latter two, such as AFDM and OTFS, may not provide the ultra-high reliability required in some 6G use cases, such as communications for autonomous driving (AD) and other use cases where message duplication and the resulting delays are undesirable or unacceptable.

[0011] Therefore, it is desirable to provide a method for modulating symbols to be transmitted that exhibits low PAPR, low bit error rate (BER), and ultra-high reliability. Summary of the Invention

[0012] This objective is achieved by the method of modulating symbols representing binary data for transmission over a wireless communication channel as proposed in claim 1, the method of demodulating such modulated symbols as proposed in claim 6, and the transmitter and receiver as proposed in claims 11 and 12, respectively. Computer program products and computer-readable media are provided in claims 13 and 14, respectively. Embodiments and developments are described in the various dependent claims.

[0013] This invention combines the concepts of AFDM and DFT-s-OFDM introduced in the background section of this specification. As will be shown below, chirping is added after the signal processed by DFT is transformed to the time domain to provide the desired properties.

[0014] Figure 2 A schematic block diagram of a chirped DFT-s-OFDM communication system according to the present invention is shown, the communication system including a transmitter 500 and a receiver 600. Figure 2 The diagram also shows the representation of the signal at each stage of processing and in the corresponding domain (time domain or frequency domain) in which the signal exists.

[0015] Transmitter 500 performs two main processing functions, including DFT-s-OFDM modulation in function blocks 510 to 550, and chirping in function block 560. In detail:

[0016] At transmitter 500, the serial sequence of data symbols is first received at serial-to-parallel (S2P) converter 510 and converted into an unmodulated time-domain data vector of length M for parallel data symbols, denoted here as... Parallel data symbols are input to block 520, which, for example, transforms the data symbols from the time domain to their corresponding representation in the frequency domain using an M-point DFT to obtain M data symbols in the frequency domain. The length M of the resulting frequency domain data vector is determined by... Given, among which Let M be the M-point DFT matrix used in this example. Block 530 receives M frequency domain data symbols and fills them with a second number of N – M padding symbols (e.g., zeros), producing a third number of N frequency domain transmit symbols. Also in block 530, the N frequency domain transmit symbols are mapped to corresponding subcarriers according to a specified subcarrier mapping scheme. Generally, although other schemes are conceivable, there are two subcarrier mapping schemes to choose from: a subband scheme and an interleaving scheme. Of the two typical mapping schemes, the subband scheme mapping, which has consecutive subcarriers for data transmission, is more likely to suffer from deep fading and exhibits a high PAPR compared to interleaving mapping. Therefore, unless otherwise stated, the mapping according to the interleaving scheme is considered in the remainder of this specification, where the subcarrier mapping matrix... The size is ,in It is the size of The identity matrix. The integer DFT spreading factor (SF) is defined as... After subcarrier mapping, a data vector of length N is written as... Note that the SF fraction can be selected, and the subcarrier mapping matrix can be defined in other ways, although this comes at the cost of increasing PAPR.

[0017] N frequency-domain transmitted symbols are input to block 540, which, for example, transforms the transmitted signal back to the time domain using an N-point IFFT to obtain a time-domain signal. The N time-domain emission symbols represent, among which This is an N-point FFT matrix. The time-domain transmit signal is input to a parallel-to-serial (P2S) converter 550, which converts the N parallel time-domain transmit symbols into a serial time-domain signal of a first length, which corresponds to the length of the cascaded parallel transmit symbols.

[0018] Note that the time-domain signal after DFT precoding is different from the unmodulated time-domain signal input to this process, such as... Figure 2 The illustrative example provided at the top illustrates this. Here, , and This is an example of usage. This indicates the subcarrier spacing, while This represents bandwidth. t and f represent time and frequency, respectively. DFT precoding causes the bandwidth to change from... arrive It has been expanded by two times, including the allocated subbands shown in different patterns and the patternless white unallocated subbands arranged in between. This bandwidth expansion is also known as DFT expansion. Furthermore, according to the magnitude of the reduction... The duration of each data symbol transmission starts from arrive Halved, and then Then repeat. Let the channel length be denoted as L, and the coherence bandwidth correspond to... Each symbol in the frequency band The frequency band is typically larger than the coherence bandwidth, thus enabling frequency diversity, as will be discussed in more detail below.

[0019] First length Serial time domain signal It is fed into chirp block 560, where Indicates the number of symbols and Indicates the duration of the symbol. Defined as ,in Corresponding to bandwidth, the chirped block 560 also receives signals with a chirped rate. length The corresponding chirping signal The chirped DFT-s-OFDM signal thus obtained is output to block 570, which is configured to insert a cyclic prefix (CP) into the time-domain chirped DFT-s-OFDM signal.

[0020] In this example, the chirped signal is linear, and the chirped rate is set to [value missing]. The indicated frequency is the same as that extended across the full bandwidth. Add one corresponding to each moment. Or subcarrier spacing, as represented by the single-carrier chirped waveform shown between pre-chirped and post-chirped symbol representations. However, the chirped signal can also be nonlinear. As a result, after chirping, data symbols represented by different patterns can "jump" to the unassigned subcarrier band. In other words, both the assigned and unassigned subbands are used for data transmission over time, i.e., the full band, thus providing more observations than unknowns and leading to enhanced noise suppression in the frequency domain. Moreover, it mitigates the effects of channel fading. This band spread is also known as chirped spread spectrum. In the figure, the arrow in the rightmost symbol representation indicates the frequency increase due to chirping. Combined DFT spread spectrum before and after the OFDM signal, and chirped spread spectrum, respectively, produce the name (moniker) for the proposed waveform: chirped DFT-s-OFDM. The chirped DFT-s-OFDM signal is represented as :

[0021] (1)

[0022] in .

[0023] Attached to before transmission The CP preferably has The length of , where The length of the channel sequence. Defined as the first Channel path in the The channel gain at time t, where and .

[0024] Note that, instead of SF=2 used in this example, increasing SF to 4 will cause each symbol to be repeated 4 times in the time domain, resulting in additional noise suppression.

[0025] The modulated signal output from block 570 is then transmitted via antenna 580 over a wireless channel subjected to biselective fading. Biselective fading refers to a wireless communication channel subjected to velocity-dependent Doppler shift or spread caused by a rapidly moving transmitter and / or receiver, as well as rapidly changing multipath reception, resulting in severe time and frequency dispersion. The time and frequency dispersion, caused by the different path lengths and speeds between the transmitter and receiver respectively, each induce signal fading at the receiver, thus giving rise to the name biselective channel fading. Biselective channel fading significantly impairs the performance of the wireless communication system. The communication channel is represented in the diagram by lightning and cloud-like symbols.

[0026] Channel matrix of communication channel Export as follows: Definition The normalized maximum Doppler frequency is relative to the subcarrier spacing, where and These are the carrier frequency, the speed of light, and the subcarrier spacing. The maximum Doppler frequency should be within one subcarrier spacing to produce... . The following formula is given

[0027] (2)

[0028] in and For channel gain and the first The normalized Doppler frequency shift of each path. The time-domain channel matrix is ​​represented as...

[0029] (3)

[0030] in and It is a forward circular shift matrix, that is, each row is a matrix that is circularly right-shifted from the previous row. It is also called a circular matrix, as shown in the following example.

[0031]

[0032] Note that this specification considers a channel model with random Doppler shifts for each channel path. However, the proposed method can be readily extended to a Doppler extended channel model with multiple Doppler shifts for each channel path.

[0033] At receiver 600, the transmitted time-domain signal is received at antenna 610 and provided to block 620, which is configured to remove CP from the signal to produce a prefix-free time-domain signal.

[0034] Length after removing the cycle prefix The received time-domain signal vector is represented as

[0035] (4)

[0036] in It has variance An additive white Gaussian noise vector. Definition In the chirping and After point FFT, the frequency domain received signal vector is

[0037] (5)

[0038] The unprefixed time-domain signal is then input to dechirping block 630, which also receives a copy of the chirped signal used in transmitter 500. Dechirping block 630 inverts the chirping performed in the transmitter and generates a signal carrying a third number of N DFT-s-OFDM modulated transmit symbols transmitted from the transmitter. The dechirped unprefixed received signal is then fed to S2P block 640, which outputs multiple parallel segments, all of which represent the dechirped unprefixed received signal. The parallel segments are provided to block 650, which transforms the parallel segments to the frequency domain, for example, by an N-point FFT. The transformed segments are then provided to block 660, configured for channel estimation (CE) and equalization (EQ), to recover the transmitted frequency-domain symbols. Various conventional OFDM channel estimation and equalization schemes can be employed for this purpose. Any frequency domain symbols that may have been recovered on any unassigned subbands can be discarded, i.e., subbands that were not assigned by the DFT at the transmitter and were filled with padding symbols, as indicated by the arrow pointing from box 670 to empty space. Then, in remapping block 670, the remaining symbols on the assigned subbands are... Each recovered data symbol is remapped to its corresponding subcarrier, and in block 680, for example, by performing... The point-based inverse discrete Fourier transform (IDFT) transforms it into time-domain symbols. Finally, in P2S block 690, the parallel time-domain symbols are converted back into a serial stream of time-domain symbols, which can be output for decoding.

[0039] Note that, for clarity, conventional components of the transmitter or receiver, such as oscillators, mixers, amplifiers, equalizers, etc., are not shown in the diagram.

[0040] Figure 3 It shows from Figure 1 The diagram illustrates a schematic comparison of the core operations required to obtain the transmitted signal using conventional modulation schemes, which are supplemented by the method described above according to the invention. Clearly, the inverse FFT exists in all modulation schemes, and subcarrier mapping exists in all modulation schemes involving the DFT. Similar to AFDM modulation, the proposed method adds a chirp operation after the inverse FFT operation. The similarity or equivalence of the post-IFFT operations is indicated in the figure by the identical shading of the various boxes.

[0041] Chirped DFT-s-OFDM modulation and the resulting waveform presented here utilize frequency diversity and introduce coding gain while maintaining the good PAPR of traditional DFT-s-OFDM. References Figure 4 The illustrative time-frequency diagrams shown, compared to OFDM, AFDM, and DFT-s-OFDM, further illustrate the benefits of the proposed DFT-chirped-s-OFDM modulation and waveform. In the diagrams, This represents the width of the subchannel, centered on the subchannel's carrier frequency and corresponding to the subcarrier spacing. and These represent time and frequency, respectively. (Combined with...) Figure 2 The discussion uses illustrative examples similar to those used here, where the number of subcarriers is... and Used for chirping.

[0042] To achieve frequency diversity, the same symbols are transmitted on different frequencies. Frequency spacing. It should be greater than the coherent bandwidth. Figure 4 An example is provided where three path channels are assumed, namely... ,produce Delay spread, Let be the bandwidth, and t and f represent time and frequency, respectively. Coherence bandwidth. It is the reciprocal of the delay extension, and therefore is obtained as .like Figure 4 As shown in a), OFDM has a total bandwidth Sending data represented by different padding patterns There are 3 symbols, each of which uses only one subcarrier band, and therefore its frequency spacing is 1. .because Usually greater than Less than Furthermore, OFDM cannot utilize frequency diversity.

[0043] like Figure 4 As shown in b), due to the use of chirp to sweep the frequency across the entire band, AFDM for each The AFDM symbols have frequency spacing that is significantly larger than that for a hypothetical 3-path channel. Therefore, AFDM enables frequency diversity. The arrows on the sub-channels indicate how the frequency of the symbols transmitted in each corresponding sub-channel increases due to chirp.

[0044] Next, refer to Figure 4 From c) to f), as single-carrier transmissions, each symbol of DFT-s-OFDM and DFT-chirped-s-OFDM is transmitted using frequencies spanning a large bandwidth, for example, respectively for... Figure 4 The spreading factors shown in c) and d) of and targeting Figure 4 The figures shown in e) and f) of . Usually greater than Therefore, DFT-s-OFDM can utilize some frequency diversity, while DFT-chirped-s-OFDM can utilize full frequency diversity, similar to AFDM.

[0045] Compared to pure DFT-assisted waveforms (e.g., DFT-s-OFDM) or pure chirped-assisted waveforms (e.g., AFDM), the chirped DFT-s-OFDM modulation proposed in this paper not only achieves frequency diversity but also brings additional coding gain. Figure 4 As shown in d), compared to DFT-s-OFDM with fixed allocated frequency bands and unused frequency bands in between, the DFT-chirped-s-OFDM presented in this paper allows full-band transmission, i.e., effectively utilizing all subcarriers. This full-bandwidth... The use of this avoids deep fading caused by channel fading and results in additional coding gain. Although both AFDM and DFT-chirped-s-OFDM allow full-band transmission, DFT-chirped-s-OFDM has symbols that are extended over multiple time slots, which brings additional coding gain compared to AFDM.

[0046] In addition, from Figure 4The comparison between d) and f) clearly shows that the proposed DFT-chirped-s-OFDM can vary with the DFT spreading factor. Increasing from 2 to 4 results in an enhanced coding gain. For DFT-s-OFDM, from Figure 4 The comparison between c) and e) clearly shows that increasing the spreading factor reduces the number of allocated frequency bands, and more frequency bands or subcarriers remain unused. This makes DFT-s-OFDM susceptible to channel fading, leading to performance degradation. In contrast, DFT-chirped-s-OFDM, by utilizing chirped signals for frequency scanning, can always maintain full-band transmission and spread each symbol across more frequency bands and time slots, which helps to enhance coding gain.

[0047] Note that for full-band transmission, the proposed chirp rate of the DFT-chirp-s-OFDM must be appropriately selected. Clearly, the chirp rate value depends on the DFT spreading factor. The frequency increase at each moment is determined by… Given, among which and It can be a zero or a positive integer value. For example, in Figure 3 middle At this point, the frequency increase at each moment should be Otherwise, the time-frequency plot of DFT-chirped-s-OFDM will be the same as that of DFT-s-OFDM without introducing additional coding gain.

[0048] It is known that DFT precoding allows single-carrier transmission with low PAPR, especially when considering interleaved subcarrier mapping. On the other hand, the chirping following OFDM proposed in this paper uses a constant amplitude chirped signal to scan only the frequency without affecting the amplitude of the data signal. Therefore, the proposed DFT-chirped-s-OFDM maintains the good PAPR of DFT-s-OFDM.

[0049] use , , And exemplary values ​​for 16-QAM (quadrature amplitude modulation). Figure 5 The image shows the complementary cumulative distribution function (CCDF) of PAPR for chirped DFT-s-OFDM relative to the proposed complementary cumulative distribution functions of OFDM, AFDM, OTFS, and DFT-s-OFDM. The CCDF of PAPR is defined as the probability that PAPR exceeds a certain value, i.e., As shown in the figure, the PAPR of DFT-s-OFDM modulation and the proposed method for implementation The CCDF value is approximated by chirped DFT-s-OFDM modulation with phase shift keying (PSK). OFDM, AFDM, and OTFS require .

[0050] Referring back to the discussion on computational complexity above, the following provides examples of using... , , and A comparison is provided between OFDM, AFDM, DFT-s-OFDM, OTFS, and chirped DFT-s-OFDM as exemplary system values. Among the different modulation schemes, OFDM has the lowest complexity at Nlog2N. Normalizing this base value to 1 will provide values ​​for the other modulation schemes discussed herein for comparison.

[0051] The complexity of AFDM follows Nlog2N + 2N, which translates to a 25% increase over OFDM, resulting in a normalized value of 1.25.

[0052] The complexity of DFT-s-OFDM follows Nlog2N + Mlog2M, which translates to a further increase of 15.2% compared to AFDM, resulting in a normalized value of 1.44.

[0053] The complexity of OTFS follows Nlog2N + Nlog2M. OTFS This translates to a 4.2% increase over DFT-s-OFDM to a normalized value of 1.5.

[0054] Finally, the complexity of DFT-chirped-s-OFDM follows N (log2N +1) + Mlog2M, representing a 4% increase over OTFS and producing a normalized value of 1.56.

[0055] It is evident that DFT-chirped-s-OFDM, OTFS, and DFT-s-OFDM have considerable complexity, less than 5% around OTFS.

[0056] Compared to multi-carrier-based waveforms (such as OFDM and AFDM), Figure 4 Some frequency bands of the DFT-s-OFDM and DFT-chirped-s-OFDM shown are unused, resulting in fewer data symbols being transmitted. Note that... Figure 4 The time-frequency graph in the image is provided for a single user. Figure 4 The remaining bandwidth shown in the diagram can be used by other users for data transmission. Therefore, DFT-s-OFDM and DFT-chirped-s-OFDM can achieve the same spectral efficiency as OFDM and AFDM.

[0057] The following sections present a performance comparison of the proposed chirped DFT-s-OFDM with simulations based on OFDM, AFDM, OTFS, and DFT-s-OFDM.

[0058] Following the simulation settings used by A. Bemani, N. Ksairi, and M. Kounturis in “Affine frequency division multiplexing for next generation wireless communications”, IEEE Transactions on Wireless Communications, vol. 22, no. 11, pp. 8214–8229, 2023, the IFFT size of OFDM, AFDM, DFT-s-OFDM, and chirped DFT-s-OFDM is... The number of bins per unit bandwidth for OTFS delay and Doppler is... and For chirped DFT-s-OFDM and DFT-s-OFDM, the size of the DFT is... This results in the DFT expansion factor being... Unless otherwise specified, the carrier frequency and subcarrier spacing are set to... and Consider a speed of The system has three equal-gain channels. The modulation scheme is 4-QAM (4-orthogonal amplitude modulation). The chirp rate of the proposed chirped DFT-s-OFDM is... Linear minimum mean square error (LMMSE) equalization is used for all waveforms.

[0059] Using a maximum likelihood (ML) equalizer as the optimal reference implementation, the error performance of the proposed chirped DFT-s-OFDM modulation was investigated using pairwise error probability (PEP) analysis.

[0060] The upper boundary of the BER of the proposed chirped DFT-s-OFDM modulation is achieved using what is considered the optimal maximum likelihood (ML) equalizer. The system model represented by equation (5) can be rewritten as:

[0061] (6)

[0062] express The received frequency domain signal vector Represented as

[0063] (7)

[0064] in and Data symbol vector This can be estimated using an ML equalizer:

[0065] (8)

[0066] in Represents a candidate vector of data symbols, and is derived from a certain modulation alphabet. The choice was made from [the data type]. Note that the ML equalizer results in high computational complexity, making it less suitable for large data vector sizes. And the size of the large symbol alphabet However, per-survival processing (PSP) can be used to approximate ML equalizers with low complexity.

[0067] Will Represented as paired error events, where Represents the transmitted signal vector and This represents the signal vector that was incorrectly detected using the ML equalizer. and From a specific modulated alphabet (i.e. and ,but Selected from ) Definition . The rank and its non-zero eigenvalues ​​are represented as and Assuming Each input item is an independent, identically distributed composite Gaussian random variable, i.e. The SNR of each data symbol is represented as... ,Right now Following the derivation proposed by Y.Ge, Q. Deng, D. González G., YL Guan and Z. Ding in “OTFS signaling for SCMA with coordinated multi-point vehicle communications”, IEEE Trans. Veh. Technol., vol. 72, no. 7, pp. 9044-9057, 2023, and Z. Sui, H. Zhang, S. Sun, L.-L. Yang and L. Hanzo in “Spacetime shift keying aided OTFS modulation for orthogonal multiple access”, IEEE Trans. Commun., vol. 71, no. 12, pp. 73937408, 2023, PEP can be expressed as:

[0068] (9)

[0069] At high SNR, equation (9) can be approximated as:

[0070] (10)

[0071] The average BER can then be obtained using the joint binding technique discussed in "OTFS signaling for SCMA with coordinated multi-point vehicle communications" (quoted in full above), as follows:

[0072] (11)

[0073] in yes and The number of bits of the difference between them.

[0074] The diversity order of chirped DFT-s-OFDM modulation is given by the following formula:

[0075] (12)

[0076] Calculate using equation (12) It was found that the diversity order is equal to the number of channel paths in chirped DFT-s-OFDM modulation.

[0077]

[0078] Table I shows the number of channel paths calculated using formula (12). Traditional DFT-s-OFDM and the proposed chirped DFT-s-OFDM with different values ​​of Doppler frequency shift. Value. As a value and The basic parameters are provided in the leftmost column. From arrive The diversity order of the randomly generated parameters is calculated and averaged over 100k implementations.

[0079] The value of DFT-s-OFDM By setting the chirping rate to The calculation shows that DFT-s-OFDM cannot achieve full-frequency diversity, and its diversity order varies with the changing Doppler frequency shift. In contrast, the proposed chirped DFT-s-OFDM can utilize full-frequency diversity and exhibits high elasticity to changing Doppler frequency shifts.

[0080] Figure 6 a) shows the upper bound of BER derived using equation (11) and the simulated BER of DFT-s-OFDM and chirped DFT-s-OFDM using the optimal ML equalizer. The DFT size and IFFT size are... and The maximum Doppler frequency, velocity, and subcarrier spacing are: as well as For each Monte Carlo simulation, the value of the Doppler frequency shift is from... arrive Randomly generated. Binary phase-shift keying (BPSK) modulation is used. From Figure 6 It is clear from a) that the assumption is... At high SNR, the simulated BER approaches the upper boundary of the derived BER. The slope of the BER curve for DFT-s-OFDM is flatter than that for chirped DFT-s-OFDM, indicating that DFT-s-OFDM has a smaller diversity order than chirped DFT-s-OFDM, consistent with the results shown in Table I. Regarding chirped DFT-s-OFDM, it was found that its diversity order calculated from its BER curve corresponds to the number of channel paths and is consistent with Equation (12), as shown from... Figure 6 It is obvious in b. In Figure 6 b) shows the range from 1 to 4. The curve of values.

[0081] In the following sections, the output SNR of the proposed chirped DFT-s-OFDM modulation is analyzed using an LMMSE equalizer compared to other waveform candidates. This is achieved by allowing... Equation (5) is rewritten as:

[0082] (13)

[0083] Use LMMSE equalizer It can be estimated as

[0084] (14)

[0085] Note that by using the LMMSE equalizer, inter-symbol interference (ISI) has been mitigated to some extent. According to equation (14), neglecting the effects of noise, the signal estimation is expressed as... ,in The impact of ISI can be seen from... From the off-diagonal elements, we can see that the off-diagonal elements are very small and can be ignored.

[0086] As shown by Y. Jiang, MK Varanasi, and J. Li in “Performance analysis of ZF and MMSE equalizers for MIMO systems: An in-depth study of the high SNR regime” (IEEE Trans. Inf. Theory, vol. 57, no. 4, pp. 2008-2026, 2011), the output SNR of chirped DFT-s-OFDM modulation using an LMMSE equalizer can be expressed as:

[0087] (15)

[0088] in Indicates the expected action. Due to the characteristics of tracking actions, the following formula can be obtained: as well as Equation (15) can therefore be rewritten as

[0089]

[0090] (16)

[0091] in, as well as .

[0092] As a result, the output signal power and output noise power correspond to respectively and The sum of the diagonal elements. and The m-th diagonal elements correspond to the output signal power and output noise power at the m-th symbol, respectively. Figure 7 and 8 The figure shows the output signal power and output noise power of the chirped DFT-s-OFDM relative to the symbol index m. Figure 9 The output SNR relative to symbol index m is shown below. After a similar derivation, for the purpose of comparison, Figures 7 to 9 The output signal power, output noise power, and output SNR of DFT-s-OFDM, AFDM, OTFS, and OFDM are also shown.

[0093] like Figure 7 , 8 As can be readily observed in Figure 9, unlike DFT-s-OFDM, chirped DFT-s-OFDM, AFDM, and OTFS, the output signal power, output noise power, and output SNR of OFDM differ significantly for some symbols. This is because OFDM symbols are modulated in the frequency domain, where different symbols experience different fading due to multipath channels. In other modulation schemes, each symbol is transmitted across almost the entire frequency band, enhancing their resilience to frequency-selective fading. Specifically, each chirped DFT-s-OFDM symbol is modulated in the time domain and transmitted over a wide frequency band. In the case of AFDM, chirping allows each symbol on a particular subcarrier to hop to all other subcarrier bands. For OTFS, each symbol is modulated in the time-delay Doppler domain and spread across the entire frequency band after ISFFT. As a result, besides OFDM, Figure 7 , 8 The other modulation schemes shown in Figure 9 exhibit nearly identical output signal power, output noise power, and output SNR across all symbols.

[0094] As further shown in the figure, the proposed chirped DFT-s-OFDM modulation produces lower output noise power and higher output SNR than DFT-s-OFDM modulation. This is because chirping is included in chirped DFT-s-OFDM, which facilitates full-band transmission. For example, when the DFT size and IFFT size of DFT-s-OFDM and chirped DFT-s-OFDM are set to M=2 and N=8 respectively without enabling full-band transmission with chirping, the two (2) unknown data symbols are mainly estimated based on the data symbols received on the two (2) subcarriers, such as... Figure 4 As shown in e). Conversely, by extending the signal across the entire frequency band, it is possible to determine, as shown in e). Figure 4 The data symbols received on the eight (8) subcarriers shown in f) are used to estimate the two (2) unknown data symbols based on the data symbols received on all subcarriers. Therefore, there are more observations than unknowns, and the redundant information from more observations can mitigate the effects of noise.

[0095] The figure further illustrates that the proposed chirped DFT-s-OFDM outperforms AFDM and OTFS in terms of output noise power and output SNR. This is because the introduction of DFT precoding in chirped DFT-s-OFDM extends the data symbols in the time domain. Figure 4 It is easy to see from e) and f) that, since the spreading factor is set to four (4), the two data symbols are transmitted four times in time. This retransmission can significantly reduce the effect of noise.

[0096] In the following sections, the communication performance of the proposed chirped DFT-s-OFDM modulation is compared with that of existing waveforms (e.g., DFT-s-OFDM, AFDM, OTFS, and OFDM). To ensure a fair comparison, the different waveforms are compared with the same bit-to-noise ratio Eb / N0. The per-active-subcarrier energy of DFT-s-OFDM and chirped DFT-s-OFDM remains the same as that of OFDM, AFDM, and OTFS. Unless otherwise specified, the values ​​of the simulation parameters are set as follows: the IFFT size of chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, and OFDM is set to... Set the DFT size of the chirped DFT-s-OFDM and the DFT size of the DFT-s-OFDM to... ,get The DFT spread factor. The chirp rate is set to... The carrier frequency and subcarrier spacing are respectively set to... and Assume a 3-path equal-gain channel. The speed is set to... , corresponding to The Doppler frequency. For each Monte Carlo simulation, the value of the Doppler frequency shift ranges from... arrive Randomly generated. For OTFS, the number of delay grids and Doppler grids is set to... and It employs quadrature phase shift keying (QPSK) modulation and an LMMSE equalizer.

[0097] Figure 10 and 11The BER and output SNR of the proposed chirped DFT-s-OFDM are shown compared to DFT-s-OFDM, AFDM, OTFS, and OFDM. The proposed chirped DFT-s-OFDM benefits from enhanced noise suppression, as further discussed above, and outperforms existing waveforms in both BER and output SNR, exhibiting an approximately 4 dB SNR gain compared to AFDM, OTFS, and DFT-s-OFDM. While OFDM has an output SNR comparable to OTFS and AFDM, it exhibits... Figure 9 The deep fading shown. At the same time, it cannot utilize frequency diversity, therefore... Figure 10 Its BER is easily identified as the worst.

[0098] Figure 12 The effect of the DFT spreading factor on the BER of DFT-s-OFDM and the proposed chirped DFT-s-OFDM is shown. The BER of DFT-s-OFDM does not benefit from increasing the DFT spreading factor. Conversely, for the proposed chirped DFT-s-OFDM, a significant BER improvement exists, especially when the DFT spreading factor increases from 2 to 4. (See diagram from...) Figure 13 As seen above, this is caused by enhanced noise suppression resulting from the increase in the DFT spreading factor. As further discussed above, this enhanced noise suppression arises from full-band transmission and symbol retransmission, implemented through chirping and DFT precoding, respectively. Note that increasing SF to more than... It will not lead to a significant performance enhancement because increasing SF results in The signal amplitude is reduced, as described by H. Gmyung, J. Lim, and DJ Goodman in “Single carrier FDMA for uplink wireless transmission”, IEEE Trans. Wireless Commun., vol. 19, no. 11, pp. 7139-7152, Nov. 2020; by Y. Shao and SC Liew in “Flexible subcarrier allocation for interleaved frequency division multiple access”, IEEE Trans. Wireless Commun., vol. 19, no. 11, pp. 7139-7152, Nov. 2020; and by MW Chia, BS Thian, and TT Tjhung in “Distributed DFT-spread OFDM”, Proc. 10th IEEE Singapore ICCS, Singapore, 2006, pp. 1-5, February, Singapore. For example, for SF=8 and SF=16, the amplitude reduction will be 0.35 and 0.25, respectively. This amplitude reduction will lead to performance degradation, offsetting the additional benefits gained from the additional repetitive transmissions.

[0099] Note that in previous results, DFT-s-OFDM and chirped DFT-s-OFDM with SF>1 exhibited different spectral efficiencies compared to AFDM and OTFS using all subcarriers. Figure 14 The diagram shows the BER values ​​for different modulation waveforms with the same bandwidth efficiency when SF=4. AFDM and OTFS use a data mapping scheme similar to DFT-s-OFDM and chirped DFT-s-OFDM, where one-quarter of the interleaved subcarriers (for AFDM) or delay grids (for OTFS) are allocated to specific users for data transmission. Unused subcarriers or delay grids can be allocated to other users for data transmission. Figure 10 compared to, Figure 14 The BER values ​​of AFDM and OTFS in the model are low, and almost the same as those of chirped DFT-s-OFDM. This means that chirped DFT-s-OFDM is superior to AFDM in terms of performance. Figure 10 and 14The excellent BER of AFDM and OTFS in [the context of the original text] is mainly generated by DFT precoding. However, even AFDM, OTFS, and chirped DFT-s-OFDM [are mentioned in the original text]. Figure 14 Similar BER values ​​were observed in AFDM and OTFS, which also showed significantly higher PAPR than chirped DFT-s-OFDM.

[0100] In practical implementations, when the input signal is too high, the power amplifier's output will reach its upper limit and cannot increase further, leading to clipping and nonlinear distortion. Considering the clipping ratio is 1, in... Figure 15 The simulation and illustration show the BER values ​​of different waveforms at the same bandwidth efficiency. It was found that AFDM and OTFS are more sensitive to clipping, and their BER values ​​are worse than those of DFT-s-OFDM and chirped DFT-s-OFDM. This is because signals with high PAPR values, such as those in AFDM and OTFS, are susceptible to clipping. Furthermore, considering the same bandwidth efficiency with SF=4, OTFS exhibits a higher PAPR than AFDM, which explains why its BER is higher than AFDM. Figure 15 The reason why the BER of AFDM is more severely downgraded.

[0101] Since 6G and above protocols are also expected to be based on communication signals for sensing, also known as Integrated Sensing and Communication (ISAC), the sensing capabilities of the proposed DFT-Chirped-s-OFDM will be discussed in the following sections.

[0102] The simulation settings used by L. Giroto de Oliveira, B. Nuss, MB Alabd, A. Diewald, M. Pauli, and T. Zwick in “Joint radar-communication systems: Modulation schemes and system design,” IEEE Transactions on Microwave Theory and Techniques, vol. 70, no. 3, pp. 1521–1551, 2022, were adopted. The IFFT sizes of DFT-s-OFDM and DFT-chirped-s-OFDM were set to... The carrier frequency and bandwidth were set to [specific values ​​to be filled in]. and Each data frame is composed of Composed of 1 symbol. The chirp rate of DFT-chirp-s-OFDM is 1. The DFT size of DFT-s-OFDM and DFT-chirped-s-OFDM is... The target's distance and speed are respectively and . The value is -33 dB.

[0103] Consider communication-assisted sensing. Data frames are transmitted and reflected by the target. The reflected data frames are processed by a sensing unit located in the same area as the transmitter to sense the target's distance and velocity. The reflected signal has two dimensions, for example... First Dimension Second dimension These correspond to the fast and slow time axes, respectively. The sensing unit generates an ambiguity function by applying a matched filter, which correlates the received signal with the time inverse conjugate and complex conjugate of the transmitted signal, implementing IFFT along the fast time axis and FFT along the slow time axis.

[0104] Figure 16 The blur function of DFT-chirped-s-OFDM is shown. A distinct peak exists, indicating that a target has been detected. The coordinates of the peak are used to estimate the target's distance and velocity, which are close to their true values. However, if the DFT-s-OFDM data frame is used for sensing, then... Figure 17 As shown, there are two peaks, one of which is an erroneous peak caused by the repetition of data symbols. The number of erroneous peaks tends to increase with the DFT spreading factor, and this erroneous peak may be incorrectly detected as the number of targets. Conversely, due to chirp, the data symbols in DFT-chirped-s-OFDM are not repetitive, thus preventing erroneous peaks and allowing for the correct detection of the number of targets.

[0105] Based on the above discussion, according to a first aspect of the invention, a method is proposed for modulating data symbols representing binary data for transmission over a wireless communication channel. The wireless channel may suffer from dual-selective fading, interference signals, and / or multipath reception. The method includes converting a serial sequence comprising a first number of M data symbols into a first number of M parallel data symbols, and transforming the M parallel data symbols from the time domain to the frequency domain. The method further includes padding the M parallel data symbols in the time domain with a second number of NM padding symbols to generate a third number of N parallel frequency domain transmission symbols. The padding can be zero-padding, although other symbols can be used, for example, symbols designed for channel estimation without causing or minimizing interference to the data symbols. In a further step of the method, the N frequency domain transmission symbols are mapped onto corresponding subcarriers before being transformed to the time domain. The time-domain transmission signal obtained in the previous step is converted into a serial time-domain signal of a first length N, and the first length N serial time-domain signal is chirped using a chirped signal of a corresponding length N. Finally, CP is added to the chirped signal before it is output for transmission.

[0106] In one or more embodiments of the method, transforming the first number M parallel data symbols from the time domain to the frequency domain includes subjecting the symbols to an M-point DFT.

[0107] In one or more embodiments of the method, mapping a third number of N parallel frequency domain transmit symbols to corresponding subcarriers includes applying an interleaving mapping scheme.

[0108] In one or more embodiments of the method, transforming the mapped third number of N frequency-domain transmitted symbols to the time domain includes subjecting the symbols to an N-point IDFT.

[0109] According to a second aspect of the invention, a method for demodulating symbols received via a wireless communication channel is provided. The demodulation method includes receiving a time-domain signal carrying symbols modulated using the method according to a first aspect of the invention, and removing the CP (prefix-coupled symbol) to generate a prefix-free representation of the received signal. In a subsequent step, the prefix-free representation of the received signal is dechirped to generate a signal carrying a third number N transmitted symbols, and the dechirped prefix-free representation of the received signal is converted into multiple parallel segments, the overall representation of which is the dechirped prefix-free received signal. The parallel segments are transformed from the time domain to the frequency domain. Symbol detection is performed on the parallel segments carrying the representations of the transmitted symbols in the frequency domain, thereby generating a representation of the third number N transmitted symbols in the frequency domain. Symbols detected on subbands allocated by the transmitter are remapped to designated carriers. Symbols detected on unallocated subbands may be discarded. The remapped symbols are transformed back to the time domain to generate a first number (M) of parallel received symbols in the time domain, which are converted into a serial stream of received data symbols in the time domain, and the serial stream is output for further processing. The first number (M) of symbols received in parallel corresponds to the third number N of symbols transmitted minus the discarded symbols.

[0110] In one or more embodiments of the method according to the second aspect of the invention, transforming parallel segments from the time domain to the frequency domain includes subjecting the parallel time-domain signals to an N-point Fast Fourier Transform.

[0111] In one or more embodiments of the method according to the second aspect of the invention, performing symbol detection further includes performing channel estimation.

[0112] In one or more embodiments of the method according to the second aspect of the present invention, performing symbol detection includes applying equalization.

[0113] In one or more embodiments of the method according to the second aspect of the invention, transforming the remapped symbols to the time domain includes applying an M-point discrete Fourier inverse transform to the transmitter-assigned subbands and the symbols mapped thereon.

[0114] According to a third aspect of the invention, a transmitter is provided for implementing the method proposed in the first aspect of the invention. The transmitter includes an antenna and circuitry for processing radio frequency signals, such as an oscillator, mixer, amplifier, filter, etc. The transmitter also includes one or more microprocessors and volatile and non-volatile memories. The aforementioned elements and components are connected via one or more data and / or signal lines or buses. The non-volatile memory stores computer program instructions that, when executed by the one or more microprocessors, configure the components of the wireless transmitter to implement or execute the method according to the first aspect of the invention for obtaining or receiving a modulated signal according to the first aspect of the invention, and transmitting the modulated signal via the circuitry for processing radio frequency signals and the antenna.

[0115] According to a fourth aspect of the invention, a receiver is provided for implementing the method proposed in the second aspect of the invention. The receiver includes an antenna and circuitry for processing radio frequency signals, such as an oscillator, mixer, amplifier, filter, etc. The receiver also includes one or more microprocessors and volatile and non-volatile memories. The aforementioned elements and components are connected via one or more data and / or signal lines or buses. The non-volatile memory stores computer program instructions that, when executed by the one or more microprocessors, configure the components of the wireless receiver to implement or perform the method according to the second aspect of the invention.

[0116] As those skilled in the art will understand, aspects of the embodiments can be implemented as a system, apparatus, method, or program product. Therefore, embodiments can take the form of entirely hardware embodiments, entirely software-implemented embodiments (including firmware, resident software, microcode, etc.), or embodiments combining software and hardware aspects.

[0117] For example, the disclosed embodiments can be implemented as hardware circuitry including custom very large-scale integration (VLSI) circuitry or gate arrays, off-the-shelf semiconductors such as logic chips, transistors, or other discrete components. The disclosed embodiments can also be implemented as programmable hardware devices such as field-programmable gate arrays, programmable array logic, programmable logic devices, etc. As another example, the disclosed embodiments may include one or more physical or logical blocks of executable code, which may, for example, be organized as objects, procedures, or functions.

[0118] The methods presented above can be represented by computer program instructions. Therefore, according to another aspect of the invention, a computer program product includes computer program instructions that, when executed by a microprocessor of a wireless transmitter according to a third aspect of the invention, cause the microprocessor to perform the method according to a first aspect of the invention and correspondingly control the hardware and / or software blocks or modules of the wireless transmitter; or, when executed by a microprocessor of a wireless receiver according to a fourth aspect of the invention, cause the microprocessor to perform the method according to a second aspect of the invention and correspondingly control the hardware and / or software blocks or modules of the wireless receiver.

[0119] The computer program instructions or code used to perform the operations of the embodiments can be any number of lines and can be written in any combination of one or more programming languages, including object-oriented programming languages ​​such as Python, Ruby, Java, Smalltalk, C++, etc., and conventional procedural programming languages ​​such as the "C" programming language, etc., and / or machine languages ​​such as assembly language. The code can be executed entirely on the user's computer, partially on the user's computer, as a standalone software package, partially on the user's computer and partially on a remote computer, or entirely on a remote computer or server. In the latter case, the remote computer can be connected to the user's computer via any type of network, including a local area network (LAN), wireless LAN (WLAN), or wide area network (WAN), or can be connected to the external computer, for example, through the Internet provided by an Internet service provider (ISP).

[0120] Computer program instructions can be retrieved and stored or transmitted on a computer-readable medium or data carrier. The medium or data carrier can be tangibly or physically implemented, for example, in the form of a hard disk, solid-state drive, flash memory device, etc. However, the medium or data carrier may also include modulated electromagnetic, electrical, or optical signals, which are received by the computer by means of a corresponding receiver and transmitted and stored in the computer's memory.

[0121] Another aspect of the present invention relates to wireless signals generated by performing the method according to the first aspect of the invention.

[0122] The features, structures, or characteristics of the described embodiments can be combined in any suitable manner. Numerous specific details, such as examples of programming, software modules, user selection, network transactions, database queries, database structures, hardware modules, hardware circuits, hardware chips, etc., are provided in this specification to provide a thorough understanding of the embodiments. However, those skilled in the art will recognize that the embodiments can be implemented without one or more specific details, or using other methods, components, materials, etc. In other instances, well-known structures, materials, or operations have not been shown or described in detail to avoid obscuring aspects of the embodiments. Throughout this specification, references to “an embodiment,” “embodiment,” or similar language mean that a particular feature, structure, or characteristic described in connection with that embodiment is included in at least one embodiment. Therefore, the phrases “in one embodiment,” “in an embodiment,” and similar language appearing throughout this specification may, but do not necessarily, refer to the same embodiment, but rather mean “one or more, but not all, embodiments,” unless expressly stated otherwise. The terms “comprising,” “including,” “having,” and variations thereof mean “including, but not limited to,” unless expressly stated otherwise. Unless expressly stated otherwise, the list of items does not imply that any or all items are mutually exclusive. The terms “a,” “one,” and “the” also mean “one or more”, unless otherwise explicitly stated.

[0123] In this specification, where aspects of embodiments are described with reference to schematic flowcharts and / or schematic block diagrams of methods, apparatus, systems, and program products according to embodiments, it will be understood that each block of the schematic flowcharts and / or schematic block diagrams, and combinations of blocks in the schematic flowcharts and / or schematic block diagrams, can be implemented by code. This code can be provided to a processor of a general-purpose computer, special-purpose computer, or other programmable data processing apparatus to generate machinery, such that instructions executable via the processor of the computer or other programmable data processing apparatus create means for implementing the functions / actions specified in the flowcharts and / or block diagrams.

[0124] It should be noted that in some implementations or embodiments, the functions mentioned in the exemplary embodiments shown in the figures may not occur in the order shown in the figures. For example, depending on the functions involved, two consecutively shown blocks may actually be executed substantially simultaneously, or these blocks may sometimes be executed in reverse order. Other steps and methods that are functionally, logically, or effectively equivalent to one or more blocks or portions thereof shown in the figures are conceivable.

[0125] The novel single-carrier DFT-chirped-s-OFDM modulation scheme proposed in this paper, obtained by chirping DFT-s-OFDM in the time domain, maintains a lower PAPR than DFT-s-OFDM modulation schemes such as AFDM, OFDM, and OTFS. Full-band transmission and symbol retransmission achieved through chirping and DFT precoding provide enhanced noise suppression in LMMSE equalization. Furthermore, the novel DFT-chirped-s-OFDM exhibits a higher output SNR and a lower BER value than conventional modulation schemes including DFT-s-OFDM, AFDM, OTFS, and OFDM. In addition, the proposed waveform achieves full frequency diversity and exhibits high resilience to Doppler shift. Last but not least, the low PAPR of the proposed chirped DFT-s-OFDM modulation provides greater resistance to clipping than multi-carrier waveforms with high PAPR.

[0126] This invention can be used in various scenarios, especially in high-speed mobile communication scenarios and autonomous driving, such as in 6G communication and above scenarios. Attached Figure Description

[0127] In the following sections, the invention will be described with reference to the accompanying drawings, wherein...

[0128] Figure 1 A schematic comparison of the core operations required to obtain the transmitted signal from the conventional modulation schemes discussed in this paper is shown.

[0129] Figure 2 A schematic block diagram of a DFT-chirped-s-OFDM communication system according to the present invention is shown.

[0130] Figure 3 This illustrates the core operations supplemented by the method according to the invention. Figure 1 A schematic comparison of the core operations required to obtain the transmitted signal using traditional modulation schemes.

[0131] Figure 4 The use of total bandwidth for a conventional modulation scheme and a modulation scheme according to the present invention is illustrated by way of example.

[0132] Figure 5 The comparison of CCDF and PAPR thresholds for OFDM, AFDM, OTFS, and DFT-based modulation schemes is shown.

[0133] Figure 6 The upper boundary of the BER and the simulated BER of DFT-s-OFDM and chirped DFT-s-OFDM are shown.

[0134] Figure 7The output signal power of the proposed chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, OTFS, and OFDM relative to the symbol index is shown.

[0135] Figure 8 The output noise power of the proposed chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, OTFS, and OFDM relative to the symbol index is shown.

[0136] Figure 9 The output SNR of the proposed chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, OTFS, and OFDM relative to the symbol index is shown.

[0137] Figure 10 The BER relative to Eb / N0 is shown for the proposed chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, OTFS, and OFDM.

[0138] Figure 11 The output SNR relative to Eb / N0 is shown for the proposed chirped DFT-s-OFDM, DFT-s-OFDM, AFDM, OTFS, and OFDM.

[0139] Figure 12 The effect of the DFT spreading factor on the BER of DFT-s-OFDM and the proposed chirped DFT-s-OFDM is shown.

[0140] Figure 13 The effect of the DFT spreading factor on the output noise power of DFT-s-OFDM and the proposed chirped DFT-s-OFDM is shown.

[0141] Figure 14 The BER values ​​for various modulation waveforms on Eb / N0 are shown, without clipping, and with the same bandwidth efficiency.

[0142] Figure 15 The BER values ​​for various modulation waveforms at Eb / N0 with the same bandwidth efficiency and under limiting are shown.

[0143] Figure 16 The blur function of DFT-chirped-s-OFDM for the exemplary scene is shown.

[0144] Figure 17 It shows Figure 16 The fuzzy function of DFT-s-OFDM for an exemplary scene.

[0145] Figure 18 Exemplary block diagrams of transmitters or receivers according to the third or fourth aspect of the present invention are shown respectively.

[0146] Figure 19 An exemplary flowchart of a method for modulating and representing symbols of binary data for transmission over a wireless communication channel according to a first aspect of the present invention is shown, and

[0147] Figure 20 An exemplary flowchart of a method for demodulating symbols received on a wireless communication channel according to a second aspect of the present invention is shown.

[0148] In the accompanying drawings, the same or similar elements may be represented by the same reference numerals. Detailed Implementation

[0149] Figures 1 to 17 This has already been described above and will not be discussed further. Figure 18 An exemplary block diagram is shown of a transmitter 500 or a receiver 600 configured to respectively perform a method according to the first or second aspect of the present invention. The transmitter 500 or receiver 600 respectively includes at least one antenna 580 or 610, and circuitry 501 for processing radio frequency signals, one or more microprocessors 502, volatile memory 503, and non-volatile memory 504. The aforementioned components or elements are connected via one or more data and / or signal lines or buses 505. The non-volatile memory 504 stores computer program instructions that, when executed by one or more microprocessors 502, respectively configure the components of the transmitter 500 to implement or perform the method according to the first aspect of the present invention, or configure the components of the receiver 600 to implement or perform the method according to the second aspect of the present invention.

[0150] Figure 19 An exemplary flowchart of a method 100 for transmitting modulated binary data symbols over a wireless communication channel according to a first aspect of the present invention is shown. After receiving a serial data sequence of data symbols in the time domain in step 105, the serial sequence is transformed to the frequency domain in step 110, and then converted into a first number of M parallel data symbols in step 120. In step 130, the first number of M parallel data symbols are padded with a second number of N–M symbols to generate a third number of N parallel frequency domain symbols. In step 140, the third number of N parallel frequency domain symbols are mapped to corresponding subcarriers. Next, in step 150, the mapped third number of N parallel frequency domain symbols are transformed to the time domain, and then converted to a first length NT in step 160. s The serial time-domain signal. Before adding CP in step 180, the serial time-domain signal is chirped in step 170. The modulated signal is output in step 190.

[0151] Figure 20An exemplary flowchart of a method 200 for demodulating symbols received on a wireless communication channel according to a second aspect of the present invention is shown. In step 210, symbols carrying [symbols] according to [reference] are received. Figure 10 After modulating the time-domain signal of the symbols discussed in method 100, any CP that may exist in the received time-domain signal is removed in step 220, thereby generating an unprefixed representation of the received signal. Before converting the unprefixed representation of the received signal into a third number of N parallel representations of the received symbols in the frequency domain in step 240, the unprefixed representation of the received signal is dechirped in step 230. In step 250, the third number of N parallel representations of the received symbols in the frequency domain are transformed into the frequency domain, and symbol detection is performed on the parallel signals in step 260. Symbol detection may include channel estimation (not shown in the figure) and generate transmitted frequency-domain symbols. In step 270, the symbols detected on the subband allocated by the transmitter are remapped to a designated carrier, and in step 280, the resulting parallel signals are converted from the frequency domain into corresponding parallel signals in the time domain. Then, before outputting the demodulated signal in step 295, the parallel signals in the time domain are converted into serial signals in step 290.

[0152] Reference Markings List (Instruction Manual Section)

[0153] 100 methods

[0154] 105 Serial sequence of received data symbols

[0155] 110 t → f domain (time domain to frequency domain) conversion

[0156] 120 S2P Conversion

[0157] 130 fill

[0158] 140 Subcarrier Mapping

[0159] 150 f → t domain (frequency domain to time domain) conversion

[0160] 160 P2S conversion

[0161] 170 chirps

[0162] 180 CP insertion

[0163] 190 Output modulation signal

[0164] 200 methods

[0165] 210 Receive time-domain signal

[0166] 220 Remove CP

[0167] 230 Go chirp

[0168] 240 S2P conversion

[0169] 250 t → f domain (time domain to frequency domain) conversion

[0170] 260 symbol detection

[0171] 270 Subcarrier Remapping

[0172] 280 f → t domain (frequency domain to time domain) conversion

[0173] 290 P2S conversion

[0174] 295 Output demodulated signal

[0175] 500 transmitters

[0176] 501 Radio Frequency Circuit

[0177] 502 microprocessor

[0178] 503 Volatile Memory

[0179] 504 Non-volatile Memory

[0180] 505 Data / Signal Line / Bus

[0181] 510 S2P Conversion

[0182] 520 t → f domain (time domain to frequency domain) T → F domain conversion

[0183] 530 Padding / Subcarrier Mapping

[0184] 540 f → t domain (frequency domain to time domain) conversion

[0185] 550 P2S conversion

[0186] 560 chirps

[0187] 570 CP insertion

[0188] 580 antenna

[0189] 600 receiver

[0190] 610 antenna

[0191] 620 CP removed

[0192] 630 Go chirp

[0193] 640 S2P Conversion

[0194] 650 t → f domain (time domain to frequency domain) conversion

[0195] 660 symbol detection

[0196] 670 Subcarrier Remapping

[0197] 680 f → t domain (frequency domain to time domain) conversion

[0198] 690 P2S conversion

Claims

1. A method (100) for modulating data symbols representing binary data for transmission over a wireless communication channel, comprising: - Convert (110) the serial sequence containing a first number (M) of data symbols into a first number (M) of parallel data symbols. - Transform the first number (M) of parallel data symbols from the time domain (120) to the frequency domain. - The first number (M) of parallel data symbols in the frequency domain (130) are filled with a second number (NM) of filling symbols to generate a third number (N) of parallel frequency domain transmission symbols. - Map (140) the third number (N) of parallel frequency-domain transmitted symbols to the corresponding subcarriers. - Transform (150) the mapped third number (N) of parallel frequency domain transmit symbols to the time domain. - Convert the third number (N) of parallel time-domain transmitted symbols (160) into a serial time-domain signal of the first length. - The serial time-domain signal of the first length is chirped (170) using a chirp signal of corresponding length. - Add (180) to the chirped signal, and - The signal modulated by the output (190) is used for transmission.

2. The method (100) of claim 1, wherein transforming the first number (M) of parallel data symbols from the time domain (120) to the frequency domain includes subjecting the symbols to an M-point Discrete Fourier Transform (DFT).

3. The method (100) according to claim 1 or 2, wherein, Padding includes zero padding.

4. The method (100) as described in any of the preceding claims, wherein mapping (140) the third number (N) of parallel frequency domain transmit symbols to the corresponding subcarriers includes applying an interleaving mapping scheme.

5. The method (100) of any of the preceding claims, wherein transforming (150) the mapped third number (N) of frequency-domain transmitted symbols to the time domain comprises subjecting the symbols to an N-point inverse discrete Fourier transform (IDFT).

6. A method (200) for demodulating symbols received on a wireless communication channel, comprising: - Receive (210) a time-domain signal carrying symbols modulated by the method (100) according to one or more of claims 1 to 5, - Remove any cyclic prefixes (220) that may exist in the received time-domain signal to produce a prefix-free representation of the received signal. - Dechirp (230) the unprefixed representation of the received signal to generate a signal carrying a third number (N) of transmit symbols. - The dechirped, prefix-free representation of the received signal is transformed (240) into multiple parallel segments, the entirety of which represents the dechirped, prefix-free received signal. - Transform the parallel segments from the time domain (250) to the frequency domain. - Perform (260) symbol detection on the parallel segments in the frequency domain to produce representations of the third number (N) of emitted symbols in the frequency domain. - The symbols detected on the subband assigned by the transmitter will be remapped (270) to the corresponding carrier. - The remapped symbols are transformed (280) into the time domain to produce a first number (M) of parallel received symbols in the time domain. - Convert (290) the first number (M) of parallel received symbols in the time domain into a serial stream of data symbols in the time domain, and - Output the serial stream of (295) data symbols in the time domain.

7. The method (200) according to claim 6, wherein, Transforming the parallel segments from the time domain (250) to the frequency domain includes subjecting the parallel time-domain signals to an N-point Fast Fourier Transform.

8. The method (200) according to claim 6 or 7, wherein performing the symbol detection (260) further includes performing channel estimation.

9. The method (200) according to one or more of claims 6 to 8, wherein, Performing the symbol detection described in (250) involves applying an equalization, including maximum likelihood (ML) equalization or minimum mean square error (MMSE) equalization.

10. The method (200) according to one or more of claims 6 to 9, wherein, Transforming the remapped symbols (280) into the time domain involves applying an M-point discrete Fourier inverse transform to the subbands assigned to the transmitter and the symbols mapped onto them.

11. A transmitter (500) comprising an antenna (502), circuitry (504) for processing radio frequency signals, one or more microprocessors (506), volatile memory (508), and non-volatile memory (510) connected via one or more data and / or signal lines or buses (512), wherein, The non-volatile memory (510) stores computer program instructions that, when executed by the one or more microprocessors (506), configure components of the wireless transmitter (500) to implement or perform the method according to one or more of claims 1 to 5 for obtaining a modulated signal, or receiving a modulated signal according to one or more of claims 1 to 5, and transmitting the modulated signal via the circuitry (504) for processing radio frequency signals and the antenna (502).

12. A receiver (600) comprising an antenna (502), circuitry (504) for processing radio frequency signals, one or more microprocessors (506), volatile memory (508), and non-volatile memory (510) connected via one or more data and / or signal lines or buses (512), wherein, The non-volatile memory (510) stores computer program instructions that, when executed by the one or more microprocessors (506), configure components of the wireless receiver (600) to implement or perform the method according to one or more of claims 6 to 10.

13. A computer program product comprising computer program instructions, wherein the computer program instructions - When executed by the microprocessor of the wireless transmitter (500) according to claim 11, the wireless transmitter (500) and / or the control hardware block, module, or component of the wireless transmitter (500) respectively implement or execute the method according to one or more of claims 1 to 5, or - When executed by the microprocessor of the wireless receiver (600) according to claim 12, the wireless receiver (600) and / or the control hardware block, module or component of the wireless receiver (600) respectively implement or perform the method according to one or more of claims 6 to 10.

14. A computer-readable medium or data carrier capable of retrievably transmitting or storing the computer program product of claim 13.

15. A wireless communication signal carrying modulation symbols representing binary data, characterized in that, The signal is generated using one or more of the methods described in claims 1 to 5.

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