Switching control method and energy storage system

Abstract

The application discloses a switching control method and an energy storage system. The switching control method comprises the following steps: determining the working mode of a DC-DC switching conversion circuit, wherein the working mode comprises a continuous conduction mode and a discontinuous conduction mode; determining the duty cycle of the DC-DC switching conversion circuit and calculating a feedforward compensation coefficient according to the topology type and the working mode of the DC-DC switching conversion circuit, wherein the topology type comprises a step-down type, a step-up type and a step-up / down type; sampling an output voltage and determining the ripple component in the output voltage; determining a compensation amount according to the product of the feedforward compensation coefficient and the ripple component; determining a target duty cycle according to the sum of the duty cycle and the compensation amount; and generating a pulse width modulation signal according to the target duty cycle to control the switch tube in the DC-DC switching conversion circuit. The influence of the double power frequency ripple on the input current can be inhibited, so that the stability and efficiency of the system comprising the DC-DC switching conversion circuit are improved.

Description

Technical Field

[0001] This application relates to the field of switching transistor control technology, and in particular to a switching control method and an energy storage system. Background Technology

[0002] In the field of battery charging and power supply, BUCK (step-down), BOOST (boost), and BUCK-BOOST (step-up / buck-down) circuits have been widely used. During actual operation, the battery charging circuit is susceptible to the influence of the downstream inverter power characteristics, causing a continuous ripple at twice the power frequency at the battery voltage terminal. This ripple cannot be completely isolated by the power loop and is conducted to the power input side via reverse coupling through the circuit power loop. This results in periodic fluctuations and distortions in the input current at the same frequency, reducing the system's stability and efficiency. Summary of the Invention

[0003] This application provides a switching control method and an energy storage system that can suppress the influence of double power frequency ripple on the input current, thereby improving the stability and efficiency of the system including a DC-DC switching converter circuit.

[0004] In a first aspect, embodiments of this application provide a switching control method for controlling a switching transistor in a DC-DC switching converter circuit. The DC-DC switching converter circuit is electrically connected between an input power supply and a battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with twice the power frequency ripple. The switching control method includes: determining the operating mode of the DC-DC switching converter circuit, wherein the operating mode includes continuous conduction mode and intermittent conduction mode; determining the duty cycle of the DC-DC switching converter circuit according to the topology type and operating mode of the DC-DC switching converter circuit, and calculating the feedforward compensation coefficient, wherein the topology type includes buck type, boost type, and buck-boost type; sampling the output voltage of the DC-DC switching converter circuit and determining the ripple component in the output voltage; determining the compensation amount according to the product of the feedforward compensation coefficient and the ripple component; determining the target duty cycle according to the sum of the duty cycle and the compensation amount; and generating a pulse width modulation signal according to the target duty cycle to control the switching transistor in the DC-DC switching converter circuit.

[0005] In one or more embodiments, the duty cycle of the DC-DC switching converter is determined according to the topology and operating mode of the DC-DC switching converter, and the feedforward compensation coefficient is calculated, including: the feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter with respect to the output voltage, wherein the partial derivative is obtained from a preset mapping relationship according to the topology and operating mode.

[0006] In one or more embodiments, the feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter circuit with respect to the output voltage of the DC-DC switching converter circuit, including: when the operating mode is continuous conduction mode and the topology type is buck converter type, the duty cycle is: V out / V in The feedforward compensation coefficient is: 1 / V in , where V out V is the output voltage. in This is the input voltage of the input power supply; when the operating mode is continuous conduction mode and the topology is boost type, the duty cycle is: 1-V. in / V out The feedforward compensation coefficient is: V in / V out 2 When the operating mode is continuous conduction mode and the topology type is buck-boost type, the duty cycle is: V out / (V in +V out The feedforward compensation coefficient is: V in / (V in +V out ) 2 .

[0007] In one or more embodiments, the feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter with respect to the output voltage of the DC-DC switching converter, including: when the operating mode is intermittent conduction mode and the topology is buck converter, the duty cycle is: The feedforward compensation coefficient is: [L×f1×I] out ×(V in +V out )] / [D×(V out -V in ) 3 ], where L is the inductance value of the inductor in the DC-DC switching converter circuit, f1 is the switching frequency of the switching transistor, and I out V is the output current of the DC-DC switching converter circuit. out V is the output voltage. in Where is the input voltage of the input power supply, and D is the duty cycle; when the operating mode is intermittent conduction mode and the topology type is boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×V in 2 When the operating mode is intermittent conduction mode and the topology type is buck-boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×Vin 2 ).

[0008] In one or more embodiments, determining the operating mode of the DC-DC switching converter circuit includes: determining the critical output current corresponding to each topology type; when the output current of the DC-DC switching converter circuit is greater than the critical output current, the operating mode is continuous conduction mode; when the output current is less than or equal to the critical output current, the operating mode is intermittent conduction mode.

[0009] In one or more embodiments, determining the critical output current corresponding to each topology type includes: when the topology type is a buck type, the critical output current is V. out ×(V in -V out ) / (2L×f1×V in ), where V out V is the output voltage. in Where is the input voltage of the input power supply, L is the inductance value of the inductor in the DC-DC switching converter circuit, and f1 is the switching frequency of the switching transistor; when the topology is boost type, the critical output current is V. in ×(V out -V in ) / (2L×f1×V out When the topology is buck-boost type, the critical output current is V. out ×V in / [2L×f1×(V in +V out )).

[0010] In one or more embodiments, determining the ripple component in the output voltage includes: the ripple component being the deviation of the output voltage from a reference voltage, wherein the reference voltage is obtained based on the ripple characteristics of the output voltage.

[0011] In one or more embodiments, the reference voltage is obtained based on the ripple characteristics of the output voltage, including: performing a sine fit on the output voltage to a second harmonic corresponding to the power frequency to obtain a fitting result; determining a fitting quality index based on the fitting result; when the fitting quality index is greater than a preset threshold, using the root mean square value of the output voltage as the reference voltage, otherwise using the average value of the output voltage as the reference voltage.

[0012] Secondly, embodiments of this application provide an energy storage system, including: a DC-DC switching converter circuit, an input power supply, and a battery. The DC-DC switching converter circuit is electrically connected between the input power supply and the battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with twice the power frequency ripple. Also included is a control processing unit electrically connected to the DC-DC switching converter circuit to control the switching transistors in the DC-DC switching converter circuit. The control processing unit includes: at least one processor and a memory. The memory is coupled to the processor and is used to store instructions or programs. When the instructions or programs are executed by the at least one processor, the at least one processor performs the switching control method as described in the first aspect.

[0013] In one or more embodiments, the DC-DC switching converter circuit includes a first switch, a second switch, a third switch, a fourth switch, and an inductor; the first switch and the second switch are connected in series between the positive and negative terminals of the input power supply, and the connection point between the first switch and the second switch is electrically connected to the first terminal of the inductor; the third switch and the fourth switch are connected in series between the positive and negative terminals of the battery, and the connection point between the third switch and the fourth switch is electrically connected to the second terminal of the inductor; the control processing unit is electrically connected to the first switch, the second switch, the third switch, and the fourth switch, respectively.

[0014] The beneficial effects of this application are as follows: The switching control method of this application embodiment is used to control the switching transistor in a DC-DC switching converter circuit. The DC-DC switching converter circuit is electrically connected between the input power supply and the battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with twice the power frequency ripple. The switching control method first determines whether the DC-DC switching converter circuit operates in continuous conduction mode or discontinuous conduction mode; then, based on the topology and operating mode of the DC-DC switching converter circuit, it determines the duty cycle of the DC-DC switching converter circuit and calculates the feedforward compensation coefficient; next, it samples the output voltage of the DC-DC switching converter circuit and determines the ripple component in the output voltage. Based on the product of the feedforward compensation coefficient and the ripple component, it determines the compensation amount. Based on the sum of the duty cycle and the compensation amount, it determines the target duty cycle; finally, it generates a pulse width modulation signal based on the target duty cycle to control the switching transistor in the DC-DC switching converter circuit. In this way, by adding feedforward compensation to the duty cycle, the double power frequency ripple can be suppressed, which helps to improve the stability of the input current of the input power supply, thereby improving the stability and efficiency of the system including the DC-DC switching converter circuit. Attached Figure Description

[0015] One or more embodiments are illustrated by way of example with reference to the accompanying drawings, which are not intended to limit the embodiments, and elements having the same reference numerals in the drawings are designated as similar elements.

[0016] Figure 1 This is a schematic diagram of the energy storage system provided in the embodiments of this application. Figure 1 ; Figure 2 This is a schematic diagram of the energy storage system provided in the embodiments of this application. Figure 2 ; Figure 3 This is a schematic diagram of the control processing unit provided in an embodiment of this application; Figure 4 This is a flowchart of the switch control method provided in the embodiments of this application; Figure 5 This is a schematic diagram of the input current when the switching control method of this application is not used, provided in an embodiment of this application; Figure 6 This is a schematic diagram of the input current when the switching control method of this application is used, provided in an embodiment of this application. Detailed Implementation

[0017] To make the objectives, technical solutions, and advantages of this application clearer, the technical solutions of this application will be described clearly and in detail below with reference to the accompanying drawings. Obviously, the embodiments in this application are only some embodiments, not all embodiments. It should be understood that the specific embodiments described herein are only used to explain this application and are not intended to limit this application.

[0018] It should be noted that when an element is described as "connected" to another element, it can be directly connected to the other element, or there can be one or more intermediate elements between them.

[0019] Furthermore, the technical features involved in the various embodiments of this application described below can be combined with each other as long as they do not conflict with each other.

[0020] Please refer to Figure 1 , Figure 1 This is a schematic diagram of the composition of an energy storage system provided in an embodiment of this application. Figure 1 As shown, the energy storage system 100 includes a DC-DC switching converter circuit 110, an input power supply 120, and a battery 130.

[0021] The DC-DC switching converter circuit 110 is electrically connected between the input power supply 120 and the battery 130. In some embodiments, the battery 130 is electrically connected to the AC-DC switching converter circuit 210, which is electrically connected to the power grid 220 and the load 230. The AC-DC switching converter circuit 210 is a high-frequency switching power electronic converter circuit that uses power switching devices to construct a bidirectional energy flow topology. It can convert the AC power from the power grid 220 to DC power and invert the DC power from the battery 130 to AC power, realizing bidirectional energy exchange between the power grid 220 and the battery 130. In a specific embodiment, the AC-DC switching converter circuit 210 is an inverter.

[0022] Figure 2 An exemplary circuit structure of a DC-DC switching converter circuit 110 is shown. For example... Figure 2 As shown, the DC-DC switching converter circuit 110 includes a first switch Q1, a second switch Q2, a third switch Q3, a fourth switch Q4, and an inductor L1.

[0023] The first switch Q1 and the second switch Q2 are connected in series between the positive and negative terminals of the input power supply 120. The connection point between the first switch Q1 and the second switch Q2 is electrically connected to the first terminal of the inductor L1. The third switch Q3 and the fourth switch Q4 are connected in series between the positive and negative terminals of the battery 130. The connection point between the third switch Q3 and the fourth switch Q4 is electrically connected to the second terminal of the inductor L1.

[0024] In this embodiment, each switch (including the first switch Q1, the second switch Q2, the third switch Q3, and the fourth switch Q4) is an NMOS transistor. The gate of the NMOS transistor is the first terminal of each switch, the source of the NMOS transistor is the second terminal of each switch, and the drain of the NMOS transistor is the third terminal of each switch.

[0025] In addition, each switching transistor can be any controllable switch, such as an insulated gate bipolar transistor (IGBT) device, an integrated gate commutated thyristor (IGCT) device, a gate turn-off thyristor (GTO) device, a silicon controlled rectifier (SCR) device, a junction gate field-effect transistor (JFET) device, a MOS controlled thyristor (MCT) device, etc.

[0026] When the first switch Q1 and the second switch Q2 are controlled by the pulse width modulation signal output by the control processing unit 140, and the third switch Q3 remains on and the fourth switch Q4 remains off, the topology of the DC-DC switching converter circuit 110 is a step-down type. At this time, the voltage of the input power supply 120 (i.e., the input voltage) is greater than the output voltage of the DC-DC switching converter circuit 110.

[0027] When the third switch Q3 and the fourth switch Q4 are controlled by the pulse width modulation signal output by the control processing unit 140, and the first switch Q1 remains on and the second switch Q2 remains off, the topology of the DC-DC switching converter circuit 110 is a boost type. At this time, the voltage of the input power supply 120 (i.e., the input voltage) is less than the output voltage of the DC-DC switching converter circuit 110.

[0028] When the first switch Q1, the second switch Q2, the third switch Q3, and the fourth switch Q4 are controlled by the pulse width modulation signal output by the control processing unit 140, the topology of the DC-DC switching converter circuit 110 is a buck-boost type. At this time, the voltage of the input power supply 120 (i.e., the input voltage) is greater than, equal to, or less than the output voltage of the DC-DC switching converter circuit 110.

[0029] Please return to the reference. Figure 1 The energy storage system 100 also includes a control processing unit 140. The control processing unit 140 is electrically connected to the DC-DC switching converter circuit 110 to control the switching transistors in the DC-DC switching converter circuit 110. Specifically, the control processing unit 140 is electrically connected to the first switching transistor Q1, the second switching transistor Q2, the third switching transistor Q3, and the fourth switching transistor Q4 to control the on or off state of the first switching transistor Q1, the second switching transistor Q2, the third switching transistor Q3, and the fourth switching transistor Q4.

[0030] The control processing unit 140 can be a microcontroller unit (MCU) or a digital signal processing (DSP) controller, etc.

[0031] like Figure 3 As shown, the control processing unit 140 includes at least one processor 141 and a memory 142. The memory 142 can be built into the control processing unit 140 or external to the control processing unit 140. The memory 142 can also be a remotely configured memory connected to the control processing unit 140 via a network.

[0032] Memory 142, as a non-volatile computer-readable storage medium, can be used to store non-volatile software programs, non-volatile computer-executable programs, and modules. Memory 142 may include a program storage area and a data storage area, wherein the program storage area may store the operating system and application programs required for at least one function; the data storage area may store data created based on the use of the terminal, etc. Furthermore, memory 142 may include high-speed random access memory and may also include non-volatile memory, such as at least one disk storage device, flash memory device, or other non-volatile solid-state storage device. In some embodiments, memory 142 may optionally include memory remotely located relative to processor 141, and these remote memories can be connected to the terminal via a network. Examples of such networks include, but are not limited to, the Internet, corporate intranets, local area networks, mobile communication networks, and combinations thereof.

[0033] The processor 141 performs various functions of the terminal and processes data by running or executing software programs and / or modules stored in the memory 142 and calling data stored in the memory 142, thereby performing overall monitoring of the terminal, such as implementing the switch control method described in any embodiment of this application.

[0034] Processor 141 can be one or more. Figure 3 The example provided is a processor 141. Processor 141 and memory 142 can be connected via a bus or other means. Processor 141 may include a central processing unit (CPU), a digital signal processor (DSP), an application-specific integrated circuit (ASIC), a controller, a field-programmable gate array (FPGA) device, etc. Processor 141 can also be implemented as a combination of computing devices, such as a combination of a DSP and a microprocessor, multiple microprocessors, one or more microprocessors combined with a DSP core, or any other such configuration.

[0035] for Figure 1 and Figure 2 In the structure shown, when the input power supply 120 charges the battery 130 through the DC-DC switching converter circuit 110, if the battery 130 is already connected to the AC-DC switching converter circuit 210, a double power frequency ripple will be generated during the operation of the AC-DC switching converter circuit 210, resulting in the battery 130 voltage being superimposed with the double power frequency ripple. This double power frequency ripple cannot be completely isolated by the power loop and will be reverse-coupled to the power input side through the power loop, thereby causing the input current to exhibit periodic fluctuation distortion at the same frequency, reducing the stability and efficiency of the system. The specific explanation is as follows: (1) If the integral of the voltage across the inductor over time is zero within one cycle, then the volt-second balance equation can be obtained: V in ×t on =V out×t off , where V in V is the input voltage. out For the output voltage, t on t is the on-time of the switching transistor. off Let V be the turn-off time of the switching transistor. Substituting the switching period T and duty cycle D into the above volt-second balance equation, we get: V in ×D×T=V out ×(1-D)×T, after simplification, the voltage conversion relationship is: V out =V in ×[D / (1-D)], where T=t on +t off The switching period is defined as D = t. on / T.

[0036] (2) In an AC input system, full-wave rectification will generate a ripple of twice the power frequency. Specifically, when the power frequency is 50Hz, the ripple is 100Hz; when the power frequency is 60Hz, the ripple is 120Hz. Therefore, the input voltage can be expressed as: V in =V dc_in +V ac_in ×sin(2π×f2×t), where V dc_in For the input DC component, V ac_in f2 is the input ripple amplitude, and f2 is twice the power frequency.

[0037] (3) Step 1: Input voltage V in Substituting the voltage conversion relationship above, we get: V out =[V dc_in +V ac_in ×sin(2π×f2×t)]×[D / (1-D)], expanding this equation gives: V out =V dc_in ×[D / (1-D)]+V ac_in ×sin(2π×f2×t)×[D / (1-D)], where V dc_in ×[D / (1-D)]=V dc_out V ac_in ×[D / (1-D)]=V ac_out V dc_out This is the DC component of the output voltage. Therefore, we can obtain: V out =V dc_out +V ac_out ×sin(2π×f2×t), which indicates the ripple V of the output voltage. ac_out With the input voltage ripple V ac_in It is directly proportional, with a proportionality constant of D / (1-D).

[0038] Step 2: Under steady state, and assuming losses are negligible, the input power equals the output power, therefore: V in ×I in =V out ×I out Therefore, we can obtain: I in =(V out / V in )×I out Assuming the output current of the DC-DC switching converter circuit 110 contains both a DC component and a ripple component, we can obtain: I out =I dc_out +I ac_out ×sin(2π×f2×t), where I ac_out This represents the ripple amplitude of the output current.

[0039] Step 3: Convert the input voltage V in (2) in V in step 1 out and I in step 2 out Substitute I in From the expression, we can obtain: . Will I in The denominator of the expression is represented as: V dc_in ×[1+(V ac_in / V dc_in )×sin(2π×f2×t)]. Based on V ac_in V dc_in and V ac_out V dc_out The approximate condition (ripple much smaller than DC component) can be obtained using Taylor expansion: 1 / (1+X)≈1-X (when |X|<1), which gives: 1 / [V dc_in +V ac_in [×sin(2π×f2×t)]≈1 / V dc_in ×[1-(V ac_in / V dc_in Substituting this formula into I... in The expression yields: .

[0040] Step 4: In actual circuits, the input ripple Vac_out is usually much smaller than the DC component V. dc_in , that is (V ac_in / V dc_in ) 1. Therefore, I in the above formula in It can ultimately be simplified to: I in ≈(V dc_out / V dc_in )×Idc_out +(V dc_out / V dc_in )×I ac_out ×sin(2π×f2×t), which indicates that the DC component of the input current of the DC-DC switching converter circuit 110 is (V dc_out / V dc_in )×I dc_out The ripple component of the input current is (V dc_out / V dc_in )×I ac_out ×sin(2π×f2×t). This is because the ripple of the input current is proportional to the ripple of the output current, and they have the same frequency (twice the power frequency). The twice-power frequency ripple at the output port is directly transmitted to the input current through the proportional relationship between the input current and the output current.

[0041] In summary, since the input current is proportional to the output current, the ripple of the output current is directly transmitted to the input current. This causes the double power frequency ripple at the output of the DC-DC switching converter circuit 110 (i.e., the double power frequency ripple superimposed on the voltage of the battery 130) to affect the input current, thereby causing the input current to exhibit periodic fluctuation distortion at the same frequency, which reduces the stability and efficiency of the system.

[0042] To suppress the impact of double-frequency ripple on the input current, this application further provides a switching control method. This switching control method controls the switching transistor in a DC-DC switching converter circuit electrically connected between the input power supply and the battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with double-frequency ripple. In some embodiments, the DC-DC switching converter circuit, the input power supply, and the battery can be connected via... Figures 1-2 The structure shown is implemented in detail in the above embodiments, and will not be repeated here.

[0043] like Figure 4 As shown, the switch control method includes the following steps S410 to S460.

[0044] Step S410: Determine the operating mode of the DC-DC switching converter circuit, wherein the operating mode includes continuous conduction mode and intermittent conduction mode.

[0045] Among them, Continuous Conduction Mode (CCM) refers to the state in which the inductor current is continuous throughout a complete switching cycle of a DC-DC switching converter circuit, and its minimum value is greater than 0. When the next switching cycle arrives, the inductor still maintains a current state.

[0046] Discontinuous Conduction Mode (DCM) refers to the idle phase in a DC-DC switching converter circuit where the inductor current drops to 0 midway through a complete switching cycle and remains at zero for a period of time until the next switching cycle begins storing energy again.

[0047] By automatically identifying the operating mode of the DC-DC switching converter circuit, it helps to select the appropriate feedforward compensation coefficient and different voltage ripple types, ensuring that output ripple can be effectively suppressed under different load conditions.

[0048] In some embodiments, the specific implementation process of step S410 includes the following steps S510 to S530.

[0049] Step S510: Determine the critical output current corresponding to each topology type.

[0050] Specifically, for a DC-DC switching converter circuit with a defined topology, under the conditions of fixed input voltage, fixed switching frequency, and rated duty cycle, the average output DC current corresponding to the moment when the inductor current drops to zero at the very beginning of the next switching cycle is called the critical output current of the topology under that operating condition.

[0051] In some embodiments, the specific implementation process of step S510 includes the following steps S610 to S630.

[0052] Step S610: When the topology type is buck, the critical output current is V. out ×(V in -V out ) / (2L×f1×V in ), where V out V is the output voltage. in Where is the input voltage of the input power supply, L is the inductance value of the inductor in the DC-DC switching converter circuit, and f1 is the switching frequency of the switching transistor.

[0053] Step S620: When the topology type is boost type, the critical output current is V. in ×(V out -V in ) / (2L×f1×V out ).

[0054] Step S630: When the topology type is buck-boost type, the critical output current is V. out ×V in / [2L×f1×(V in +V out )).

[0055] Step S520: When the output current of the DC-DC switching converter circuit is greater than the critical output current, the operating mode is continuous conduction mode.

[0056] The actual average output current of the DC-DC switching converter circuit is greater than the critical output current, which means that the instantaneous current of the inductor always remains greater than zero within a complete switching cycle. The inductor current will not drop to zero. The circuit will immediately enter the energy storage freewheeling state of the next cycle at the end of the cycle, with no current idle interval. This working state is defined as continuous conduction mode.

[0057] Step S530: When the output current is less than or equal to the critical output current, the operating mode is intermittent conduction mode.

[0058] The actual average output current of the DC-DC switching converter circuit is less than or equal to the critical output current. This means that within a complete switching cycle, the instantaneous inductor current has dropped to zero before the end of the cycle. The inductor experiences a period of idle freewheeling with zero current until the next switching cycle restarts the inductor's energy storage. This operating state is defined as the intermittent conduction mode.

[0059] Step S420: Determine the duty cycle of the DC-DC switching converter circuit according to its topology and operating mode, and calculate the feedforward compensation coefficient. The topology includes buck, boost, and buck-boost types.

[0060] In this way, the solution proposed in this application can be applied to a variety of topologies and has strong versatility.

[0061] In some embodiments, the feedforward compensation coefficient in step S420 is the partial derivative of the duty cycle of the DC-DC switching converter circuit with respect to the output voltage of the DC-DC switching converter circuit, wherein the partial derivative is obtained from a preset mapping relationship according to the topology type and operating mode.

[0062] Specifically, in a DC-DC switching converter circuit, the first-order partial derivative of the circuit duty cycle with respect to the output voltage is the feedforward compensation coefficient. Furthermore, this partial derivative is directly determined by referencing a pre-established parameter mapping relationship based on the circuit topology and the actual operating mode.

[0063] In some embodiments, the specific implementation process of determining the feedforward compensation coefficient includes the following steps S710 to S760.

[0064] Step S710: When the operating mode is continuous conduction mode and the topology type is buck, the duty cycle is: V out / V in The feedforward compensation coefficient is: 1 / V in , where V out V is the output voltage.in This refers to the input voltage of the input power supply.

[0065] Step S720: When the operating mode is continuous conduction mode and the topology type is boost type, the duty cycle is: 1-V in / V out The feedforward compensation coefficient is: V in / V out 2 .

[0066] Step S730: When the operating mode is continuous conduction mode and the topology type is buck-boost type, the duty cycle is: V out / (V in +V out The feedforward compensation coefficient is: V in / (V in +V out ) 2 .

[0067] Step S740: When the operating mode is intermittent conduction mode and the topology type is buck, the duty cycle is: The feedforward compensation coefficient is: [L×f1×I] out ×(V in +V out )] / [D×(V out -V in ) 3 ], where L is the inductance value of the inductor in the DC-DC switching converter circuit, and f1 is the switching transistor (i.e., Figure 2 The switching frequency of the first switching transistor Q1 in the circuit, I out V is the output current of the DC-DC switching converter circuit. out V is the output voltage. in denoted as , where is the input voltage of the input power supply, and D is the duty cycle.

[0068] Specifically, when the switching transistor is turned on and the inductor current rises, the inductor current i pk =(V in -V out )×D×T / L, according to the volt-second balance, we can get: (V in -V out )×D×T=V out ×D'×T, then D'=(V in -V out )×D / V out Output current I out =1 / 2×i pk ×D'=1 / 2×(V in -V out ) 2 ×D2 ×T / (L×V out ), then, D 2 =2L×f1×I out ×V out / (V in -V out ) 2 .thereby, V in L, f1 and I out Here, all are considered constants. Therefore, the partial derivative of the output voltage (i.e., the feedforward compensation coefficient) is [L×f1×I]. out ×(V in +V out )] / [D×(V out -V in ) 3 ].

[0069] Step S750: When the operating mode is intermittent conduction mode and the topology type is boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×V in 2 ).

[0070] Specifically, when the switching transistor is turned on and the inductor current rises, the inductor current i pk =V in ×D×T / L, according to the volt-second balance, V in ×D×T=(V out -V in If ) × D' × T, then D' = V in ×D / (V out -V in Output current I out =1 / 2×i pk ×D'=1 / 2×(V in ×D×T / L)×[V in ×D / (V out -V in )], then, D 2 =2L×f1×I out ×(V out -V in ) / V in 2 .thereby, V in L, f1 and I out Here, all are considered constants. Therefore, the partial derivative of the output voltage (i.e., the feedforward compensation coefficient) is (L×f1×I). out ) / (D×V in2 ).

[0071] Step S760: When the operating mode is intermittent conduction mode and the topology type is buck-boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×V in 2 ).

[0072] Specifically, when the switching transistor is turned on and the inductor current rises, the inductor current i pk =V in ×D×T / L, according to the volt-second balance, V in ×D×T=V out ×D'×T, then D'=V in ×D / V out Output current I out =1 / 2×i pk ×D'=1 / 2×(V in ×D×T / L)×(V in ×D / V out ), then, D 2 =2L×f1×I out ×V out / V in 2 .thereby, V in L, f1 and I out Here, all are considered constants. Therefore, the partial derivative of the output voltage (i.e., the feedforward compensation coefficient) is (L×f1×I). out ) / (D×V in 2 ).

[0073] In summary, precise partial derivative expressions for each topology in both continuous and intermittent conduction modes are provided, which helps to improve compensation accuracy.

[0074] Step S430: Sample the output voltage of the DC-DC switching converter circuit and determine the ripple component in the output voltage.

[0075] Specifically, by executing step S430, it is beneficial to enable subsequent feedforward compensation to be generated based on real-time detected output ripple, which has a fast response speed and can effectively suppress low-frequency disturbances.

[0076] In some embodiments, the specific implementation process of step S430 includes the following step S810.

[0077] Step S810: The ripple component is the deviation of the output voltage from the reference voltage, wherein the reference voltage is obtained based on the ripple characteristics of the output voltage.

[0078] That is, ripple component = output voltage - reference voltage. The reference voltage is not a fixed constant voltage value, but is determined based on the actual ripple characteristics of the output voltage itself, such as ripple amplitude and fluctuation frequency.

[0079] In some embodiments, the specific implementation process of obtaining the reference voltage in step S810 based on the ripple characteristics of the output voltage includes the following steps S910 to S930.

[0080] Step S910: Perform sinusoidal fitting of the output voltage to the second harmonic of the power frequency to obtain the fitting result.

[0081] Step S920: Based on the fitting results, determine the fitting quality index.

[0082] Step S930: When the fitting quality index is greater than the preset threshold, the root mean square value of the output voltage is used as the reference voltage; otherwise, the average value of the output voltage is used as the reference voltage.

[0083] Specifically, first, construct the matrix: [sin(2π×f2×t)] s cos(2π×f2×t) s ]], where n is the sampling point number, n=0, 1, 2, ...; f2 is the inverter power frequency; t s =n×T s =n / f s For the time of the nth sampling point, T s =1 / f s The sampling interval is sin(2π×f²×n / t). s ) represents the sinusoidal component based on twice the power frequency, cos(2π×f²×t) s () represents the cosine component based on twice the power frequency.

[0084] The first fitting weight coefficient (denoted as B) and the second fitting weight coefficient (denoted as C) are solved using the least squares method.

[0085] The amplitude (denoted as A) and phase (denoted as φ) of the double power frequency ripple are synthesized using the first fitting weight coefficient B and the second fitting weight coefficient C. , Based on the obtained amplitude A and phase φ, the fitting result Y[n] is calculated as follows: Y[n] = A × sin(2π × f² × t) s +φ).

[0086] Determine the fit quality index R 2 for: , where x[n] is the discretely sampled ripple signal sequence, that is, x[n] is the ripple voltage value collected at the nth sampling time; This represents the average value of the ripple signal sequence. : Characterizes the difference between the original ripple component and the average ripple value, which represents the overall fluctuation level of the ripple; : Characterizes the difference between the original ripple component and the fitted ripple.

[0087] Fit quality index R 2 R is used to measure how well a model fits the original data. 2 The closer R is to 1, the higher the explanatory power of the fitted model for the ripple, and the closer the ripple is to a sine wave at twice the power frequency; 2 =1 indicates that the model perfectly fits the data, and the residual is 0; R 2 =0 indicates that the model fit is as bad as using the average value directly, meaning there is almost no ripple in the data.

[0088] When the fitting quality index R 2 When the ripple exceeds a preset threshold, it indicates that the ripple is almost a pure double-frequency sine wave. In this case, using the root mean square (RMS) value as a reference voltage can more accurately reflect the impact of the ripple on the effective value of the battery voltage, improving calculation accuracy. The RMS value is a characteristic quantity obtained by first squaring all sampled values ​​in the discrete sampled signal, then calculating the arithmetic mean, and finally taking the square root. The RMS value represents the effective value of the signal, reflecting the true impact of AC ripple on system power and energy.

[0089] When the fitting quality index R 2 When the ripple is less than or equal to a preset threshold, it indicates that the ripple is not a simple double-frequency sine wave and may contain noise or higher harmonics. In this case, using the average value is more robust and avoids bias caused by an incorrect ripple model. The average value is the arithmetic mean of the signal over the entire sampling window, calculated by summing all sampled values ​​and dividing by the number of sampling points. The average value represents the DC component of the signal, which is the stable value after the ripple has been averaged out.

[0090] Step S440: Determine the compensation amount based on the product of the feedforward compensation coefficient and the ripple component.

[0091] Step S450: Determine the target duty cycle based on the sum of the duty cycle and the compensation amount.

[0092] Step S460: Generate a pulse width modulation signal based on the target duty cycle to control the switching transistor in the DC-DC switching converter circuit.

[0093] Specifically, the partial derivative of the duty cycle with respect to the output voltage is calculated, and this partial derivative is used as the feedforward compensation coefficient of the duty cycle relative to the output voltage. That is, the feedforward compensation coefficient characterizes the amount by which the duty cycle needs to change when the output voltage changes by one unit, reflecting how much duty cycle needs to be adjusted to correct a unit output voltage deviation. Simultaneously, the ripple component separated from the output voltage is extracted. The feedforward compensation coefficient is multiplied by the ripple component to obtain the compensation amount, which is used to determine how much duty cycle needs to be corrected to cancel the ripple. The current duty cycle is algebraically summed with the above compensation amount to obtain the target duty cycle after superimposed ripple correction. The target duty cycle includes both the reference duty cycle required for steady-state voltage regulation and the dynamic correction amount required to suppress output ripple. According to a preset switching cycle, using the target duty cycle as the modulation reference, a pulse width modulation signal corresponding to the on-time is generated. This pulse width modulation signal is sent to the control terminals of each switch in the DC-DC switching converter circuit to adjust the on and off times of each switch.

[0094] In this way, by adding feedforward compensation to the duty cycle, the double-frequency ripple is suppressed, which helps improve the stability of the input current of the input power supply, thereby improving the stability and efficiency of the system, including the DC-DC switching converter circuit. Furthermore, the feedforward compensation is an open-loop structure, which does not affect system stability and forms a good match with feedback control, resulting in better system stability.

[0095] Figure 5 An exemplary diagram of the input current when the switching control method of this application is not used is shown, wherein the horizontal axis is time and the vertical axis is current, and the input current at this time is waveform LI1; Figure 6 An exemplary schematic diagram of the input current when using the switching control method of this application is shown, wherein the horizontal axis represents time and the vertical axis represents current, and the input current at this time is waveform LI2.

[0096] like Figure 5 As shown, without the switching control method of this application, the maximum current is approximately 9.74A and the minimum current is approximately 9.07A, as can be seen from waveform LI1. Therefore, the ripple is approximately 0.67A, which is quite noticeable. Figure 6 As shown, when using the switching control method of this application, the maximum current value is approximately 9.45A and the minimum current value is approximately 9.37A, as can be obtained from waveform LI2. Therefore, the ripple is approximately 0.08A, indicating that the ripple is significantly suppressed. It is evident that the switching control method provided by this application can effectively suppress double-frequency ripple, thereby improving the stability of the input current and enhancing the stability and efficiency of the system including the DC-DC switching converter circuit.

[0097] In some embodiments, the switch control method further includes steps S1010 to S1030.

[0098] Step S1010: When controlling the first and second switching transistors based on the pulse width modulation signal, and controlling the third switching transistor to remain on, and controlling the fourth switching transistor to remain off, determine that the topology of the DC-DC switching converter circuit is a buck type.

[0099] Step S1020: When controlling the third and fourth switching transistors based on the pulse width modulation signal, and controlling the first switching transistor to remain on, and controlling the second switching transistor to remain off, determine that the topology of the DC-DC switching converter circuit is a boost type.

[0100] Step S1030: When controlling the first, second, third, and fourth switching transistors based on the pulse width modulation signal, determine that the topology of the DC-DC switching converter circuit is a buck-boost type.

[0101] It is understood that the method for determining the topology type of the DC-DC switching converter circuit and the resulting beneficial effects in the method embodiments can be referred to the corresponding descriptions in the above embodiments of the DC-DC switching converter circuit, which will not be repeated here for the sake of brevity.

[0102] The above description is merely an embodiment of this application and does not limit the patent scope of this application. Any equivalent structural or procedural transformations made using the content of this application's specification and drawings, or direct or indirect applications in other related technical fields, are similarly included within the patent protection scope of this application.

[0103] The above embodiments are only used to illustrate the technical solutions of this application, and are not intended to limit them. Within the framework of this application, the technical features of the above embodiments or different embodiments can also be combined, and the steps can be implemented in any order. Those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features; and these modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope of the technical solutions of the embodiments of this application.

Claims

1. A switching control method, characterized in that, This method is used to control the switching transistor in a DC-DC switching converter circuit, which is electrically connected between an input power supply and a battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with twice the power frequency ripple. The switching control method includes: The operating mode of the DC-DC switching converter circuit is determined, wherein the operating mode includes continuous conduction mode and intermittent conduction mode; Based on the topology of the DC-DC switching converter circuit and the operating mode, the duty cycle of the DC-DC switching converter circuit is determined, and the feedforward compensation coefficient is calculated. The feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter circuit with respect to the output voltage of the DC-DC switching converter circuit. The topology includes buck, boost, and buck-boost types, and the partial derivative is obtained from a preset mapping relationship based on the topology and the operating mode. Sample the output voltage of the DC-DC switching converter circuit and determine the ripple component in the output voltage; The compensation amount is determined by multiplying the feedforward compensation coefficient by the ripple component. The target duty cycle is determined based on the sum of the duty cycle and the compensation amount; A pulse width modulation signal is generated based on the target duty cycle to control the switching transistors in the DC-DC switching converter circuit; The feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter circuit with respect to the output voltage of the DC-DC switching converter circuit, including: When the operating mode is intermittent conduction mode and the topology type is buck type, the duty cycle is: The feedforward compensation coefficient is: [L×f1×I out ×(V in +V out )] / [D×(V out -V in ) 3 Where L is the inductance value of the inductor in the DC-DC switching converter circuit, f1 is the switching frequency of the switching transistor, and I out V is the output current of the DC-DC switching converter circuit. out V is the output voltage. in Where is the input voltage of the input power supply, and D is the duty cycle; When the operating mode is intermittent conduction mode and the topology type is boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×V in 2 ); When the operating mode is intermittent conduction mode and the topology type is buck-boost type, the duty cycle is: The feedforward compensation coefficient is: (L×f1×I) out ) / (D×V in 2 ).

2. The switching control method according to claim 1, characterized in that, The feedforward compensation coefficient is the partial derivative of the duty cycle of the DC-DC switching converter circuit with respect to the output voltage of the DC-DC switching converter circuit, including: When the operating mode is continuous conduction mode and the topology type is buck type, the duty cycle is: V out / V in The feedforward compensation coefficient is: 1 / V in , where V out V is the output voltage. in The input voltage of the input power supply; When the operating mode is continuous conduction mode and the topology type is boost type, the duty cycle is: 1-V. in / V out The feedforward compensation coefficient is: V in / V out 2 ; When the operating mode is continuous conduction mode and the topology type is buck-boost type, the duty cycle is: V out / (V in +V out The feedforward compensation coefficient is: V in / (V in +V out ) 2 .

3. The switching control method according to claim 1, characterized in that, Determining the operating mode of the DC-DC switching converter circuit includes: Determine the critical output current corresponding to each of the aforementioned topology types; When the output current of the DC-DC switching converter circuit is greater than the critical output current, the operating mode is the continuous conduction mode; When the output current is less than or equal to the critical output current, the operating mode is the intermittent conduction mode.

4. The switch control method according to claim 3, characterized in that, Determining the critical output current corresponding to each of the aforementioned topology types includes: When the topology is the buck type, the critical output current is V. out ×(V in -V out ) / (2L×f1×V in ), where V out V is the output voltage. in The input voltage of the input power supply is L, the inductance value of the inductor in the DC-DC switching converter circuit is L, and the switching frequency of the switching transistor is f1. When the topology is the boost type, the critical output current is V. in ×(V out -V in ) / (2L×f1×V out ); When the topology type is the buck-boost type, the critical output current is V. out ×V in / [2L×f1×(V in +V out )).

5. The switching control method according to claim 1, characterized in that, Determining the ripple component in the output voltage includes: The ripple component is the deviation of the output voltage from the reference voltage, wherein the reference voltage is obtained based on the ripple characteristics of the output voltage.

6. The switching control method according to claim 5, characterized in that, The reference voltage is obtained based on the ripple characteristics of the output voltage, including: The output voltage is fitted with a sine wave corresponding to the second harmonic of the power frequency to obtain the fitting result; Based on the fitting results, the fitting quality index is determined; When the fitting quality index is greater than a preset threshold, the root mean square value of the output voltage is used as the reference voltage; otherwise, the average value of the output voltage is used as the reference voltage.

7. An energy storage system, characterized in that, include: The system comprises a DC-DC switching converter circuit, an input power supply, and a battery. The DC-DC switching converter circuit is electrically connected between the input power supply and the battery. When the input power supply charges the battery through the DC-DC switching converter circuit, the battery voltage is superimposed with twice the power frequency ripple. And a control processing unit, which is electrically connected to the DC-DC switching converter circuit to control the switching transistors in the DC-DC switching converter circuit; The control processing unit includes: at least one processor and a memory; the memory is coupled to the processor and is used to store instructions or programs, which, when executed by the at least one processor, cause the at least one processor to perform the switching control method as described in any one of claims 1-6.

8. The energy storage system according to claim 7, characterized in that, The DC-DC switching converter circuit includes a first switching transistor, a second switching transistor, a third switching transistor, a fourth switching transistor, and an inductor; The first switch and the second switch are connected in series between the positive and negative terminals of the input power supply. The connection point between the first switch and the second switch is electrically connected to the first terminal of the inductor. The third switch and the fourth switch are connected in series between the positive and negative terminals of the battery. The connection point between the third switch and the fourth switch is electrically connected to the second terminal of the inductor. The control processing unit is electrically connected to the first switch, the second switch, the third switch, and the fourth switch, respectively.

Images