Motor drive system and stress optimization method for ZVS series winding inverter

Abstract

The application discloses a motor driving system and a ZVS series winding inverter stress optimization method, relates to the technical field of motor driving, and solves the technical problem of uneven current stress of middle bridge arms and two side bridge arms, which influences inverter efficiency. The system comprises a current stress optimization module, a series winding inverter system and a series winding motor which are sequentially connected. The series winding inverter system comprises a series winding inverter and a coupling inductor. The series winding inverter comprises eight parallel bridge arms. Each bridge arm comprises two parallel switch tubes. Two adjacent bridge arms are connected with a coupling inductor. The current stress optimization module generates a switch driving signal meeting the timing requirements of the series winding inverter system through the current stress optimization method. The switch driving signal adjusts the voltage output of the series winding inverter and drives the series winding motor. The application reduces the current stress and improves the system efficiency while ensuring that all switch tubes realize zero voltage switching.

Description

Technical Field

[0001] This invention relates to the field of motor drive technology, and in particular to a motor drive system and a stress optimization method for ZVS series winding inverters. Background Technology

[0002] Traditional motor drive systems often employ hard switching (in power electronic converters, the switching transistor turns on or off under the simultaneous presence of voltage and current), resulting in significant switching losses. The rapid development of wide-bandgap semiconductor devices such as silicon carbide (SiC) and gallium nitride (GaN) has theoretically increased their switching frequencies far beyond those of traditional silicon-based devices, paving the way for higher frequency motor drives. Although third-generation semiconductors can effectively reduce energy loss during a single switching cycle, overall switching losses still increase dramatically when the system switching frequency rises to hundreds of kHz, leading to a significant decrease in inverter efficiency. Traditional control methods based on hard switching remain a key factor limiting further improvements in system efficiency and frequency.

[0003] Against this backdrop, zero-voltage switching (ZVS, a power electronic soft-switching technology that reduces switching losses, improves power conversion efficiency, and reduces electromagnetic interference by reducing the voltage across the switching transistor to zero or near zero before it is turned on) technology is particularly suitable for wide-bandgap semiconductor devices such as SiCMOS transistors because it can effectively improve switching frequency and system efficiency.

[0004] However, in topologies such as series-wound inverters, the current stress in the middle arm (the current surge and rate of change that power electronic devices experience during operation, a key factor affecting device reliability and lifespan) can be up to 1.73 times that of the two side arms. Using traditional space vector pulse width modulation (SVPWM) strategies will further increase the current stress, leading to reduced power capacity utilization. Discontinuous pulse width modulation (DPWM) can alleviate current stress and improve performance under some operating conditions, but it still struggles to achieve global optimization across the entire operating range. Therefore, a modulation algorithm based on dynamic zero-sequence injection is urgently needed to simultaneously achieve soft-switching technology in series-wound topologies, optimize the magnitude of current stress, and comprehensively improve system efficiency.

[0005] In the process of realizing this invention, the inventors discovered at least the following problems in the prior art: In a series-winding inverter topology, the current stress in the middle arm is unbalanced with that in the two side arms, which affects the inverter's efficiency. Summary of the Invention

[0006] The purpose of this invention is to provide a motor drive system and a stress optimization method for a ZVS series winding inverter, to solve the technical problem in existing series winding inverter topologies where the current stress is unbalanced between the middle arm and the two side arms, affecting inverter performance. The various technical effects of the preferred solutions among the many technical solutions provided by this invention are detailed below.

[0007] To achieve the above objectives, the present invention provides the following technical solution: This invention provides a motor drive system, comprising a current stress optimization module, a series winding inverter system, and a series winding motor connected in sequence. The series winding inverter system includes a series winding inverter and a coupling inductor. The series winding inverter includes eight parallel bridge arms, each bridge arm including two parallel switching transistors. The number of coupling inductors is four, with two adjacent bridge arms connected to one coupling inductor. The current stress optimization module generates a switching drive signal that meets the timing requirements of the series winding inverter system using a current stress optimization method. The switching drive signal adjusts the voltage output of the series winding inverter by controlling the on / off state of the switching transistors of the series winding inverter, thereby driving the series winding motor.

[0008] Preferably, the bridge arm includes a first bridge arm, a second bridge arm, a third bridge arm, a fourth bridge arm, a fifth bridge arm, a sixth bridge arm, a seventh bridge arm, and an eighth bridge arm; the first bridge arm and the second bridge arm form a first group of bridge arms, the third bridge arm and the fourth bridge arm form a second group of bridge arms, the fifth bridge arm and the sixth bridge arm form a third group of bridge arms, and the seventh bridge arm and the eighth bridge arm form a fourth group of bridge arms; wherein, the first bridge arm includes a first switching transistor. T a1 Second switching transistor T a2 The second bridge arm includes a third switching transistor. T b1 Fourth switching transistor T b2 The third bridge arm includes a fifth switching transistor. T c1 Sixth switch tube T c2 The fourth bridge arm includes a seventh switching transistor. T d1 Eighth switch tube T d2 The fifth bridge arm includes a ninth switching transistor. T a3 10th switch tube T a4 The sixth bridge arm includes an eleventh switching transistor. Tb3 12th switch tube T b4 The seventh bridge arm includes a thirteenth switching transistor. T c3 Fourteenth switch tube T c4 The eighth bridge arm includes a fifteenth switching transistor. T d3 Sixteenth switch tube T d4 The coupling inductors include a first coupling inductor, a second coupling inductor, a third coupling inductor, and a fourth coupling inductor; the series-wound motor includes a first stator winding A, a second stator winding B, and a third stator winding C; the first switching transistor... T a1 emitter and second switch T a2 The collector of the transistor and the corresponding terminal of the first coupling inductor are both connected, and the ninth switching transistor is connected. T a3 The emitter and the tenth switch T a4 The collector of the first coupling inductor and the non-identical terminals of the second coupling inductor are both connected; the third switching transistor T b1 emitter and fourth switch T b2 The collector of the first transistor and the corresponding terminal of the second coupling inductor are both connected, and the eleventh switching transistor is connected. T b3 The emitter and the twelfth switch T b4 The collector of the fifth switching transistor and the non-identical terminals of the second coupling inductor are both connected; T c1 The emitter and the sixth switch T c2 The collector of the transistor and the corresponding terminal of the third coupling inductor are both connected, and the thirteenth switching transistor is connected. T c3 The emitter and the fourteenth switch T c4 The collector and the non-identical terminals of the third coupling inductor are both connected; the seventh switch transistor T d1 The emitter and the eighth switch T d2 The collector of the transistor and the corresponding terminal of the fourth coupling inductor are both connected, and the fifteenth switching transistor is connected. T d3 The emitter and the sixteenth switch T d4The collector of the first winding and the non-same-name terminals of the fourth coupling inductor are all connected; the two ends of the first winding A are respectively connected to the first coupling inductor and the second coupling inductor; the two ends of the second winding B are respectively connected to the second coupling inductor and the third coupling inductor; the two ends of the third winding C are respectively connected to the third coupling inductor and the fourth coupling inductor.

[0009] Preferably, the motor drive system further includes a DC bus capacitor and a DC power supply; the first plate and the first switching transistor of the DC bus capacitor. T a1 collector, third switching transistor T b1 collector, fifth switch T c1 collector, seventh switch T d1 collector, ninth switch T a3 collector, eleventh switching transistor T b3 collector, thirteenth switching transistor T c3 collector and fifteenth switching transistor T d3 The collectors of all components are connected to the positive terminal of the DC power supply; the second plate of the DC bus capacitor and the second switching transistor... T a2 emitter, fourth switch T b2 emitter, sixth switch T c2 emitter, eighth switch T d2 emitter, tenth switch T a4 The emitter and the twelfth switch T b4 emitter, fourteenth switch T c4 The emitter and the sixteenth switch T d4 The emitters of all components are connected to the negative terminal of the DC power supply.

[0010] Preferably, the motor drive system further includes a current sampling module, a position encoding module, and a speed calculation module. T 1 Coordinate transformation unit, speed PI control module, d Axis PI control module q Axis PI control module T 2 and T3The system includes a coordinate transformation unit, a first difference calculation unit, a second difference calculation unit, and a third difference calculation unit. The input terminal of the current sampling module is connected between the series winding inverter system and the series winding AC motor. The input terminal of the position encoding module is connected to the series winding AC motor, and its output terminal is connected to the speed calculation module. T 1 Coordinate transformation unit T 2 and T 3 The coordinate transformation unit is connected, and the output of the speed calculation module is connected to the input of the speed PI control module through the first difference calculation unit. The output of the speed PI control module is connected to the input of the speed PI control module through the second difference calculation unit. q The input terminal of the axis PI control module is connected, the d Axis PI control module q The output terminals of the axis PI control module are all connected to the T 2 and T 3 The input terminal of the coordinate transformation unit is connected, the T 2 and T 3 The output terminals of the coordinate transformation unit and the current sampling module are both connected to the current stress optimization module. d The input terminal of the axis PI control module is connected to the third difference calculation unit, and the third difference calculation unit is connected to the output terminal of the T1 coordinate transformation unit.

[0011] A stress optimization method for a ZVS series winding inverter, operating through the above motor drive system, includes the following steps: S100: Transforming the voltage of the series winding AC motor in the rotating coordinate system into a three-phase reference voltage in the natural coordinate system through coordinate transformation. V a , V b , V c S200: Three-phase reference voltage for AC motors with series windings V a , V b , V c Four sets of bridge arm voltages were used to calculate the modulation waveforms of the series winding inverter under three different zero-sequence voltage injection scenarios; S300: A current ripple prediction model was established to calculate the maximum bridge arm current; S400: Based on the maximum bridge arm current, the modulation waveform was obtained. d x0 , dxmax and d xmin Calculate the soft-switching constraints and the magnitude of the four-arm current and the optimal zero-sequence voltage under this clamping state. V cm and the operating frequency of the switching transistor f s S500: Based on the operating frequency of the switching transistor. f s The reconstructed reference voltage after dynamic zero-sequence voltage injection is obtained. The reconstructed reference voltage is compared with the interleaved carrier to generate the switching signal of the switching tube.

[0012] Preferably, step S100 includes: S110: Obtain the current position angle of the series-wound AC motor through the position encoding module. θ Position angle θ The input speed calculation module obtains the actual speed of the series-wound AC motor. N , give the speed N ref Compared with actual speed N The difference is calculated by PI control via the speed PI control module to obtain... q First given current of shaft I q_ref ; S120: Obtains the three-phase current output of the series-wound AC motor through the current sampling module. i a , i b , i c Through the first coordinate transformation unit T 1 get q First actual current of shaft i q and d First actual current of shaft i d Set the d-axis reference current. I d_ref =0 and d First actual current of shaft i d To make differences, through d The axis PI control module controls the rotation to obtain the coordinate system. d Shaft given voltage V d ;Will q First given current of shaft I q_ref and q First actual current of shaft iq Take the difference, and then pass the difference through... q The axis PI control module performs PI control to obtain the coordinates in the rotating coordinate system. q Shaft given voltage V q ; in, T 1 The transformation formula for the coordinate transformation unit is: ; S130: The given voltage in the rotating coordinate system... V d and V q conduct T 2 Coordinate transformation and T 3 Coordinate transformation yields the three-phase reference voltage of the series-wound AC motor in the natural coordinate system. V a , V b , V c , V a , V b , V c These are the voltages of the first stator winding A, the second stator winding B, and the third stator winding C, respectively. in, T 2 The formula for coordinate transformation is: ; T 3 The formula for coordinate transformation is: .

[0013] Preferably, step S200 specifically includes: S210: The three-phase reference voltage of the series-wound AC motor. V a , V b , V c Represented as: , S220: Based on topological symmetry and voltage balance principles, the three-phase reference voltage of the series-wound AC motor is represented by four sets of bridge arm reference voltages: ; The four-phase modulated wave signals generated by the four sets of bridge arm reference voltages are obtained: , in, m a 、m b , m c , m d These represent the four-phase modulated wave signals generated by the reference voltages of the first, second, third, and fourth bridge arms, respectively. V leg1 , V leg2 , V leg3 , V leg4 These represent the reference voltages for the first, second, third, and fourth bridge arms, respectively. m Represents the modulation coefficient. w This indicates the electrical angular frequency of an AC motor with series windings. S230: For the initial bridge arm reference voltage, three zero-sequence voltage injection methods are set: no injection, maximum injection, and minimum injection. These correspond to three clamping modes: bridge arm non-clamping, bridge arm maximum clamping, and bridge arm minimum clamping, respectively. The calculation formula is as follows: , in x = a, b, c, d max (m x ) The maximum voltage amplitude in the bridge arm reference voltage, min (m x ) This is the minimum voltage amplitude of the bridge arm reference voltage; S240: Based on three different zero-sequence voltage injections, the modulation waveform for driving the series winding inverter is obtained as follows: , in, x = a, b, c, d .

[0014] Preferably, in step S300, the series winding inverter A The phase includes the first bridge arm, the fifth bridge arm, and the first coupling inductor, and its specific operation process is as follows: S310: Based on the first switching transistor T a1 and the third switching transistor T a3 The on / off status will be the status within an on / off cycle. A The slope of the phase bridge arm current is expressed as: , in, V dc Indicates the DC bus voltage. di cir / dt Indicates the slope of the current. L This represents the self-inductance of the first coupled inductor. M This represents the mutual inductance of the first coupled inductor. S a1 Indicates the first switching transistor T a1 The switch signal, S a3 Indicates the third switching transistor T a3 The switch signal; S320: List the first switching transistor T a1 Second switching transistor T a2 Ninth switch tube T a3 10th switch tube T a4 All of these can achieve zero-voltage conduction, and the constraints that must be met are: , in, i a1 (t 1 ) For the first bridge arm in t 1 First switching transistor at all times T a1 Bridge arm current at start-up i a1 (t 4 ) For the first bridge arm in t 4 Second switching transistor T a2 Bridge arm current at start-up i a2 (t 2 ) For the fifth bridge arm t 2 Ninth switch at time T a3 Bridge arm current at start-up i a1 (t 3 ) For the fifth bridge arm t3 Tenth switch tube at time T a4 Bridge arm current at start-up i bias The minimum current required to turn the switching transistor on and off to meet the charging and discharging requirements of the output capacitor of the switching transistor. S330: The duration of different switching states is expressed by duty cycle and switching period as follows: , in, d a Representative is the series winding inverter A Phase reference voltage, T s Indicates the switching cycle. T 1 Corresponding switching transistor S a1 Turn-off and switching transistors S a3 Switching from on state to switching transistor S a1 conduction and switching transistors S a3 The time interval between conduction states. T 2 The corresponding switch state remains as a switching transistor. S a1 conduction and switching transistors S a3 The duration of the shutdown; S340: Based on the bridge arm current slope and switching state time, t 1 First switching transistor at all times T a1 When it is opened and t 4 Second switching transistor T a2 The first arm current at the start-up point is expressed as follows: ; S350: Based on the reciprocal relationship between switching frequency fs and switching period Ts, the soft-switching condition is represented by frequency degrees of freedom, and the frequency constraint of phase A is defined. Represented as: ; S360: Calculate the soft-switching frequency constraints for phases B, C, and D of the series-wound inverter, and select the minimum switching frequency as the operating frequency. : ; S370: Series winding inverterA Maximum value of circulating ripple in phase icir a Represented as: , in V dc Bus voltage f s The final selected system switching frequency; S380: Based on the maximum value of circulating ripple icir a AC motor with series winding A Phase current i a The relationship between the maximum value of the bridge arm current and the bridge arm current. Represented as: .

[0015] Preferably, step S400 specifically includes: S410: The maximum peak current of the four phases of the series winding inverter under different zero-sequence voltages is used as the core comparison benchmark, and its expression is defined as: , in ipeak v0 , ipeak vmax , ipeak vmin These represent the reference voltage modulation waveforms when there is no injection, maximum injection, and minimum injection zero-sequence voltage, respectively. d x0 , d xmax and d xmin ; S420: Defines the minimum peak current under three operating conditions: no injection, maximum injection, and minimum injection zero-sequence voltage. peak opt And use it as the threshold for determining the clamping mode selection, that is: ; S430: Obtain the optimal zero-sequence voltage for minimum current stress in the bridge arm. V cm : , like peak opt = ipeak v0 ,choose v 0 The corresponding clamping mode; if peakopt = ipeak vmax or peak opt = ipeak vmin Select respectively v max and v max Corresponding clamping mode; S440: The operating frequency of the switching transistor... f s Represented as: , Operating frequency of the switching transistor f s The switching is calculated based on the dynamic clamping interval. f 0 represent V cm for v 0 The switching frequency required for soft switching of the switching transistor. f max represent V cm for V max The switching frequency required for soft switching of the switching transistor. f min represent V cm for V min The switching frequency required for soft switching of the switching transistor.

[0016] Preferably, the specific operation of step S500 is as follows: the reference voltage reconstructed after dynamic zero-sequence injection is used as the modulation signal, compared with the variable switching frequency interleaved carrier, to generate the final PWM switching signal, and output to the series winding inverter system, so that the three-phase current is output through the series winding motor.

[0017] Implementing one of the above-described technical solutions of the present invention has the following advantages or beneficial effects: This invention ensures zero-voltage switching for all switching transistors while reducing current stress, improving overall system efficiency, and is highly versatile. It effectively reduces conduction losses and switching transistor current stress requirements, significantly improving the efficiency, reliability, and service life of inverter systems, precisely meeting the core requirements of current AC motor drive systems for miniaturization, high power density, and high reliability. Attached Figure Description

[0018] To more clearly illustrate the technical solutions of the embodiments of the present invention, the accompanying drawings used in the description of the embodiments will be briefly introduced below. Obviously, the accompanying drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort. In the drawings: Figure 1 This is a topology diagram of a motor drive system according to an embodiment of the present invention; Figure 2 This is a circuit diagram of a series winding inverter system for a motor drive system according to Embodiment 1 of the present invention; Figure 3 This is a flowchart illustrating the stress optimization method for ZVS series winding inverters according to Embodiment 2 of the present invention. Figure 4 This is a flowchart of the stress optimization method for ZVS series winding inverter according to Embodiment 2 of the present invention; Figure 5 This is a reference voltage waveform diagram generated by three zero-sequence voltage injection methods in the stress optimization method for ZVS series winding inverters in Embodiment 2 of the present invention; Figure 6 This is a comparison of ripple prediction results for three zero-sequence voltage injection methods in the stress optimization method for ZVS series winding inverters in Embodiment 2 of the present invention. Figure 7 This is the modulation wave generated by three zero-sequence voltage injection methods that have the best current stress optimization effect in the stress optimization method of ZVS series winding inverter in Embodiment 2 of the present invention; Figure 8 These are the current stress waveforms of traditional SVPWM, DPWM, and the ZVS series winding inverter stress optimization method of Embodiment 2 of this invention under the same operating conditions. Figure 1 ; Figure 9 These are the current stress waveforms of traditional SVPWM, DPWM, and the ZVS series winding inverter stress optimization method of Embodiment 2 of this invention under the same operating conditions. Figure 2 ; Figure 10 These are the current stress waveforms of traditional SVPWM, DPWM, and the ZVS series winding inverter stress optimization method of Embodiment 2 of this invention under the same operating conditions. Figure 3 ; Figure 11 This is a comparison chart of the efficiency of SVPWM, DPWM and the stress optimization method of ZVS series winding inverter in Embodiment 2 of the present invention under variable speed conditions; Figure 12 This is a comparison chart of the efficiency of SVPWM, DPWM and the stress optimization method of ZVS series winding inverter in Embodiment 2 of the present invention under variable load conditions; Figure 13 This is the current stress scan diagram of the SVPWM algorithm under the full operating range; Figure 14 Current stress scan diagram of DPWM algorithm under full operating conditions; Figure 15 Current stress scan diagram of the stress optimization method for ZVS series winding inverter in Embodiment 2 of the present invention, covering the entire operating range; In the diagram: 1. Series winding inverter system; 2. Series winding motor; 3. DC power supply; 4. DC bus capacitor; 5. Current sampling module; 6. Position encoding module; 7. Speed ​​calculation module; 8. T1 coordinate transformation unit; 9. Speed ​​PI control module; 10. q-axis PI control module; 11. d-axis PI control module; 12. T2 and T3 coordinate transformation units; 13. Current stress optimization module; 14. First difference calculation unit; 15. Second difference calculation unit; 16. Third difference calculation unit. Detailed Implementation

[0019] To make the objectives, technical solutions, and advantages of the present invention clearer, various exemplary embodiments described below will be referenced to the accompanying drawings, which form part of the exemplary embodiments, illustrating various exemplary embodiments that may be used to implement the present invention. Unless otherwise indicated, the same numbers in different drawings represent the same or similar elements. The embodiments described in the following exemplary embodiments do not represent all embodiments consistent with this disclosure. It should be understood that they are merely examples of processes, methods, and apparatuses consistent with some aspects of the present invention disclosed as detailed in the appended claims, and other embodiments may be used, or structural and functional modifications may be made to the embodiments listed herein without departing from the scope and spirit of the present invention.

[0020] In the description of this invention, it should be understood that the terms "center," "longitudinal," "lateral," etc., indicate the orientation or positional relationship based on the accompanying drawings, and are only for the convenience of describing the invention and simplifying the description, and do not indicate or imply that the referred element must have a specific orientation, or be constructed and operated in a specific orientation. The terms "first," "second," etc., are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features. The term "multiple" means two or more. The terms "connected" and "linked" should be interpreted broadly, for example, they can be fixed connections, detachable connections, integral connections, mechanical connections, electrical connections, communication connections, direct connections, indirect connections through an intermediate medium, and can be the internal connection of two elements or the interaction relationship between two elements. The term "and / or" includes any and all combinations of one or more of the related listed items. Those skilled in the art can understand the specific meaning of the above terms in this invention according to the specific circumstances.

[0021] To illustrate the technical solution described in this invention, specific embodiments are described below, showing only the parts related to the embodiments of this invention.

[0022] Example 1: like Figure 1 , Figure 2 As shown, this invention provides a motor drive system, including a current stress optimization module, a series winding inverter system, and a series winding motor connected in sequence. The series winding inverter system includes a series winding inverter and a coupling inductor. The series winding inverter includes eight parallel bridge arms, each bridge arm including two parallel switching transistors. The number of coupling inductors is four, with two adjacent bridge arms connected to one coupling inductor. The current stress optimization module generates switching drive signals that meet the timing requirements of the series winding inverter system through a current stress optimization method. The switching drive signals regulate the voltage output of the series winding inverter by controlling the on / off state of the switching transistors, thereby driving the series winding motor. The current stress optimization module of this system can effectively reduce conduction losses and switching transistor current stress requirements, significantly improving the efficiency, reliability, and service life of the inverter system, and meeting the core requirements of AC motor drive systems for small size, high power density, and high reliability.

[0023] As an optional implementation method, such as Figure 2 As shown, the bridge arm includes a first bridge arm, a second bridge arm, a third bridge arm, a fourth bridge arm, a fifth bridge arm, a sixth bridge arm, a seventh bridge arm, and an eighth bridge arm; the first and second bridge arms form a first group of bridge arms, the third and fourth bridge arms form a second group of bridge arms, the fifth and sixth bridge arms form a third group of bridge arms, and the seventh and eighth bridge arms form a fourth group of bridge arms; wherein, the first bridge arm includes a first switching transistor. T a1 Second switching transistor T a2 The second bridge arm includes the third switching transistor. T b1 Fourth switching transistor T b2 The third bridge arm includes the fifth switching transistor. T c1 Sixth switch tube T c2 The fourth bridge arm includes the seventh switch. T d1 Eighth switch tube T d2 The fifth bridge arm includes the ninth switch. T a3 10th switch tube T a4 The sixth bridge arm includes the eleventh switch. Tb3 12th switch tube T b4 The seventh bridge arm includes the thirteenth switching transistor. T c3 Fourteenth switch tube T c4 The eighth bridge arm includes the fifteenth switching transistor. T d3 Sixteenth switch tube T d4 The coupled inductors include a first coupled inductor, a second coupled inductor, a third coupled inductor, and a fourth coupled inductor; the series-wound motor includes a first stator winding A, a second stator winding B, and a third stator winding C; the first switching transistor... T a1 emitter and second switch T a2 The collector of the transistor and the corresponding terminal of the first coupling inductor are both connected, and the ninth switching transistor is connected. T a3 The emitter and the tenth switch T a4 The collector of the first coupling inductor and the non-identical terminals of the second coupling inductor are both connected; the third switching transistor T b1 emitter and fourth switch T b2 The collector of the first transistor and the corresponding terminal of the second coupling inductor are both connected, and the eleventh switching transistor is connected. T b3 The emitter and the twelfth switch T b4 The collector of the fifth switching transistor and the non-identical terminals of the second coupling inductor are both connected; T c1 The emitter and the sixth switch T c2 The collector of the transistor and the corresponding terminal of the third coupling inductor are both connected, and the thirteenth switching transistor is connected. T c3 The emitter and the fourteenth switch T c4 The collector and the non-identical terminals of the third coupling inductor are both connected; the seventh switch transistor T d1 The emitter and the eighth switch T d2 The collector of the transistor and the corresponding terminal of the fourth coupling inductor are both connected, and the fifteenth switching transistor is connected. T d3 The emitter and the sixteenth switch T d4The collector of the first winding and the non-same-name terminals of the fourth coupling inductor are all connected; the two ends of the first winding A are connected to the first coupling inductor and the second coupling inductor respectively; the two ends of the second winding B are connected to the second coupling inductor and the third coupling inductor respectively; the two ends of the third winding C are connected to the third coupling inductor and the fourth coupling inductor respectively.

[0024] As an optional implementation method, such as Figure 2 As shown, the motor drive system also includes a DC bus capacitor, a DC power supply, the first plate of the DC bus capacitor, and a first switching transistor. T a1 collector, third switching transistor T b1 collector, fifth switch T c1 collector, seventh switch T d1 collector, ninth switch T a3 collector, eleventh switching transistor T b3 collector, thirteenth switching transistor T c3 collector and fifteenth switching transistor T d3 The collectors of all components are connected to the positive terminal of the DC power supply; the second plate of the DC bus capacitor and the second switching transistor... T a2 emitter, fourth switch T b2 emitter, sixth switch T c2 emitter, eighth switch T d2 emitter, tenth switch T a4 The emitter and the twelfth switch T b4 emitter, fourteenth switch T c4 The emitter and the sixteenth switch T d4 The emitters of all are connected to the negative terminal of the DC power supply.

[0025] As an optional implementation method, such as Figure 1 As shown, the motor drive system also includes a current sampling module, a position encoding module, and a speed calculation module. T 1 Coordinate transformation unit, speed PI control module, d Axis PI control module q Axis PI control module T 2 andT3 The system includes a coordinate transformation unit, a first difference operation unit, a second difference operation unit, and a third difference operation unit. The input of the current sampling module is connected between the series winding inverter system and the series winding AC motor. The input of the position encoding module is connected to the series winding AC motor, and its output is connected to the speed calculation module. T 1 Coordinate transformation unit T 2 and T 3 The coordinate transformation unit is connected, and the output of the speed calculation module is connected to the input of the speed PI control module through the first difference calculation unit. The output of the speed PI control module is connected to the input of the speed PI control module through the second difference calculation unit. q The input terminal of the axis PI control module is connected. d Axis PI control module q The output terminals of the axis PI control module are all connected to T 2 and T 3 The input terminal of the coordinate transformation unit is connected. T 2 and T 3 The output terminals of the coordinate transformation unit and the current sampling module are both connected to the current stress optimization module. d The input terminal of the axis PI control module is connected to the third difference calculation unit, and the third difference calculation unit is connected to the output terminal of the T1 coordinate transformation unit.

[0026] The embodiment is merely a specific example and does not indicate that this is the only way to implement the present invention.

[0027] Example 2: A stress optimization method for a ZVS series winding inverter, implemented through the motor drive system in Example 1, such as... Figure 3 , Figure 4 As shown, the process includes the following steps: S100: Transforming the voltage of the series-wound AC motor in the rotating coordinate system into a three-phase reference voltage in the natural coordinate system through coordinate transformation. V a , V b , V c S200: Three-phase reference voltage for AC motors with series windings V a , V b , V cFour sets of bridge arm voltages were used to calculate the modulation waveforms of the series winding inverter under three different zero-sequence voltage injection scenarios; S300: A current ripple prediction model was established to calculate the maximum bridge arm current; S400: Based on the maximum bridge arm current, the modulation waveform was obtained. d x0 , d xmax and d xmin Calculate the soft-switching constraints and the magnitude of the four-arm current and the optimal zero-sequence voltage under this clamping state. V cm and the operating frequency of the switching transistor f s S500: Based on the operating frequency of the switching transistor. f s The reconstructed reference voltage after dynamic zero-sequence voltage injection is obtained. This reconstructed reference voltage is compared with an interleaved carrier wave to generate the switching signal for the switching diodes. This invention ensures zero-voltage switching for all switching transistors while reducing current stress, improving overall system efficiency, and offering strong versatility. It effectively reduces conduction losses and switching transistor current stress requirements, significantly improving the efficiency, reliability, and lifespan of the inverter system. It precisely meets the core requirements of current AC motor drive systems for miniaturization, high power density, and high reliability. The control method of this invention is easy to integrate and implement in existing systems, effectively improving system efficiency and power density, and enhancing the operational reliability and overall performance of the motor drive system under high-frequency switching conditions. It provides an advanced modulation solution for the application of wide-bandgap semiconductor devices in high-performance drive fields.

[0028] As an optional implementation, step S100 includes: S110: Obtain the current position angle of the series-wound AC motor through the position encoding module. θ Position angle θ The input speed calculation module obtains the actual speed of the series-wound AC motor. N , give the speed N ref Compared with actual speed N The difference is calculated by PI control via the speed PI control module to obtain... q First given current of shaft I q_ref ; S120: Obtains the three-phase current output of the series-wound AC motor through the current sampling module. i a , i b , i c Through the first coordinate transformation unit T1 get q First actual current of shaft i q and d First actual current of shaft i d Set the d-axis reference current. I d_ref =0 and d First actual current of shaft i d To make differences, through d The axis PI control module controls the rotation to obtain the coordinate system. d Shaft given voltage V d ;Will q First given current of shaft I q_ref and q First actual current of shaft i q Take the difference, and then pass the difference through... q The axis PI control module performs PI control to obtain the coordinates in the rotating coordinate system. q Shaft given voltage V q ; in, T 1 The transformation formula for the coordinate transformation unit is: ; S130: The given voltage in the rotating coordinate system... V d and V q conduct T 2 Coordinate transformation and T 3 Coordinate transformation yields the three-phase reference voltage of the series-wound AC motor in the natural coordinate system. V a , V b , V c , V a , V b , V c These are the voltages of the first stator winding A, the second stator winding B, and the third stator winding C, respectively. in, T 2 The formula for coordinate transformation is: ; T 3 The formula for coordinate transformation is: .

[0029] As an optional implementation, step S200 specifically includes: S210: The three-phase reference voltage of the series-wound AC motor. V a , V b , V c Represented as: , S220: Based on topological symmetry and voltage balance principles, the three-phase reference voltage of the series-wound AC motor is represented by four sets of bridge arm reference voltages: ; The four-phase modulated wave signals generated by the four sets of bridge arm reference voltages are obtained: , in, m a 、m b , m c , m d These represent the four-phase modulated wave signals generated by the reference voltages of the first, second, third, and fourth bridge arms, respectively. V leg1 , V leg2 , V leg3 , V leg4 These represent the reference voltages for the first bridge arm (corresponding to phase A of the series winding inverter), the second bridge arm (corresponding to phase B of the series winding inverter), the third bridge arm (corresponding to phase C of the series winding inverter), and the fourth bridge arm (corresponding to phase D of the series winding inverter), respectively. m Represents the modulation coefficient. w This indicates the electrical angular frequency of a series-winding AC motor; and the frequency of a current-winding inverter. D Phase reference voltage and the inverter with the current winding A The phase reference voltages are the same, and the amplitude of the bridge arm reference voltage is equal to that of the three-phase voltage. The phase is 30 degrees ahead; S230: For the initial bridge arm reference voltage, three zero-sequence voltage injection methods are set: no injection, maximum injection, and minimum injection. These correspond to three clamping modes: bridge arm non-clamping, bridge arm maximum clamping, and bridge arm minimum clamping, respectively. The calculation formula is as follows: , in x = a, b, c, d max (m x ) The maximum voltage amplitude in the bridge arm reference voltage, min (m x ) This is the minimum voltage amplitude of the bridge arm reference voltage; S240: Based on three different zero-sequence voltage injections, the modulation waveform for driving the series winding inverter is obtained as follows: , in, x = a, b, c, d like Figure 5 As shown. Based on this modulation waveform, a current ripple prediction model can be further established, which provides a key basis for the subsequent comparison and selection of clamping modes.

[0030] As an optional implementation, in step S300, the calculation of the bridge arm current slope at each time period can be derived based on the electromagnetic characteristics of the coupled inductor and the circuit topology, for the series winding inverter. A The phase includes the first bridge arm, the fifth bridge arm, and the first coupling inductor, and its specific operation process is as follows: S310: The first and fifth bridge arms use the same modulation waveform, which is compared with an interleaved 180° carrier wave to obtain the switching signal. Based on the first switching transistor... T a1 and the third switching transistor T a3 The on / off status will be the status within an on / off cycle. A The slope of the phase bridge arm current is expressed as: , in, V dc Indicates the DC bus voltage. di cir / dt Indicates the slope of the current. L This represents the self-inductance of the first coupled inductor. M This represents the mutual inductance of the first coupled inductor. S a1 Indicates the first switching transistor T a1 The switch signal, S a3 Indicates the third switching transistor T a3 The switch signal; S320: List the first switching transistor T a1 Second switching transistorT a2 Ninth switch tube T a3 10th switch tube T a4 All of these can achieve zero-voltage conduction, and the constraints that must be met are: , in, i a1 (t 1 ) For the first bridge arm in t 1 First switching transistor at all times T a1 Bridge arm current at start-up i a1 (t 4 ) For the first bridge arm in t 4 Second switching transistor T a2 Bridge arm current at start-up i a2 (t 2 ) For the fifth bridge arm t 2 Ninth switch at time T a3 Bridge arm current at start-up i a1 (t 3 ) For the fifth bridge arm t 3 Tenth switch tube at time T a4 Bridge arm current at start-up i bias The minimum current required to turn the switching transistor on and off to meet the charging and discharging requirements of the output capacitor of the switching transistor. S330: The duration of different switching states is expressed by duty cycle and switching period as follows: , in, d a Representative is the series winding inverter A Phase reference voltage, T s Indicates the switching cycle. T 1 Corresponding switching transistor S a1Turn-off and switching transistors S a3 Switching from on state to switching transistor S a1 conduction and switching transistors S a3 The time interval between conduction states. T 2 The corresponding switch state remains as a switching transistor. S a1 conduction and switching transistors S a3 The duration of the shutdown; S340: Based on the bridge arm current slope and switching state time, t 1 First switching transistor at all times T a1 When it is opened and t 4 Second switching transistor T a2 The first arm current at the start-up point is expressed as follows: ; S350: Based on the reciprocal relationship between switching frequency fs and switching period Ts, the soft-switching condition is represented by frequency degrees of freedom, and the frequency constraint of phase A is defined. Represented as: ; S360: Calculate the soft-switching frequency constraints for phases B, C, and D of the series-wound inverter, and select the minimum switching frequency as the operating frequency. : ; S370: Series winding inverter A Maximum value of circulating ripple in phase icir a Represented as: , in V dc Bus voltage f s The final selected system switching frequency; S380: Based on the maximum value of circulating ripple icir a AC motor with series winding A Phase current i a The relationship between the two is that the maximum value of the bridge arm current corresponding to inverter A with series windings is determined. Represented as: Using the same method, the maximum value of the bridge arm current corresponding to inverter B with series windings can be calculated. The maximum value of the bridge arm current corresponding to C of the series winding inverter. The maximum value of the bridge arm current corresponding to the series winding inverter D. .

[0031] As an optional implementation, step S400 specifically includes: S410: The maximum peak current of the four phases of the series winding inverter under different zero-sequence voltages is used as the core comparison benchmark, and its expression is defined as: , like Figure 6 As shown, where ipeak v0 , ipeak vmax , ipeak vmin These represent the reference voltage modulation waveforms when there is no injection, maximum injection, and minimum injection zero-sequence voltage, respectively. d x0 , d xmax and d xmin ; S420: Defines the minimum peak current under three operating conditions: no injection, maximum injection, and minimum injection zero-sequence voltage. peak opt And use it as the threshold for determining the clamping mode selection, that is: ; S430: Obtain the optimal zero-sequence voltage for minimum current stress in the bridge arm. V cm : , like peak opt = ipeak v0 ,choose v 0 The corresponding clamping mode; if peak opt = ipeak vmax or peak opt = ipeak vmin Select respectively v max and v max The corresponding clamping modes, such as Figure 6 As shown; S440: The operating frequency of the switching transistor... f s Represented as: , Operating frequency of the switching transistor f s The switching is calculated based on the dynamic clamping interval. f 0 represent V cm for v 0 The switching frequency required for soft switching of the switching transistor. f max represent V cm for V max The switching frequency required for soft switching of the switching transistor. f min represent V cm for V min The switching frequency required for soft switching of the switching transistor.

[0032] As an optional implementation, the specific operation of step S500 is as follows: the reference voltage reconstructed after dynamic zero-sequence injection is used as a modulation signal, compared with the variable switching frequency interleaved carrier, to generate the final PWM switching signal, and output to the series winding inverter system, so that the three-phase current is output through the series winding motor.

[0033] This invention can significantly reduce the current stress of the four bridge arms across the entire operating range and effectively improve system efficiency, thereby bringing significant improvements in system efficiency, reliability, adaptability, and cost. Specifically, this is reflected in the following aspects: (1) Improved efficiency and reduced energy consumption. For example... Figures 13-15 As shown, scanning the current stress across the entire operating range, the peak current stress of the traditional SVPWM method is 14.724 times the phase current, the DPWM method is 7.70352 times, while the dynamic zero-sequence method in this embodiment is only 5.71463 times. The reduction in current stress directly reduces the conduction and switching losses of switching devices and magnetic components, thereby significantly improving the overall system drive efficiency and reducing operating energy consumption. The efficiency improvement has been fully verified in actual operation: as shown... Figure 11 , Figure 12As shown, under a wide range of operating conditions with different speeds and load currents, the efficiency of the dynamic zero-sequence injection algorithm in this embodiment is superior to both the DPWM and SVPWM algorithms. Compared to the SVPWM algorithm, its efficiency improvement is particularly significant—reaching a 5.2% improvement at a speed of 1200 r / min and a 5.3% improvement at a load current of 3A. Compared to the DPWM algorithm, it still achieves a 2.3% efficiency improvement at 1200 r / min and a 4.2% efficiency improvement at a load current of 3A.

[0034] (2) Current stress distribution optimization and reliability enhancement: The dynamic zero-sequence injection method in this embodiment overcomes the limitation that the stress of traditional DPWM may not be lower than that of SVPWM under certain low modulation ratios by calculating the circulating current in real time and dynamically selecting the clamping interval, thus achieving the optimal current stress distribution across the entire operating range. This is directly verified in the actual current waveform, such as... Figure 8 As shown, under the same load conditions with a modulation ratio of 1.1 and a power factor of 0.8, the peak-to-peak value of the bridge arm current of SVPWM reaches as high as 35.4A, while DPWM and the dynamic zero-sequence method of this embodiment are optimized to 24.9A and 23.8A, respectively. Figure 9 As shown, under the same load conditions with a modulation ratio of 1.5 and a power factor of 1, the peak-to-peak value of the bridge arm current of SVPWM reaches as high as 45.8A, while DPWM and the dynamic zero-sequence method of this embodiment are optimized to 31.8A and 27.3A, respectively. Figure 10 As shown, under the same load conditions with a modulation ratio of 1.5 and a power factor of 0.8, the peak-to-peak value of the bridge arm current of SVPWM reaches as high as 42.8A, while that of DPWM and the dynamic zero-sequence method of this embodiment is optimized to 28.3A and 23.1A, respectively. This indicates that the dynamic zero-sequence injection method of this embodiment has superior peak current suppression capability. This not only reduces the current capacity requirements of power devices but also helps to reduce thermal stress and improve the long-term operational reliability of the system.

[0035] (3) Improved control flexibility and adaptability: Compared to DPWM with a fixed clamping range, the dynamic zero-sequence injection method can flexibly adjust the clamping strategy according to operating conditions, thereby maintaining low current stress under various load and modulation ratio conditions. This flexibility brings advantages that become more pronounced with changing operating conditions: for example... Figure 9 As shown, when the modulation ratio is low and the power factor is close to 1, the dynamic zero-sequence injection method in this embodiment is similar to the DPWM current stress; however, as the modulation ratio increases and the power factor changes, the optimization effect of the dynamic zero-sequence injection method is significantly enhanced, achieving better current stress control across the entire operating range (compared to...). Figures 13-15 (The scan results are consistent). This adaptability allows the system to maintain efficient and stable operation under a wide range of working conditions, improving overall control accuracy and quality.

[0036] (4) Reduced requirements for components: Significant reduction in peak current stress (due to...) Figures 8-10 and Figures 13-15 (As confirmed by both parties) This allows for the use of power semiconductor devices and magnetic components with smaller current specifications, thereby saving raw material costs and helping to achieve system miniaturization and weight reduction.

[0037] The above description is merely a preferred embodiment of the present invention. Those skilled in the art will understand that various changes or equivalent substitutions can be made to these features and embodiments without departing from the spirit and scope of the present invention. Furthermore, under the teachings of the present invention, these features and embodiments can be modified to adapt to specific situations and materials without departing from the spirit and scope of the present invention. Therefore, the present invention is not limited to the specific embodiments disclosed herein, and all embodiments falling within the scope of the claims of this application are within the protection scope of the present invention.

Claims

1. A motor drive system, characterized in that, The system includes a current stress optimization module, a series winding inverter system, and a series winding motor connected in sequence. The series winding inverter system includes a series winding inverter and a coupling inductor. The series winding inverter includes eight parallel bridge arms, each bridge arm including two parallel switching transistors. The number of coupling inductors is four, with two adjacent bridge arms connected to one coupling inductor. The current stress optimization module generates a switching drive signal that meets the timing requirements of the series winding inverter system using a current stress optimization method. The switching drive signal adjusts the voltage output of the series winding inverter by controlling the on / off state of the switching transistors of the series winding inverter, thereby driving the series winding motor.

2. The motor drive system according to claim 1, characterized in that, The bridge arm includes a first bridge arm, a second bridge arm, a third bridge arm, a fourth bridge arm, a fifth bridge arm, a sixth bridge arm, a seventh bridge arm, and an eighth bridge arm; the first and second bridge arms form a first group of bridge arms, the third and fourth bridge arms form a second group of bridge arms, the fifth and sixth bridge arms form a third group of bridge arms, and the seventh and eighth bridge arms form a fourth group of bridge arms; wherein, the first bridge arm includes a first switching transistor. T a1 Second switching transistor T a2 The second bridge arm includes a third switching transistor. T b1 Fourth switching transistor T b2 The third bridge arm includes a fifth switching transistor. T c1 Sixth switch tube T c2 The fourth bridge arm includes a seventh switching transistor. T d1 Eighth switch tube T d2 The fifth bridge arm includes a ninth switching transistor. T a3 10th switch tube T a4 The sixth bridge arm includes an eleventh switching transistor. T b3 12th switch tube T b4 The seventh bridge arm includes a thirteenth switching transistor. T c3 Fourteenth switch tube T c4 The eighth bridge arm includes a fifteenth switching transistor. T d3 Sixteenth switch tube T d4 The coupling inductors include a first coupling inductor, a second coupling inductor, a third coupling inductor, and a fourth coupling inductor; the series-wound motor includes a first stator winding A, a second stator winding B, and a third stator winding C; the first switching transistor... T a1 emitter and second switch T a2 The collector of the transistor and the corresponding terminal of the first coupling inductor are both connected, and the ninth switching transistor is connected. T a3 The emitter and the tenth switch T a4 The collector of the first coupling inductor and the non-identical terminals of the second coupling inductor are both connected; the third switching transistor T b1 emitter and fourth switch T b2 The collector of the first transistor and the corresponding terminal of the second coupling inductor are both connected, and the eleventh switching transistor is connected. T b3 The emitter and the twelfth switch T b4 The collector of the fifth switching transistor and the non-identical terminals of the second coupling inductor are both connected; T c1 The emitter and the sixth switch T c2 The collector of the transistor and the corresponding terminal of the third coupling inductor are both connected, and the thirteenth switching transistor is connected. T c3 The emitter and the fourteenth switch T c4 The collector and the non-identical terminals of the third coupling inductor are both connected; the seventh switch transistor T d1 The emitter and the eighth switch T d2 The collector of the transistor and the corresponding terminal of the fourth coupling inductor are both connected, and the fifteenth switching transistor is connected. T d3 The emitter and the sixteenth switch T d4 The collector of the first winding and the non-same-name terminals of the fourth coupling inductor are all connected; the two ends of the first winding A are respectively connected to the first coupling inductor and the second coupling inductor; the two ends of the second winding B are respectively connected to the second coupling inductor and the third coupling inductor; the two ends of the third winding C are respectively connected to the third coupling inductor and the fourth coupling inductor.

3. The motor drive system according to claim 2, characterized in that, The motor drive system also includes a DC bus capacitor and a DC power supply; the first plate and the first switching transistor of the DC bus capacitor. T a1 collector, third switching transistor T b1 collector, fifth switch T c1 collector, seventh switch T d1 collector, ninth switch T a3 collector, eleventh switching transistor T b3 collector, thirteenth switching transistor T c3 collector and fifteenth switching transistor T d3 The collectors of all components are connected to the positive terminal of the DC power supply; the second plate of the DC bus capacitor and the second switching transistor... T a2 emitter, fourth switch T b2 emitter, sixth switch T c2 emitter, eighth switch T d2 emitter, tenth switch T a4 The emitter and the twelfth switch T b4 emitter, fourteenth switch T c4 The emitter and the sixteenth switch T d4 The emitters of all components are connected to the negative terminal of the DC power supply.

4. The motor drive system according to claim 3, characterized in that, The motor drive system also includes a current sampling module, a position encoding module, and a speed calculation module. T 1 Coordinate transformation unit, speed PI control module, d Axis PI control module q Axis PI control module T 2 and T3 The system includes a coordinate transformation unit, a first difference calculation unit, a second difference calculation unit, and a third difference calculation unit. The input terminal of the current sampling module is connected between the series winding inverter system and the series winding AC motor. The input terminal of the position encoding module is connected to the series winding AC motor, and its output terminal is connected to the speed calculation module. T 1 Coordinate transformation unit T 2 and T 3 The coordinate transformation unit is connected, and the output of the speed calculation module is connected to the input of the speed PI control module through the first difference calculation unit. The output of the speed PI control module is connected to the input of the speed PI control module through the second difference calculation unit. q The input terminal of the axis PI control module is connected, the d Axis PI control module q The output terminals of the axis PI control module are all connected to the T 2 and T 3 The input terminal of the coordinate transformation unit is connected, the T 2 and T 3 The output terminals of the coordinate transformation unit and the current sampling module are both connected to the current stress optimization module. d The input terminal of the axis PI control module is connected to the third difference calculation unit, and the third difference calculation unit is connected to the output terminal of the T1 coordinate transformation unit.

5. A stress optimization method for a ZVS series winding inverter, characterized in that, Operating the motor drive system of claim 4 includes the following steps: S100: Transforms the voltage of a series-wound AC motor in a rotating coordinate system into a three-phase reference voltage in a natural coordinate system through coordinate transformation. V a , V b , V c ; S200: Three-phase reference voltage for AC motors with series windings V a , V b , V c Four sets of bridge arm voltages were used to calculate the modulation waveforms of the series winding inverter under three different zero-sequence voltage injection scenarios. S300: Establish a current ripple prediction model and calculate the maximum value of the bridge arm current; S400: Based on the maximum value of the bridge arm current, through modulation wave d x0 , d xmax and d xmin Calculate the soft-switching constraints and the magnitude of the four-arm current and the optimal zero-sequence voltage under this clamping state. V cm and the operating frequency of the switching transistor f s ; S500: Based on the operating frequency of the switching transistor f s The reconstructed reference voltage after dynamic zero-sequence voltage injection is obtained. The reconstructed reference voltage is compared with the interleaved carrier to generate the switching signal of the switching tube.

6. The stress optimization method for ZVS series winding inverters according to claim 5, characterized in that, The S100 step includes: S110: Obtain the current position angle of the series-wound AC motor through the position encoding module. θ Position angle θ The input speed calculation module obtains the actual speed of the series-wound AC motor. N , give the speed N ref Compared with actual speed N The difference is calculated by PI control via the speed PI control module to obtain... q First given current of shaft I q_ref ; S120: Obtains the three-phase current output of the series-wound AC motor through the current sampling module. i a , i b , i c Through the first coordinate transformation unit T 1 get q First actual current of shaft i q and d First actual current of shaft i d Set the d-axis reference current. I d_ref =0 and d First actual current of shaft i d To make a difference, through d The axis PI control module controls the rotation to obtain the coordinate system. d Shaft given voltage V d ;Will q First given current of shaft I q_ref and q First actual current of shaft i q Take the difference, and then pass the difference through... q The axis PI control module performs PI control to obtain the coordinates in the rotating coordinate system. q Shaft given voltage V q ; in, T 1 The transformation formula for the coordinate transformation unit is: ； S130: The given voltage in the rotating coordinate system... V d and V q conduct T 2 Coordinate transformation and T 3 Coordinate transformation yields the three-phase reference voltage of the series-wound AC motor in the natural coordinate system. V a , V b , V c , V a , V b , V c These are the voltages of the first stator winding A, the second stator winding B, and the third stator winding C, respectively. in, T 2 The formula for coordinate transformation is: ； T 3 The formula for coordinate transformation is: 。 7. The stress optimization method for ZVS series winding inverters according to claim 6, characterized in that, The specific steps of S200 are as follows: S210: The three-phase reference voltage of the series-wound AC motor. V a , V b , V c Represented as: ， S220: Based on topological symmetry and voltage balance principles, the three-phase reference voltage of the series-wound AC motor is represented by four sets of bridge arm reference voltages: ； The four-phase modulated wave signals generated by the four sets of bridge arm reference voltages are obtained: ， in, m a 、m b , m c , m d These represent the four-phase modulated wave signals generated by the reference voltages of the first, second, third, and fourth bridge arms, respectively. V leg1 , V leg2 , V leg3 , V leg4 These represent the reference voltages for the first, second, third, and fourth bridge arms, respectively. m Represents the modulation coefficient. w This indicates the electrical angular frequency of an AC motor with series windings. S230: For the initial bridge arm reference voltage, three zero-sequence voltage injection methods are set: no injection, maximum injection, and minimum injection. These correspond to three clamping modes: bridge arm non-clamping, bridge arm maximum clamping, and bridge arm minimum clamping, respectively. The calculation formula is as follows: ， in x = a, b, c, d max (m x ) The maximum voltage amplitude in the bridge arm reference voltage, min (m x ) This is the minimum voltage amplitude of the bridge arm reference voltage; S240: Based on three different zero-sequence voltage injections, the modulation waveform for driving the series winding inverter is obtained as follows: ， in, x = a, b, c, d .

8. The stress optimization method for ZVS series winding inverters according to claim 7, characterized in that, In step S300, the series winding inverter A The phase includes the first bridge arm, the fifth bridge arm, and the first coupling inductor, and its specific operation process is as follows: S310: Based on the first switching transistor T a1 and the third switching transistor T a3 The on / off status will be the status within an on / off cycle. A The slope of the phase bridge arm current is expressed as: ， in, V dc Indicates the DC bus voltage. di cir / dt Indicates the slope of the current. L This represents the self-inductance of the first coupled inductor. M This represents the mutual inductance of the first coupled inductor. S a1 Indicates the first switching transistor T a1 The switch signal, S a3 Indicates the third switching transistor T a3 The switch signal; S320: List the first switching transistor T a1 Second switching transistor T a2 Ninth switch tube T a3 10th switch tube T a4 All of these can achieve zero-voltage conduction, and the constraints that must be met are: ， in, i a1 (t 1 ) For the first bridge arm in t 1 First switching transistor at all times T a1 Bridge arm current at start-up i a1 (t 4 ) For the first bridge arm in t 4 Second switching transistor T a2 Bridge arm current at start-up i a2 (t 2 ) For the fifth bridge arm t 2 Ninth switch at time T a3 Bridge arm current at start-up i a1 (t 3 ) For the fifth bridge arm t 3 Tenth switch tube at time T a4 Bridge arm current at start-up i bias The minimum current required to turn the switching transistor on and off to meet the charging and discharging requirements of the output capacitor of the switching transistor. S330: The duration of different switching states is expressed by duty cycle and switching period as follows: ， in, d a Representative is the series winding inverter A Phase reference voltage, T s Indicates the switching cycle. T 1 Corresponding switching transistor S a1 Turn-off and switching transistors S a3 Switching from on state to switching transistor S a1 conduction and switching transistors S a3 The time interval between conduction states. T 2 The corresponding switch state remains as a switching transistor. S a1 conduction and switching transistors S a3 The duration of the shutdown; S340: Based on the bridge arm current slope and switching state time, t 1 First switching transistor at all times T a1 When it is opened and t 4 Second switching transistor T a2 The first arm current at the start-up point is expressed as follows: ； S350: Based on the reciprocal relationship between switching frequency fs and switching period Ts, the soft-switching condition is represented by frequency degrees of freedom, and the frequency constraint of phase A is defined. Represented as: ； S360: Calculate the soft-switching frequency constraints for phases B, C, and D of the series-wound inverter, and select the minimum switching frequency as the operating frequency. : ； S370: Series winding inverter A Maximum value of circulating ripple in phase icir a Represented as: ， in V dc Bus voltage f s The final selected system switching frequency; S380: Based on the maximum value of circulating ripple icir a AC motor with series winding A Phase current i a The relationship between the maximum value of the bridge arm current and the bridge arm current. Represented as: 。 9. The stress optimization method for ZVS series winding inverters according to claim 8, characterized in that, The S400 step specifically includes: S410: The maximum peak current of the four phases of the series winding inverter under different zero-sequence voltages is used as the core comparison benchmark, and its expression is defined as: ， in ipeak v0 , ipeak vmax , ipeak vmin These represent the reference voltage modulation waveforms when there is no injection, maximum injection, and minimum injection zero-sequence voltage, respectively. d x0 , d xmax and d xmin ; S420: Defines the minimum peak current under three operating conditions: no injection, maximum injection, and minimum injection zero-sequence voltage. peak opt And use it as the threshold for determining the clamping mode selection, that is: ； S430: Obtain the optimal zero-sequence voltage for minimum current stress in the bridge arm. V cm : ， like peak opt = ipeak v0 ,choose v 0 The corresponding clamping mode; if peak opt = ipeak vmax or peak opt = ipeak vmin Select respectively v max and v max Corresponding clamping mode; S440: The operating frequency of the switching transistor... f s Represented as: ， Operating frequency of the switching transistor f s The switching is calculated based on the dynamic clamping interval. f 0 represent V cm for v 0 The switching frequency required for soft switching of the switching transistor. f max represent V cm for V max The switching frequency required for soft switching of the switching transistor. f min represent V cm for V min The switching frequency required for soft switching of the switching transistor.

10. The stress optimization method for ZVS series winding inverters according to claim 9, characterized in that, The specific operation of step S500 is as follows: the reference voltage reconstructed after dynamic zero-sequence injection is used as the modulation signal, compared with the alternating carrier of the variable switching frequency, to generate the final PWM switching signal, and output to the series winding inverter system, so that the three-phase current is output through the series winding motor.

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