Magnetorheological damper temperature self-adaptive compensation control circuit and method thereof

Abstract

The present application relates to the technical field of magneto-rheological shock absorber, and particularly relates to a temperature self-adaptive compensation control circuit and method of magneto-rheological shock absorber, which comprises: a power management power supply control circuit, a Boost voltage boosting circuit, an H-bridge driving output circuit, a magneto-rheological shock absorber two-end voltage sampling circuit and a microcontroller MCU which are electrically connected in sequence; the microcontroller MCU is configured to: acquire a real-time current signal collected by the H-bridge driving output circuit and a real-time voltage signal collected by the voltage sampling circuit; in the present application, the voltage across the excitation coil and the current flowing through the coil are acquired in real time through the voltage sampling circuit and the current collection circuit, and the current resistance value of the excitation coil is calculated in real time by the microcontroller MCU according to Ohm's law. The present scheme directly reflects the coil body temperature, has fast response speed, high measurement accuracy, and is not affected by environmental temperature, heat dissipation conditions and installation process differences.

Description

Technical Field

[0001] This invention relates to the field of magnetorheological damper technology, and in particular to a temperature adaptive compensation control circuit and method for a magnetorheological damper. Background Technology

[0002] Magnetorheological dampers are widely used in modern automotive semi-active suspension systems due to their significant advantages such as fast response speed and continuously adjustable damping force. Their working principle involves adjusting the current flowing through the excitation coil to change the magnetic field strength of the magnetorheological fluid, thereby altering the yield stress of the fluid and achieving continuous, stepless adjustment of the damping force.

[0003] Currently, in actual operation, the excitation coil of a magnetorheological damper experiences a temperature rise due to the thermal effect of the current. The excitation coil is typically wound with enameled copper wire, whose resistivity increases significantly with temperature, causing a noticeable shift in the coil's internal resistance with changes in operating temperature. This change in the excitation coil's internal resistance directly leads to a decrease in the actual driving current under the same control duty cycle. Consequently, the damping force output by the control algorithm designed based on a fixed parameter model deviates from the expected target value, severely affecting the control accuracy and consistency of the suspension system.

[0004] To address the aforementioned temperature drift problem, a common solution in existing technologies is the software lookup table compensation method. This involves pre-calibrating the resistance parameters or current correction coefficients of the excitation coil at different temperatures through bench tests, and then acquiring temperature signals using temperature sensors installed on or near the shock absorber housing during actual control. The control parameters are then compensated using a lookup table. However, this method has inherent drawbacks: the temperature sensors measure the ambient temperature or the temperature of the shock absorber housing, not the actual temperature inside the excitation coil. Significant thermal hysteresis and temperature gradients exist between these two temperatures, resulting in slow temperature sensing response, untimely compensation, and poor accuracy.

[0005] Therefore, in order to solve the above problems, we propose a temperature adaptive compensation control circuit and method for magnetorheological dampers. Summary of the Invention

[0006] The purpose of this invention is to address the shortcomings of existing technologies where temperature sensors measure ambient temperature or the temperature of the damper housing, rather than the actual temperature inside the excitation coil. Significant thermal conduction hysteresis and temperature gradient exist between the two, resulting in slow temperature sensing response, untimely compensation, and poor accuracy. Therefore, this invention proposes a magnetorheological damper temperature adaptive compensation control circuit and method.

[0007] To achieve the above objectives, the present invention adopts the following technical solution:

[0008] Design a temperature adaptive compensation control circuit for a magnetorheological damper, comprising: a power management control circuit, a boost circuit, an H-bridge drive output circuit, and a voltage sampling circuit and a microcontroller (MCU) connected in sequence.

[0009] The microcontroller (MCU) is configured as follows:

[0010] The real-time current signal collected by the H-bridge drive output circuit and the real-time voltage signal collected by the voltage sampling circuit are obtained, and the current internal resistance value of the excitation coil is calculated.

[0011] Based on the current internal resistance value, the reference internal resistance value, and the temperature coefficient of resistance of the enameled wire, the real-time temperature of the excitation coil is calculated in reverse to compensate for the temperature of the drive current, and the Boost circuit is controlled to boost the voltage when the required drive voltage is higher than the input voltage.

[0012] Furthermore, it also includes a power protection circuit, which includes:

[0013] A surge suppression unit is connected in parallel to the power input terminal to absorb surge voltage;

[0014] The reverse connection protection control unit includes an ideal diode controller and an N-channel MOSFET connected to the ideal diode controller. The ideal diode controller detects the voltage difference between the source and drain of the N-channel MOSFET, and turns off the N-channel MOSFET when a reverse voltage is detected to achieve reverse connection protection. When the MOSFET is forward-biased, the controller adjusts the gate voltage to maintain the N-channel MOSFET in a low on-state voltage drop.

[0015] Furthermore, it also includes a power management circuit, which includes:

[0016] The Buck step-down circuit has its input terminal connected to the output terminal of the power protection circuit, and is used to step down the vehicle power supply voltage to the first intermediate voltage.

[0017] A low-dropout linear regulator, whose input is connected to the output of the Buck step-down circuit, is used to step down the first intermediate voltage to a constant low-voltage power supply for the microcontroller MCU and peripheral chips.

[0018] Furthermore, the Boost circuit includes a synchronous four-switch Buck-Boost power chip, which operates in continuous conduction mode, and its switching frequency and output voltage are configured to meet the drive voltage requirements of the excitation coil at the highest operating temperature; the Boost circuit also includes an output inductor connected to the synchronous four-switch Buck-Boost power chip, a capacitor for current slope compensation, and a resistor-capacitor network for type II compensation.

[0019] Furthermore, the H-bridge drive output circuit adopts a four-switch H-bridge structure, which achieves current direction switching by controlling the conduction state of the four switching transistors, wherein:

[0020] When the first and fourth switches are turned on and the second and third switches are turned off, the current flows from the bus voltage through the first switch, the excitation coil, and the fourth switch to ground.

[0021] When the second and third switches are turned on and the first and fourth switches are turned off, the current flows from the bus voltage through the second switch, the excitation coil, and the third switch to ground.

[0022] Furthermore, the voltage sampling circuit includes:

[0023] The first and second resistor voltage divider networks are used to attenuate the voltage between the two ends of the magnetorheological damper and ground.

[0024] A voltage follower, connected to the voltage divider point of the resistor divider network, is used to isolate the influence of downstream loads;

[0025] A voltage reference circuit is used to raise the reference level of the voltage follower to a preset potential that is compatible with the input range of the analog-to-digital converter of the microcontroller (MCU).

[0026] An RC filter circuit is used to filter out high-frequency switching noise generated by the drive circuit, and outputs a voltage signal that is compatible with the analog-to-digital converter of the microcontroller (MCU).

[0027] A second aspect of this invention also proposes a temperature adaptive compensation control method for a magnetorheological damper, applied to the aforementioned control circuit, specifically including the following steps:

[0028] S1: Real-time acquisition of the voltage value across the magnetorheological damper and the current value flowing through the magnetorheological damper at the corresponding moment, and calculation of the internal resistance of the current excitation coil;

[0029] S2: Obtain the pre-calibrated reference temperature and the reference cold resistance at the reference temperature, and calculate the current temperature of the magnetorheological damper by back-calculating based on the internal resistance of the current excitation coil, the reference cold resistance, and the temperature coefficient of resistance of the enameled wire;

[0030] S3: Based on the current temperature, calculate and output the compensated target driving current according to the preset magnetorheological damper damping force inverse model;

[0031] S4: Determine whether the current system bus voltage can meet the requirements of the target drive current output. If not, control the Boost circuit to increase the bus voltage.

[0032] S5: Control the H-bridge drive output circuit to drive the magnetorheological damper according to the target drive current.

[0033] Furthermore, in step S2, the formula for calculating the current temperature of the magnetorheological damper is as follows:

[0034] Tt = T0 + (Rt - R0) / (α × R0)

[0035] Where Tt is the current temperature obtained by reverse calculation, T0 is the reference temperature, Rt is the internal resistance of the current excitation coil, R0 is the reference cold resistance at the reference temperature, and α is the temperature coefficient of resistance of the enameled wire inside the excitation coil.

[0036] Furthermore, in step S1, the process of obtaining the voltage value across the magnetorheological damper includes:

[0037] The voltage to ground at both ends of the magnetorheological damper is collected and attenuated by voltage division.

[0038] Perform differential calculation on the attenuated voltages at both ends, and superimpose a preset DC reference voltage on the differential calculation result;

[0039] The superimposed voltage signal is low-pass filtered to remove the high-frequency noise from the PWM chopping generated by the H-bridge drive, thus obtaining the final voltage value.

[0040] Furthermore, a thermal protection determination step is included after step S2:

[0041] Determine whether the current temperature obtained by reverse calculation exceeds the preset safe temperature threshold;

[0042] If so, the thermal protection mode is triggered, forcibly reducing the amplitude of the target drive current to prevent the excitation coil from burning out or the oil seal from aging.

[0043] This invention proposes a temperature adaptive compensation control circuit and method for a magnetorheological damper. The advantages are as follows: In this invention, the voltage across the excitation coil and the current flowing through the coil are acquired in real time through a voltage sampling circuit and a current acquisition circuit. The microcontroller (MCU) then calculates the current internal resistance of the excitation coil in real time based on Ohm's law. Since the temperature coefficient of resistance of the enameled wire used in the excitation coil is a known physical constant, the MCU can directly infer the true internal temperature of the coil based on the change in internal resistance, without relying on a temperature sensor installed in the housing or environment. Compared to the indirect temperature measurement method in existing technologies that uses external temperature sensors for table lookup compensation, this solution directly reflects the coil body temperature, has a fast response speed, high measurement accuracy, and is unaffected by differences in ambient temperature, heat dissipation conditions, and installation processes. Attached Figure Description

[0044] Figure 1 This is a system hardware block diagram of the present invention;

[0045] Figure 2 This is a power protection circuit diagram for the present invention;

[0046] Figure 3 This is a power control circuit diagram of the present invention;

[0047] Figure 4 This is a Boost converter circuit diagram of the present invention;

[0048] Figure 5 The H-bridge drive output circuit of the present invention Figure 1 ;

[0049] Figure 6 The H-bridge drive output circuit of the present invention Figure 2 ;

[0050] Figure 7 This is a voltage sampling circuit diagram of the present invention;

[0051] Figure 8 This is a flowchart of the control method in Embodiment 2 of the present invention. Detailed Implementation

[0052] The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings of the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments.

[0053] Example 1

[0054] Reference Figures 1-7As an embodiment of the present invention, a temperature adaptive compensation control circuit for a magnetorheological damper is disclosed. The control circuit includes: a power management power supply control circuit, a boost circuit, an H-bridge drive output circuit, and a voltage sampling circuit and a microcontroller (MCU) connected in sequence.

[0055] The microcontroller (MCU) is configured as follows:

[0056] The real-time current signal collected by the H-bridge drive output circuit and the real-time voltage signal collected by the voltage sampling circuit are obtained, and the current internal resistance value of the excitation coil is calculated.

[0057] Based on the current internal resistance value, the reference internal resistance value, and the temperature coefficient of resistance of the enameled wire, the real-time temperature of the excitation coil is calculated in reverse to compensate for the temperature of the drive current, and the Boost circuit is controlled to boost the voltage when the required drive voltage is higher than the input voltage.

[0058] Specifically, this invention uses a voltage sampling circuit and a current acquisition circuit to acquire the voltage across the excitation coil and the current flowing through the coil in real time. The microcontroller (MCU) then calculates the current internal resistance of the excitation coil in real time based on Ohm's law. Since the temperature coefficient of resistance of the enameled wire used in the excitation coil is a known physical constant, the MCU can directly deduce the true internal temperature of the coil based on the change in internal resistance, without relying on a temperature sensor installed in the housing or environment. Compared to the indirect temperature measurement method in existing technologies that uses an external temperature sensor for table lookup compensation, this solution directly reflects the coil body temperature, has a fast response speed, high measurement accuracy, and is unaffected by differences in ambient temperature, heat dissipation conditions, and installation processes.

[0059] Secondly, during operation, the magnetorheological damper experiences a temperature increase and internal resistance increase in the excitation coil due to the current heating effect, which in turn causes the actual drive current to decrease under the same control duty cycle, resulting in the damping force deviating from the target value. In this solution, the microcontroller (MCU) calls the preset inverse damping force model based on real-time temperature information to perform dynamic temperature compensation on the target drive current, ensuring that the drive current that matches the desired damping force can be output at different operating temperatures, effectively overcoming the adverse effects of temperature drift on suspension performance.

[0060] Reference Figure 2 The present invention also includes a power protection circuit, which comprises:

[0061] A surge suppression unit is connected in parallel to the power input terminal to absorb surge voltage;

[0062] The reverse connection protection control unit includes an ideal diode controller and an N-channel MOSFET connected to the ideal diode controller. The ideal diode controller detects the voltage difference between the source and drain of the N-channel MOSFET, and turns off the N-channel MOSFET when a reverse voltage is detected to achieve reverse connection protection. When the MOSFET is forward-biased, the controller adjusts the gate voltage to maintain the N-channel MOSFET in a low on-state voltage drop.

[0063] The main function of power protection circuits is to provide surge protection and reverse connection protection, such as... Figure 2 As shown, a D010 and a TVS diode SM8S24A are added to the battery power input port VBATJ for surge protection. The reverse connection protection circuit uses a combination of an ideal diode controller SCT53600 and an NMOS transistor YJB5D0G06HQ. The source and drain voltages of the MOSFET YJB5D0G06HQ are detected between the ANODE and CATHODE pins of the ideal diode controller SCT53600. When the voltage across these pins is detected to be -11 to 50 mV, the SCT53600 operates in forward regulation mode. By adjusting the voltage from GATE to ANODE, the voltage from ANODE to CATHODE can be adjusted to 20 mV. This closed-loop regulation scheme can adjust and shut down the external MOSFET under very small loads and ensure zero DC reverse current. When the voltage across the terminals is typically greater than 50 mV, the SCT53600 operates in full-on mode, where the voltage drop between GATE and ANODE is at its maximum to ensure that the RDS(ON) of the external MOSFET is fully conducted. When the voltage from ANODE to CATHODE is less than -11 mV, reverse current protection mode is activated. The GATE is internally connected to the ANODE anode to shut down the external MOSFET, thus achieving reverse connection protection. Output capacitors C011~C013 are used to prevent power supply fluctuations, C010 is a charge pump capacitor used to drive the external MOSFET to its maximum gate drive voltage, and R010 is a current-limiting resistor.

[0064] The parameters and connection methods of each electrical component can be found in the following references. Figure 2 As shown, I will not go into detail here.

[0065] Based on the above embodiments, the present invention further includes a power management circuit, the power management circuit comprising:

[0066] The Buck step-down circuit has its input terminal connected to the output terminal of the power protection circuit, and is used to step down the vehicle power supply voltage to the first intermediate voltage.

[0067] A low-dropout linear regulator, whose input is connected to the output of the Buck step-down circuit, is used to step down the first intermediate voltage to a constant low-voltage power supply for the microcontroller MCU and peripheral chips.

[0068] Specifically, the power management control circuit, such as Figure 3 As shown, its main function is to step down the vehicle battery voltage to 3.3V, providing a power input that meets the operating requirements of the MUC and other ICs. This case uses the automotive-grade Buck chip SCT2632QSTER and LDO chip SCT71405F33BQ-STER from Chipsys Technology to convert the battery voltage to 3.3V. An LC circuit consisting of L3 and C304 is added to the VIN port of the Buck chip to filter out power ripple; R303 is used to set the switching frequency of the chip; by matching different resistance values ​​according to Fsw=100000 / R303, the switching frequency of the chip can be changed; the output inductor L3_1 is...

[0069]

[0070] The appropriate inductance value of output inductor L3_1 is matched, where ∆Iout is the ripple current; the output voltage is Vout=0.8*(1+R304 / R305), and the ratio of feedback resistors R304 and R305 is matched according to the required output 5V voltage; output capacitors C308 and C309 are used as energy storage capacitors to ensure the stability of the secondary power input; the 5V voltage generated by the Buck circuit is input to LDOSCT71405F33BQ-STER to generate a 3.3V voltage to power the MCU and other ICs.

[0071] The parameters and connection methods of each electrical component can be found in the following references. Figure 3 As shown, I will not go into detail here.

[0072] Reference Figure 4 The Boost circuit includes a synchronous four-switch Buck-Boost power chip, which operates in continuous conduction mode. Its switching frequency and output voltage are configured to meet the drive voltage requirements of the excitation coil at the highest operating temperature. The Boost circuit also includes an output inductor connected to the synchronous four-switch Buck-Boost power chip, a capacitor for current slope compensation, and a resistor-capacitor network for type II compensation.

[0073] Boost converter circuit, such as Figure 4As shown, the selected TI synchronous four-switch Buck-Boost power supply chip LM5175 operates as follows: When the input voltage is lower than the output voltage, the chip operates in Boost mode, controlling Q1 to be normally closed; when the LM5175 controls Q2 to open and Q4 to close, the current path is Q1→L1→Q4→GND, charging inductor L1 to store energy; when the LM5175 controls Q4 to open and Q2 to close, since the current in the inductor will not change abruptly, a back electromotive force (EMF) with the left side negative and the right side positive will be generated across the inductor to maintain the current magnitude. At this time, the inductor voltage is superimposed in series with the power supply voltage, thereby achieving the purpose of boosting the voltage. The function of this circuit is that when the resistance of the excitation coil in the magnetorheological damper increases with the increase of temperature, causing the output drive current under the original voltage to be insufficient, the voltage across it can be increased through the Boost circuit to achieve the required output drive current.

[0074] Among them, EN is used to enable the LM5175 chip. When the input voltage of the EN pin is >1.23V, the chip enters normal operating mode. The voltage divider R1 and R2 determine that the chip operates normally when the input voltage VBATJ ≥ 6V. MODE is used to set the operating mode of the LM5175 chip. When this pin is pulled down to GND through a 93.1kΩ resistor, the chip is in continuous conduction mode (CCM). The operating mode selected in this case is CCM. RT / SYNC is used to set the switching frequency of the chip. Considering that the higher the frequency, the lower the power supply output ripple, the switching frequency of the chip in this case is designed to be 500kHz. ;

[0075] Therefore, R6 is selected as 48.7kΩ; SS is used to set the soft-start time of the chip. In this case, the soft-start time is designed to be 16ms, which is determined by the formula... C3 is set to 0.1uF; C6 and R7 form an RC compensation network for Type II (PI) compensation; C5 adds a pole to the compensation network to suppress high-frequency noise; C7 provides slope compensation to stabilize the current in Boost mode.

[0076] Therefore, C7 is set to 880pF; ISN3(+) and ISNS(-) are used to set the average current limit, as shown in the formula IAGV. This design sets the average current limit IAGV to 20A, so R13 = 2.5mΩ; the output inductance is given by the formula...

[0077] Therefore, L1 is 3.3uH; in this design, the voltage is boosted to 30V, i.e., UBR = Vout = 30V, according to the formula... Since VFB = 0.8V, R9 is 750kΩ and R8 is 20kΩ. C8 and C10 are bootstrap capacitors, providing the required turn-on voltage for the upper-bridge MOSFET to discharge when it turns on. The maximum average current flowing through inductor L1 occurs under the conditions of minimum input voltage and maximum load current.

[0078] Based on the system requirements, Iout(max) is set to 20A, resulting in Imax = 74A; the sampling resistor between pins CS and CSG is obtained from...

[0079] With Rsense = 1.5mΩ, the N-channel MOSFET used in this case is Infineon IAUCN10S7N021, with a drain-source voltage of 100V, a continuous drain current of 220A, and an on-resistance of 2.1mΩ, which meets the functional requirements.

[0080] Reference Figure 5 In some embodiments, the H-bridge drive output circuit of the present invention adopts a four-switch H-bridge structure, and the current direction switching is achieved by controlling the conduction state of the four switching transistors, wherein:

[0081] When the first and fourth switches are turned on and the second and third switches are turned off, the current flows from the bus voltage through the first switch, the excitation coil, and the fourth switch to ground.

[0082] When the second and third switches are turned on and the first and fourth switches are turned off, the current flows from the bus voltage through the second switch, the excitation coil, and the third switch to ground.

[0083] Specifically, the working principle of the H-bridge drive output circuit is as follows: Figure 4 As shown, the microcontroller (MCU) controls the MOSFETs Q1-Q4 via a driver chip to achieve current flow in different directions. When MOSFETs Q1 and Q4 are closed and Q2 and Q3 are open, the current flow is UBR→Q1→OUT1→load→OUT2→Q4→GND. When the current is reversed, MOSFETs Q2 and Q3 are closed and Q1 and Q4 are open, and the current flow is UBR→Q2→OUT2→load→OUT1→Q3→GND.

[0084] In this embodiment, the driving scheme uses the TI DRV8244-Q1 automotive-grade H-bridge driver with integrated current sensing and feedback functions. The specific driving circuit design is as follows: Figure 6 As shown;

[0085] The DRV8244-Q1 driver chip is powered directly from a constant-voltage UBR supply. This allows it to adapt to magnetorheological dampers with varying internal resistances, ensuring that the input current covers a wide range of dampers with different resistances. This prevents the required output current from failing to meet standards due to supply voltage limitations when the coil's internal resistance increases. The chip integrates an internal current sensor; only an external sampling resistor is needed. The actual current is determined by the sensor's internal current acquisition mechanism. Where I: the actual current output of the H-bridge; VADC_Current: the acquired voltage; Rsense: the adapted current sampling resistor; 4750: the current scaling factor.

[0086] The truth table controlled by this chip is shown in Table 1.

[0087] Table 1 Control Truth Table

[0088] 0 X X X High resistance state High resistance state 1 1 0 0 High resistance state High resistance state 1 0 0 1 L H 1 0 1 0 H L 1 0 1 1 High resistance state High resistance state

[0089] like Figure 7 As shown, in some embodiments, the voltage sampling circuit of the present invention includes:

[0090] The first and second resistor voltage divider networks are used to attenuate the voltage between the two ends of the magnetorheological damper and ground.

[0091] A voltage follower, connected to the voltage divider point of the resistor divider network, is used to isolate the influence of downstream loads;

[0092] A voltage reference circuit is used to raise the reference level of the voltage follower to a preset potential that is compatible with the input range of the analog-to-digital converter of the microcontroller (MCU).

[0093] An RC filter circuit is used to filter out high-frequency switching noise generated by the drive circuit, and outputs a voltage signal that is compatible with the analog-to-digital converter of the microcontroller (MCU).

[0094] The working principle of the voltage sampling circuit at both ends of the magnetorheological damper is as follows: Figure 6As shown, a scheme using a resistor divider network + voltage follower + RC filter is adopted. First, the voltages at both ends of the magnetorheological damper to ground are attenuated to the range (0~3.3V) of the MCU's ADC through a resistor divider network composed of R40, R41 and R46, R47. In this case, the maximum voltage difference is designed to be 16V, and the resistor divider ratio is 11:1. Taking advantage of the extremely high input impedance and extremely low output impedance of the operational amplifier, a dual-channel operational amplifier LM2904 is used to isolate the fragile voltage divider point from the subsequent filtering and ADC sampling circuits. To ensure the voltage divider accuracy is unaffected by the downstream load, and considering the bidirectional control of the magnetorheological damper, the differential voltage may be negative. Therefore, the reference voltage needs to be raised. A voltage reference chip, REF2033, was used to raise the voltage reference from 0V to 1.65V. The magnetorheological damper's drive output is accompanied by PWM chopping, which generates significant high-frequency switching noise. This noise is removed by RC filtering to prevent the ADC from acquiring erratic values. The frequency to be filtered out is determined by f = 1 / 2πRC, ultimately yielding the voltage across the magnetorheological damper. Based on the collected voltage Vmrd across the magnetorheological damper and the current Imrd driving the magnetorheological damper, the internal resistance of the excitation coil of the magnetorheological damper can be monitored in real time using Ohm's law R=U / I.

[0095] Example 2

[0096] Please refer to Figure 8 This is a flowchart of the method of the present invention. Specifically, the present invention also proposes a temperature adaptive compensation control method for a magnetorheological damper, applied to the above-mentioned control circuit, which specifically includes the following steps:

[0097] S1: Real-time acquisition of the voltage value Vmrd across the magnetorheological damper and the current value Imrd flowing through the magnetorheological damper at the corresponding moment, and calculation of the internal resistance Rmrd of the current excitation coil;

[0098] S2: Obtain the pre-calibrated reference temperature T0 and the reference cold resistance R0 at the reference temperature. Based on the internal resistance Rmrd of the current excitation coil, the reference cold resistance R0, and the temperature coefficient of resistance α of the enameled wire, calculate the current temperature Tt of the magnetorheological damper. Specifically, in this embodiment, the magnetorheological damper is placed in a constant temperature test chamber and kept warm until thermally stable. The resistance R0 of the excitation coil in the magnetorheological damper is measured using a high-precision resistance tester, and the temperature T0 at this time is recorded.

[0099] S3: Based on the current temperature, calculate and output the compensated target driving current according to the preset magnetorheological damper damping force inverse model;

[0100] S4: Determine whether the current system bus voltage can meet the requirements of the target drive current output. If not, control the Boost circuit to increase the bus voltage.

[0101] S5: Control the H-bridge drive output circuit to drive the magnetorheological damper according to the target drive current.

[0102] Based on the above embodiments, in step S2 of this embodiment, the formula for calculating the current temperature of the magnetorheological damper is as follows:

[0103] Tt = T0 + (Rt - R0) / (α × R0)

[0104] Where Tt is the current temperature obtained by reverse calculation, T0 is the reference temperature, Rt is the internal resistance of the current excitation coil, R0 is the reference cold resistance at the reference temperature, and α is the temperature coefficient of resistance of the enameled wire inside the excitation coil.

[0105] Specifically, in this invention, T0 is the reference temperature of 25°C; R0 is the cold resistance at the reference temperature, which is known and determined through calibration; α is the temperature coefficient of resistance of the enameled wire in the excitation coil, which is a known parameter once the enameled wire is determined.

[0106] In an optional embodiment, step S1 of the present invention, the process of obtaining the voltage value across the magnetorheological damper, includes:

[0107] The voltage to ground at both ends of the magnetorheological damper is collected and attenuated by voltage division.

[0108] Perform differential calculation on the attenuated voltages at both ends, and superimpose a preset DC reference voltage on the differential calculation result;

[0109] The superimposed voltage signal is low-pass filtered to remove the high-frequency noise from the PWM chopping generated by the H-bridge drive. Specifically, in this embodiment, the noise frequency range is from 500kHz to 2MHz, to obtain the final voltage value.

[0110] In addition, a thermal protection determination step is included after step S2:

[0111] Determine whether the current temperature obtained by reverse calculation exceeds the preset safe temperature threshold;

[0112] If so, the thermal protection mode is triggered, forcibly reducing the amplitude of the target drive current to prevent the excitation coil from burning out or the oil seal from aging.

[0113] The above description is only a preferred embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any equivalent substitutions or modifications made by those skilled in the art within the scope of the technology disclosed in the present invention, based on the technical solution and inventive concept of the present invention, should be covered within the scope of protection of the present invention.

Claims

1. A temperature adaptive compensation control circuit for a magnetorheological damper, characterized in that, include: The power management control circuit, the boost circuit, the H-bridge drive output circuit, the voltage sampling circuit at both ends of the magnetorheological damper, and the microcontroller (MCU) are connected in sequence. The microcontroller (MCU) is configured as follows: The real-time current signal collected by the H-bridge drive output circuit and the real-time voltage signal collected by the voltage sampling circuit are obtained, and the current internal resistance value of the excitation coil is calculated. Based on the current internal resistance value, the reference internal resistance value, and the temperature coefficient of resistance of the enameled wire, the real-time temperature of the excitation coil is calculated in reverse to compensate for the temperature of the drive current, and the Boost circuit is controlled to boost the voltage when the required drive voltage is higher than the input voltage.

2. The temperature adaptive compensation control circuit for a magnetorheological damper according to claim 1, characterized in that: It also includes a power protection circuit, which includes: A surge suppression unit is connected in parallel to the power input terminal to absorb surge voltage; The reverse connection protection control unit includes an ideal diode controller and an N-channel MOSFET connected to the ideal diode controller. The ideal diode controller detects the voltage difference between the source and drain of the N-channel MOSFET, and turns off the N-channel MOSFET when a reverse voltage is detected to achieve reverse connection protection. When the MOSFET is forward-biased, the controller adjusts the gate voltage to maintain the N-channel MOSFET in a low on-state voltage drop.

3. The temperature adaptive compensation control circuit for a magnetorheological damper according to claim 2, characterized in that: It also includes a power management circuit, which includes: The Buck step-down circuit has its input terminal connected to the output terminal of the power protection circuit, and is used to step down the vehicle power supply voltage to the first intermediate voltage. A low-dropout linear regulator, whose input is connected to the output of the Buck step-down circuit, is used to step down the first intermediate voltage to a constant low-voltage power supply for the microcontroller MCU and peripheral chips.

4. The temperature adaptive compensation control circuit for a magnetorheological damper according to claim 1, characterized in that: The Boost circuit includes a synchronous four-switch Buck-Boost power chip that operates in continuous conduction mode. Its switching frequency and output voltage are configured to meet the drive voltage requirements of the excitation coil at the highest operating temperature. The Boost circuit also includes an output inductor connected to the synchronous four-switch Buck-Boost power chip, a capacitor for current slope compensation, and a resistor-capacitor network for type II compensation.

5. The temperature adaptive compensation control circuit for a magnetorheological damper according to claim 1, characterized in that: The H-bridge drive output circuit adopts a four-switch H-bridge structure, which achieves current direction switching by controlling the conduction state of the four switching transistors, wherein: When the first and fourth switches are turned on and the second and third switches are turned off, the current flows from the bus voltage through the first switch, the excitation coil, and the fourth switch to ground. When the second and third switches are turned on and the first and fourth switches are turned off, the current flows from the bus voltage through the second switch, the excitation coil, and the third switch to ground.

6. The temperature adaptive compensation control circuit for a magnetorheological damper according to claim 1, characterized in that: The voltage sampling circuit includes: The first and second resistor voltage divider networks are used to attenuate the voltage between the two ends of the magnetorheological damper and ground. A voltage follower, connected to the voltage divider point of the resistor divider network, is used to isolate the influence of downstream loads; A voltage reference circuit is used to raise the reference level of the voltage follower to a preset potential that is compatible with the input range of the analog-to-digital converter of the microcontroller (MCU). An RC filter circuit is used to filter out high-frequency switching noise generated by the drive circuit, and outputs a voltage signal that is compatible with the analog-to-digital converter of the microcontroller (MCU).

7. A temperature adaptive compensation control method for a magnetorheological damper, characterized in that, The control circuit applied to any one of claims 1 to 6 comprises the following steps: S1: Real-time acquisition of the voltage value across the magnetorheological damper and the current value flowing through the magnetorheological damper at the corresponding moment, and calculation of the internal resistance of the current excitation coil; S2: Obtain the pre-calibrated reference temperature and the reference cold resistance at the reference temperature, and calculate the current temperature of the magnetorheological damper based on the internal resistance of the current excitation coil, the reference cold resistance, and the temperature coefficient of resistance of the enameled wire. S3: Based on the current temperature, calculate and output the compensated target driving current according to the preset magnetorheological damper damping force inverse model; S4: Determine whether the current system bus voltage can meet the requirements of the target drive current output. If not, control the Boost circuit to increase the bus voltage. S5: Control the H-bridge drive output circuit to drive the magnetorheological damper according to the target drive current.

8. The temperature adaptive compensation control method for a magnetorheological damper according to claim 7, characterized in that: In step S2, the formula for calculating the current temperature of the magnetorheological damper is as follows: Tt = T0 + (Rt - R0) / (α × R0) Where Tt is the current temperature obtained by reverse calculation, T0 is the reference temperature, Rt is the internal resistance of the current excitation coil, R0 is the reference cold resistance at the reference temperature, and α is the temperature coefficient of resistance of the enameled wire inside the excitation coil.

9. The temperature adaptive compensation control method for a magnetorheological damper according to claim 7, characterized in that: In step S1, the process of obtaining the voltage value across the magnetorheological damper includes: The voltage to ground at both ends of the magnetorheological damper is collected and attenuated by voltage division. Perform differential calculation on the attenuated voltages at both ends, and superimpose a preset DC reference voltage on the differential calculation result; The superimposed voltage signal is low-pass filtered to remove the high-frequency noise from the PWM chopping generated by the H-bridge drive, thus obtaining the final voltage value.

10. The temperature adaptive compensation control method for a magnetorheological damper according to claim 7, characterized in that: Following step S2, a thermal protection determination step is also included: Determine whether the current temperature obtained by reverse calculation exceeds the preset safe temperature threshold; If so, the thermal protection mode is triggered, forcibly reducing the amplitude of the target drive current to prevent the excitation coil from burning out or the oil seal from aging.

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