Converter control method, control terminal, and storage medium

Abstract

The application provides a converter control method, a control terminal and a storage medium, and relates to the technical field of power electronics. The method is applied to a bidirectional CLLC resonant isolation DC-AC converter; the method comprises the following steps: establishing a full-cycle resonant current model according to operation parameters at a current time; constructing a control optimization model with a power deviation of 0 as a constraint condition and a current effective value minimum as an objective function according to the full-cycle resonant current model; and solving the control optimization model to obtain a phase-shifted duty ratio and a switching period. The application optimizes control by establishing a full-cycle resonant current model and taking the current effective value minimum as an objective, is more in line with actual working conditions, can effectively inhibit invalid circulating current in a resonant cavity, significantly reduces the conduction loss of a switching tube and the copper loss of a magnetic element, improves the accuracy of parameter design and control calculation, and enables the resonant converter to approach optimal efficiency operation in a full working condition and a wide load range, and improves working performance in a light load and a non-resonant point working condition.

Description

Technical Field

[0001] This invention relates to the field of power electronics technology, and in particular to a converter control method, control terminal, and storage medium. Background Technology

[0002] The bidirectional CLLC resonant isolated DC-AC converter directly applies the CLLC topology to a single-stage DC-AC converter, comprising a bidirectional CLLC resonant isolated DC-DC converter and a low-frequency expansion module connected in sequence. The bidirectional CLLC resonant isolated DC-DC converter consists of a primary-side high-frequency conversion circuit, a CLLC resonant network, a secondary-side high-frequency conversion circuit, and a high-frequency filter circuit. The DC-side port of the primary-side high-frequency conversion circuit is connected to a DC power supply and configured for bidirectional conversion between DC and high-frequency AC. The CLLC resonant network is connected between the primary-side and secondary-side high-frequency conversion circuits to achieve electrical isolation and energy transfer. The AC side of the secondary-side high-frequency conversion circuit is connected to the secondary-side output of the CLLC resonant network. The high-frequency filter circuit is connected in parallel across the DC side of the secondary-side high-frequency conversion circuit. The input of the low-frequency expansion circuit is connected across the high-frequency filter circuit and configured to perform polarity reversal or synchronous rectification of the pulsating DC bus voltage at the power frequency.

[0003] In existing technologies, the fundamental frequency approximation analysis method (FHA) is typically used for parameter design and gain calculation. However, since FHA only considers the fundamental frequency component and ignores the influence of higher harmonics, the theoretical and practical deviations are large when the frequency deviates significantly from the resonant frequency or under light load conditions. This leads to inaccurate calculation of control parameters, the presence of large ineffective circulating currents within the resonant cavity, increased conduction losses of the switching transistors and copper losses of the magnetic components, making it difficult to achieve optimal efficiency under all operating conditions. Summary of the Invention

[0004] This invention provides a converter control method, control terminal, and storage medium to solve the problems of inaccurate calculation of control parameters, large ineffective circulating current in the resonant cavity, and inability to achieve optimal efficiency under all operating conditions in the fundamental wave approximation analysis method.

[0005] In a first aspect, embodiments of the present invention provide a converter control method applied to a bidirectional CLLC resonant isolation DC-AC converter; the converter includes: a bidirectional CLLC resonant isolation DC-DC converter and a low-frequency expansion module connected in sequence; the method includes: Obtain the current operating parameters of the converter; Based on the operating parameters at the current moment, establish a full-cycle resonant current model; Based on the full-cycle resonant current model, a control optimization model is constructed with zero power deviation as the constraint and minimum effective current value as the objective function. Solving the control optimization model yields the target control parameters, which include the phase shift duty cycle and the switching period.

[0006] In a second aspect, embodiments of the present invention provide a control terminal, including a memory and a processor. The memory stores a computer program, and the processor executes the computer program to implement the converter control method as described in the first aspect or any possible implementation of the first aspect.

[0007] Thirdly, embodiments of the present invention provide a computer-readable storage medium storing a computer program that, when executed by a processor, implements the converter control method as described in the first aspect or any possible implementation thereof.

[0008] This invention provides a converter control method, a control terminal, and a storage medium. The converter control method is applied to a bidirectional CLLC resonant isolation DC-AC converter. The converter includes a bidirectional CLLC resonant isolation DC-DC converter and a low-frequency expansion module connected in sequence. The method includes: acquiring the converter's current operating parameters; establishing a full-cycle resonant current model based on the current operating parameters; constructing a control optimization model based on the full-cycle resonant current model, with a power deviation of 0 as a constraint and the minimum effective current value as the objective function; solving the control optimization model to obtain the target control parameters; wherein the target control parameters include: phase shift duty cycle and switching period. This invention, by establishing a full-cycle resonant current model and optimizing control with the minimum effective current value as the objective, better reflects actual operating conditions. It can effectively suppress ineffective circulating currents within the resonant cavity, significantly reduce the conduction losses of the switching transistors and the copper losses of the magnetic components, improve the accuracy of parameter design and control calculations, and enable the resonant converter to approach optimal efficiency operation across all operating conditions and a wide load range, improving its performance under light load and non-resonant point conditions. Attached Figure Description

[0009] Figure 1 This is a schematic diagram of the circuit structure of a bidirectional CLLC resonant isolation DC-AC converter provided in an embodiment of the present invention; Figure 2 This is a flowchart illustrating the implementation of the converter control method provided in this embodiment of the invention. Figure 3 This is a target control parameter transformation trajectory diagram provided in an embodiment of the present invention; Figure 4 These are the forward AC side voltage and current waveforms when applying the converter control method provided in the embodiments of the present invention; Figure 5 It is the harmonic distortion rate of the positive alternating current when the converter control method provided in the embodiments of the present invention is applied; Figure 6 These are the reverse AC side voltage and current waveforms when applying the converter control method provided in the embodiments of the present invention; Figure 7 It is the harmonic distortion rate of the reverse alternating current when applying the converter control method provided in the embodiments of the present invention; Figure 8 This is a schematic diagram of the converter control device provided in an embodiment of the present invention; Figure 9 This is a schematic diagram of the control terminal provided in an embodiment of the present invention. Detailed Implementation

[0010] The embodiments of the present invention will now be described in detail with reference to the accompanying drawings.

[0011] Figure 1 A schematic diagram of the circuit structure of a bidirectional CLLC resonant isolated DC-AC converter provided in an embodiment of the present invention. Figure 1 As shown, the bidirectional CLLC resonant isolated DC-AC converter includes: a bidirectional CLLC resonant isolated DC-DC converter 1 and a low-frequency expansion module 2 connected in sequence.

[0012] The bidirectional CLLC resonant isolated DC-DC converter 1 includes: a primary-side high-frequency conversion circuit 11, a CLLC resonant network 12, a secondary-side high-frequency conversion circuit 13, and a high-frequency filter circuit 14.

[0013] The primary-side high-frequency conversion circuit 11 has a DC-side port for connecting to a DC power supply. The primary-side high-frequency conversion circuit 11 is configured to convert DC power to high-frequency AC power or high-frequency AC power to DC power.

[0014] The primary-side high-frequency converter circuit 11 adopts a full-bridge topology, including a first bridge arm and a second bridge arm connected in parallel to the DC side port; The first bridge arm consists of a first switch (Q) connected in series. 1a ) and the second switching transistor (Q) 1b The second bridge arm consists of a third switch (Q) connected in series. 2a ) and the fourth switch (Q) 2b )composition; The AC side port of the primary-side high-frequency conversion circuit 11 is drawn from the midpoint of the first bridge arm and the midpoint of the second bridge arm.

[0015] CLLC resonant network 12, whose primary input is connected to the AC side port of primary high-frequency conversion circuit 11; CLLC resonant network 12 includes a primary resonant inductor ( L 1) Primary resonant capacitor ( C 1) Magnetizing inductance ( L m ), High-frequency isolation transformer (HFT), Secondary resonant inductor (L 2) and secondary resonant capacitor ( C 2); Primary resonant inductor ( L 1) Connected between the first AC output terminal of the primary-side high-frequency conversion circuit 11 and the first terminal of the primary winding of the high-frequency isolation transformer (HFT); Primary resonant capacitor ( C 1) Connected between the second AC output terminal of the primary-side high-frequency conversion circuit 11 and the second terminal of the primary winding of the high-frequency isolation transformer (HFT); Magnetizing inductor ( L m It is connected in parallel between the first end of the primary winding and the second end of the primary winding of the high-frequency isolation transformer (HFT); Secondary resonant inductor ( L 2) Connected between the first terminal of the secondary winding of the high-frequency isolation transformer (HFT) and the first AC input terminal of the secondary high-frequency conversion circuit 13; Secondary resonant capacitor ( C 2) Connected between the second terminal of the secondary winding of the high-frequency isolation transformer (HFT) and the second AC input terminal of the secondary high-frequency conversion circuit 13.

[0016] Among them, the primary resonant inductor ( L 1) The inductance value and the secondary resonant inductance ( L 2) The ratio of inductance values, and the secondary resonant capacitance ( C 2) The capacitance value and the primary resonant capacitance ( C The ratio of the capacitance values ​​of 1) is equal to the square of the turns ratio of the high-frequency isolation transformer (HFT).

[0017] The secondary high-frequency conversion circuit 13 has its AC side port connected to the secondary output terminal of the CLLC resonant network 12; The secondary high-frequency converter circuit 13 also adopts a full-bridge topology, including a third bridge arm and a fourth bridge arm connected in parallel across the high-frequency filter circuit 14; the third bridge arm consists of a fifth switch (Q) connected in series. 3a ) and the sixth switch (Q) 3b The fourth bridge arm consists of a seventh switch (Q) connected in series. 4a ) and the eighth switch (Q) 4b The secondary high-frequency conversion circuit 13 is composed of the AC side ports of the secondary side, which are derived from the midpoint of the third bridge arm and the midpoint of the fourth bridge arm.

[0018] The high-frequency filter circuit 14 is connected in parallel to both ends of the DC side port of the secondary high-frequency conversion circuit 13 to filter out high-frequency switching ripple to form a pulsating DC bus voltage. The high-frequency filter circuit 14 includes a high-frequency filter capacitor ( C hf), high-frequency filter capacitor ( C hf The capacitance value is configured to filter out voltage ripple at the switching frequency only, so that the voltage across it has a sinusoidal absolute value envelope waveform (i.e., pulsating DC).

[0019] The low-frequency expansion module 2 has its input terminal connected to both ends of the high-frequency filter circuit 14. The low-frequency expansion module 2 is configured to perform polarity reversal or synchronous rectification of the pulsating DC bus voltage at the power frequency. The low-frequency expansion module 2 adopts a full-bridge topology, including a fifth and sixth bridge arm connected in parallel across the high-frequency filter circuit 14; the fifth bridge arm consists of a ninth switch (Q) connected in series. 5a ) and the tenth switch (Q) 5b The sixth bridge arm consists of an eleventh switch (Q) connected in series. 6a ) and the twelfth switch (Q) 6b It consists of ) and the switching frequency of the switching transistor in the low-frequency expansion module 2 is synchronized with the fundamental frequency of the AC power grid.

[0020] The output filter circuit is connected between the output of the low-frequency expansion module 2 and the AC power grid to filter out high-frequency harmonics.

[0021] See Figure 2 The diagram illustrates a flowchart of a converter control method provided by an embodiment of the present invention, which is described in detail below: The above converter control method is applied to Figure 1 The bidirectional CLLC resonant isolated DC-AC converter shown; Reference Figure 1 The converter includes: a bidirectional CLLC resonant isolated DC-DC converter and a low-frequency expansion module connected in sequence; the above method includes: S101: Obtain the current operating parameters of the converter; For example, the input voltage of a bidirectional CLLC resonant isolated DC-DC converter ( u in ) and output voltage ( u out AC side voltage ( u ac ) and current ( i ac ), resonant cavity voltage and current, etc.

[0022] S102: Establish a full-cycle resonant current model based on the current operating parameters; The modal decoupling method based on CLLC resonant cavity decomposes the fourth-order coupled circuit into two independent second-order linear systems: common-mode and differential-mode. By establishing the time-domain state-space differential equation, solving the state transition matrix, and substituting the steady-state boundary conditions, a full-cycle resonant current model containing higher harmonics is derived, replacing the traditional fundamental wave approximation method (FHA) and achieving high-precision modeling.

[0023] In one possible implementation, S102 may include: S1021: Based on the operating parameters at the current moment, establish the state-space differential equations; wherein, the state-space differential equations include: common-mode equations and differential-mode equations; This application introduces a common-mode to differential-mode decoupling transformation matrix to linearly transform the originally strongly coupled fourth-order circuit system into two completely independent, uncoupled second-order linear subsystems, thereby halving the system order and completely eliminating the coupling relationship.

[0024] The common-mode equation describes the reactive power exchange process of the excitation branch in the resonant cavity, which corresponds to the charging and discharging behavior of the transformer excitation inductance and directly determines the magnitude of the reactive circulating current in the resonant cavity and the soft-switching characteristics of the switching transistor. The differential mode equation describes the power transfer process between the primary and secondary sides of the resonant cavity, and directly determines the power transfer accuracy and gain characteristics of the converter.

[0025] The state-space differential equations are completely linear, uncoupled, and without approximations, thus fully preserving the higher harmonic components of the resonant current.

[0026] In one possible implementation, the common-mode equations may include:

[0027]

[0028] Differential mode equations may include:

[0029] in, and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. For magnetizing inductance; and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at all times; and They are respectively Common-mode voltage and common-mode current at time _____. and They are respectively Differential-mode voltage and differential-mode current at time t; and They are respectively The current flowing through the primary resonant inductor and the current flowing through the secondary resonant inductor at all times; and They are respectively The primary capacitor voltage and the secondary capacitor voltage at time t.

[0030] This application decomposes a fourth-order strongly coupled system into two independent second-order subsystems, halving the system order and exponentially reducing the difficulty of solving the differential equations. Furthermore, the state-space differential equations completely preserve the higher harmonic components of the resonant current, exhibiting minimal errors even under extreme conditions of light load, wide gain, and deviation from the resonant frequency.

[0031] It should be noted that, The current flowing from the primary-side high-frequency converter circuit to the primary-side resonant inductor is defined as positive. The current flowing from the secondary resonant inductor to the secondary high-frequency conversion circuit is defined as positive; Defined as the primary capacitor voltage calculated with the primary-side high-frequency conversion circuit as the positive terminal. Defined as the secondary capacitor voltage calculated with the secondary high-frequency conversion circuit as the negative terminal.

[0032] S1022: Determine the state transition matrix based on the state-space differential equation; The state transition matrix describes the time-domain evolution of the system's state variables within a switching mode, from the initial time to any other time.

[0033] This application addresses the decoupled common-mode and differential-mode independent second-order linear subsystems by solving their state-space differential equations in the time domain, yielding a general time-domain solution based on trigonometric functions. Based on this solution, a block-diagonal state transition matrix is ​​constructed. The two second-order block matrices on the diagonal correspond to the state evolution of the common-mode and differential-mode, respectively. Each block matrix contains trigonometric function terms related to the resonant angular frequency and characteristic impedance of the corresponding mode, comprehensively describing the changes in state variables for each mode within a switching cycle. A system of linear equations can be directly constructed using the state transition matrix, and the initial state vector can be solved to reconstruct the full-cycle resonant current model.

[0034] In one possible implementation, S1022 may include: 1. Solve the state-space differential equations to obtain the common mode matrix exponent and the differential mode matrix exponent; The common-mode equation describes the reactive power exchange process of the excitation branch in a CLLC resonant cavity, directly determining the magnitude of the reactive circulating current, the soft-switching range of the switching transistor, and the peak / RMS value of the resonant current. Solving the common-mode equation yields the common-mode matrix exponent, which can accurately quantify the time-domain oscillation characteristics of the excitation branch, providing a calculation basis for minimizing the RMS value of the resonant current.

[0035] The differential-mode equation describes the active power transfer process between the primary and secondary sides of a CLLC resonant cavity, directly determining the converter's power transfer gain, output power accuracy, and wide-gain adjustment capability. Solving the differential-mode equation yields the differential-mode matrix exponent, which can accurately quantify the time-domain transmission law of active power, providing a calculation basis for achieving zero power deviation.

[0036] 2. Obtain the state transition matrix based on the common mode matrix exponent and the differential mode matrix exponent; The state transition matrix can include:

[0037]

[0038]

[0039]

[0040] in, for The state transition matrix at time t, and They are respectively Two block diagonal matrices at time 1; and These are the common-mode resonant angular frequency and the differential-mode resonant angular frequency, respectively. and These are the common-mode characteristic impedance and the differential-mode characteristic impedance, respectively. and They are respectively The common mode matrix exponent and the differential mode matrix exponent at time t; and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. It is the magnetizing inductor.

[0041] Since the common-mode and differential-mode subsystems are completely decoupled, the state transition matrix of the entire CLLC fourth-order system is a standard block diagonal matrix; the two second-order blocks on the diagonal correspond to the common-mode matrix exponent and differential-mode matrix exponent obtained in the first step, respectively. The final state transition matrix can completely describe the time-domain evolution of the system's state variables in the switching modes.

[0042] S1023: Based on the state transition matrix, the full-cycle resonant current model is obtained.

[0043] In one possible implementation, S1023 may include: 1. Construct an initial state vector based on the current operating parameters; 2. Construct a full-cycle resonant current model based on the initial state vector and the state transition matrix.

[0044] To obtain the full-cycle resonant current during stable operation of the converter, the initial state vector at the beginning of the switching cycle must first be solved. Then, using the initial state vector as the initial boundary and the state transition matrix as the dynamic evolution rule, a continuous time-domain analytical solution of the resonant current over a complete switching cycle is obtained through linear recursion. Finally, a full-cycle resonant current model without any fundamental approximations is constructed.

[0045] In one possible implementation, the initial state vector may include:

[0046] in, for The initial state vector at time t. and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at all times; and They are respectively Common-mode voltage and common-mode current at time _____. and They are respectively Differential-mode voltage and differential-mode current at time t; The full-cycle resonant current model can include:

[0047]

[0048]

[0049] in, It is a full-cycle resonant current. This is the port excitation voltage related vector. It is the identity matrix. This represents the total state transition matrix within half a switching cycle; and They are respectively The input voltage and output voltage of the bidirectional CLLC resonant isolated DC-DC converter in the next switching mode adjacent to the current switching mode.

[0050] Based on the modal cascade relationship within the switching cycle and the half-cycle odd-symmetric steady-state characteristics of the CLLC symmetric topology, a system of linear equations about the initial state vector is constructed and solved to obtain the initial state vector, thereby reconstructing the full-cycle resonant current.

[0051] S103: Based on the full-cycle resonant current model, construct a control optimization model with zero power deviation as the constraint and minimum effective current value as the objective function. Using zero power deviation as an equality constraint, the output power strictly tracks the command power; using the minimum effective value of current as the objective function, the circulating current loss, copper loss, and conduction loss are minimized; the phase shift duty cycle and switching cycle are used as optimization variables to form a constrained nonlinear optimization problem.

[0052] In one possible implementation, the objective function may include:

[0053] Constraints may include:

[0054]

[0055] in, For the switching cycle, For phase shift duty cycle, Average output power, For power deviation, For target power command, for The resonant cavity port voltage at time ( u p ), It is the full-cycle resonant current.

[0056] S104: Solve the control optimization model to obtain the target control parameters; among which, the target control parameters include: phase shift duty cycle and switching period.

[0057] The control optimization model is solved to obtain the phase shift duty cycle and switching period. Then, a PWM drive signal is generated based on the phase shift duty cycle and switching period to control the operation of the converter's switching transistors.

[0058] In one possible implementation, S104 may include: S1041: Solve the control optimization model using a nonlinear programming algorithm or table lookup interpolation method to obtain the target control parameters.

[0059] The lookup table interpolation method involves traversing all operating conditions, using the converter control method provided in this application to obtain the optimal solution, and generating a lookup table (LUT) which is stored in the control terminal. During runtime, the target control parameters are obtained by looking up the table based on the real-time status. This method is mostly used in offline environments.

[0060] For online environments, nonlinear programming algorithms, such as Sequential Quadratic Programming (SQP), can be used to iteratively solve the control optimization model in real time to obtain the target control parameters. Nonlinear programming algorithms include, but are not limited to, Sequential Quadratic Programming; the specific algorithm chosen depends on the specific application requirements.

[0061] The target control parameters are solved using the converter control method provided in this application. The trajectories of the target control parameters under half-load and full-load conditions are as follows: Figure 3 As shown, the THD measurement results of the alternating current are as follows: Figures 4-7 As shown. By Figures 4-7 It is known that the converter control method provided in this application can accurately control the waveform, effectively improve power quality and increase energy conversion efficiency. In bidirectional power flow mode, the total harmonic distortion (THD) of the converter's AC output current can be controlled within 3.5%, and the overall peak conversion efficiency remains above 98%, verifying the effectiveness of the strategy in this application.

[0062] This application leverages the symmetry of CLLC topology, introducing a decoupling transformation matrix to decompose the complex fourth-order system into two second-order subsystems: common-mode and differential-mode. A time-domain analytical model (full-cycle resonant current model) without approximations is established, fully preserving higher harmonics and addressing the large errors of traditional fundamental frequency analysis methods. Under conditions of light load, wide gain, and distance from the resonant frequency, the model error is significantly reduced. Simultaneously, with power tracking as a constraint and minimizing the effective value of the resonant current as the objective, the phase shift duty cycle and switching period are optimized simultaneously. This fundamentally suppresses ineffective circulating current, reduces copper losses in the resonant cavity and conduction losses in the switching transistors, achieving minimum circulating current loss and fully soft switching across the entire operating range. The optimal phase shift duty cycle and switching period are obtained, ensuring the converter always operates at the optimal efficiency point under the current conditions, thus improving overall system efficiency.

[0063] It should be understood that the sequence number of each step in the above embodiments does not imply the order of execution. The execution order of each process should be determined by its function and internal logic, and should not constitute any limitation on the implementation process of the embodiments of the present invention.

[0064] The following are device embodiments of the present invention. For details not described in detail, please refer to the corresponding method embodiments described above.

[0065] Figure 8A schematic diagram of the converter control device provided in an embodiment of the present invention is shown. For ease of explanation, only the parts related to the embodiment of the present invention are shown, and are described in detail below: like Figure 8 As shown, the converter control device is applied to a bidirectional CLLC resonant isolated DC-AC converter; the converter includes: a bidirectional CLLC resonant isolated DC-DC converter and a low-frequency expansion module connected in sequence; the converter control device includes: The parameter acquisition module 21 is used to acquire the current operating parameters of the converter; The current model establishment module 22 is used to establish a full-cycle resonant current model based on the operating parameters at the current moment. The optimization model building module 23 is used to construct a control optimization model based on the full-cycle resonant current model, with the power deviation being 0 as a constraint and the minimum effective value of the current as the objective function. The model solving module 24 is used to solve the control optimization model to obtain the target control parameters; among which, the target control parameters include: phase shift duty cycle and switching period.

[0066] In one possible implementation, the current model building module 22 may include: The differential equation establishment unit is used to establish state-space differential equations based on the operating parameters at the current moment; wherein, the state-space differential equations include: common mode equations and differential mode equations; The state transition matrix establishment unit is used to determine the state transition matrix based on the state-space differential equation. The current model establishment unit is used to obtain the full-cycle resonant current model based on the state transition matrix.

[0067] In one possible implementation, the common-mode equations may include:

[0068]

[0069] Differential mode equations may include:

[0070] in, and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. For magnetizing inductance; and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at all times; and They are respectively Common-mode voltage and common-mode current at time _____. and They are respectively Differential-mode voltage and differential-mode current at time t; and They are respectively The current flowing through the primary resonant inductor and the current flowing through the secondary resonant inductor at all times; and They are respectively The primary capacitor voltage and the secondary capacitor voltage at time t.

[0071] In one possible implementation, the state transition matrix establishment unit can be specifically used for: 1. Solve the state-space differential equations to obtain the common mode matrix exponent and the differential mode matrix exponent; 2. Obtain the state transition matrix based on the common mode matrix exponent and the differential mode matrix exponent; The state transition matrix can include:

[0072]

[0073]

[0074]

[0075] in, for The state transition matrix at time t, and They are respectively Two block diagonal matrices at time 1; and These are the common-mode resonant angular frequency and the differential-mode resonant angular frequency, respectively. and These are the common-mode characteristic impedance and the differential-mode characteristic impedance, respectively. and They are respectively The common mode matrix exponent and the differential mode matrix exponent at time t; and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. It is the magnetizing inductor.

[0076] In one possible implementation, the current model building unit can be specifically used for: 1. Construct an initial state vector based on the current operating parameters; 2. Construct a full-cycle resonant current model based on the initial state vector and the state transition matrix.

[0077] In one possible implementation, the initial state vector may include:

[0078] in, for The initial state vector at time t. and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at all times; and They are respectively Common-mode voltage and common-mode current at time _____. and They are respectively Differential-mode voltage and differential-mode current at time t; The full-cycle resonant current model can include:

[0079]

[0080]

[0081] in, It is a full-cycle resonant current. This is the port excitation voltage related vector. It is the identity matrix. This represents the total state transition matrix within half a switching cycle; and They are respectively The input voltage and output voltage of the bidirectional CLLC resonant isolated DC-DC converter in the next switching mode adjacent to the current switching mode.

[0082] In one possible implementation, the objective function may include:

[0083] Constraints may include:

[0084]

[0085] in, For the switching cycle, For phase shift duty cycle, Average output power, For power deviation, For target power command, for The resonant cavity port voltage at time t. It is the full-cycle resonant current.

[0086] In one possible implementation, the model solving module 24 may include: The solver unit is used to solve the control optimization model using nonlinear programming algorithms or lookup table interpolation methods to obtain the target control parameters.

[0087] Figure 9 This is a schematic diagram of the control terminal provided in an embodiment of the present invention. Figure 9 As shown, the control terminal 3 in this embodiment includes a processor 30 and a memory 31. The memory 31 stores a computer program 32. When the processor 30 executes the computer program 32, it implements the steps in the various method embodiments described above. Alternatively, when the processor 30 executes the computer program 32, it implements the functions of each module / unit in the various device embodiments described above.

[0088] For example, computer program 32 can be divided into one or more modules / units, which are stored in memory 31 and executed by processor 30 to complete the present invention. The one or more modules / units can be a series of computer program instruction segments capable of performing specific functions, which describe the execution process of computer program 32 in control terminal 3.

[0089] The control terminal 3 may include, but is not limited to, a processor 30 and a memory 31. Those skilled in the art will understand that... Figure 9 This is merely an example of control terminal 3 and does not constitute a limitation on control terminal 3. It may include more or fewer components than shown, or combine certain components, or different components. For example, control terminal 3 may also include input / output devices, network access devices, buses, etc.

[0090] The processor 30 can be a central processing unit (CPU), or other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. The general-purpose processor can be a microprocessor or any conventional processor.

[0091] The memory 31 can be an internal storage unit of the control terminal 3, such as a hard disk or RAM of the control terminal 3. The memory 31 can also be an external storage device of the control terminal 3, such as a plug-in hard disk, smart media card (SMC), secure digital (SD) card, flash card, etc., equipped on the control terminal 3. Furthermore, the memory 31 can include both internal and external storage units of the control terminal 3. The memory 31 is used to store the computer program 32 and other programs and data required by the control terminal 3. The memory 31 can also be used to temporarily store data that has been output or will be output.

[0092] For the sake of simplicity and clarity, only the above-described functional modules / units are used as examples. In practical applications, the functions described above can be assigned to different functional modules / units as needed. These modules / units can be implemented in hardware, software, or a combination of both.

[0093] This invention also provides a computer-readable storage medium storing a computer program. When the computer program is executed by a processor, it implements the methods described in the above-described method embodiments.

[0094] This invention also provides a computer program product, including a computer program. When the computer program is executed by a processor, it implements the methods described in the above-described method embodiments.

[0095] Computer programs include computer program code, which can be in the form of source code, object code, executable files, or certain intermediate forms. Computer-readable media can include: any entity or device capable of carrying computer program code, recording media, USB flash drives, portable hard drives, magnetic disks, optical disks, computer memory, read-only memory (ROM), random access memory (RAM), electrical carrier signals, telecommunication signals, and software distribution media, etc.

[0096] In the above embodiments, the descriptions of each embodiment have their own emphasis. Parts not detailed or described in a particular embodiment can be referred to in the relevant descriptions of other embodiments. Unless otherwise specified or in conflict with logic, the terminology and / or descriptions between different embodiments are consistent and can be referenced interchangeably. Technical features in different embodiments can be combined to form new embodiments based on their inherent logical relationships.

[0097] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention, and should all be included within the protection scope of the present invention.

Claims

1. A converter control method, characterized in that, Applied to a bidirectional CLLC resonant isolation DC-AC converter; the converter includes: a bidirectional CLLC resonant isolation DC-DC converter and a low-frequency expansion module connected in sequence; the method includes: Obtain the current operating parameters of the converter; Based on the operating parameters at the current moment, establish a full-cycle resonant current model; Based on the full-cycle resonant current model, a control optimization model is constructed with zero power deviation as the constraint and minimum effective current value as the objective function. The target control parameters are obtained by solving the control optimization model; wherein the target control parameters include: phase shift duty cycle and switching period.

2. The converter control method according to claim 1, characterized in that, The step of establishing a full-cycle resonant current model based on the operating parameters at the current moment includes: Based on the operating parameters at the current moment, a state-space differential equation is established; wherein, the state-space differential equation includes: common-mode equation and differential-mode equation; Determine the state transition matrix based on the state-space differential equation; The full-cycle resonant current model is obtained based on the state transition matrix.

3. The converter control method according to claim 2, characterized in that, The common-mode equations include: The differential mode equations include: in, and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. For magnetizing inductance; and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at the specified time. and They are respectively Common-mode voltage and common-mode current at time 1. and They are respectively Differential-mode voltage and differential-mode current at time t; and They are respectively The current flowing through the primary resonant inductor and the current flowing through the secondary resonant inductor at all times; and They are respectively The primary capacitor voltage and the secondary capacitor voltage at time t.

4. The converter control method according to claim 2, characterized in that, Determining the state transition matrix based on the state-space differential equation includes: Solving the state-space differential equation yields the common mode matrix exponent and the differential mode matrix exponent. The state transition matrix is ​​obtained based on the common mode matrix index and the differential mode matrix index; The state transition matrix includes: in, for The state transition matrix at time t, and They are respectively Two block diagonal matrices at time 1; and These are the common-mode resonant angular frequency and the differential-mode resonant angular frequency, respectively. and These are the common-mode characteristic impedance and the differential-mode characteristic impedance, respectively. and They are respectively The common mode matrix index and the differential mode matrix index at time t; and These are the equivalent resonant inductance and capacitance referred to the primary side of the transformer, respectively. It is the magnetizing inductor.

5. The converter control method according to claim 2, characterized in that, The step of obtaining the full-cycle resonant current model based on the state transition matrix includes: Construct an initial state vector based on the operating parameters at the current moment; A full-cycle resonant current model is constructed based on the initial state vector and the state transition matrix.

6. The converter control method according to claim 5, characterized in that, The initial state vector includes: in, for The initial state vector at time t, and They are respectively The input and output voltages of the bidirectional CLLC resonant isolated DC-DC converter at the specified time. and They are respectively Common-mode voltage and common-mode current at time 1. and They are respectively Differential-mode voltage and differential-mode current at time t; The full-cycle resonant current model includes: in, This refers to the full-cycle resonant current. This is the port excitation voltage related vector. It is the identity matrix. This is the total state transition matrix within half of the switching cycle; and They are respectively The input voltage and output voltage of the next switching mode adjacent to the switching mode of the bidirectional CLLC resonant isolated DC-DC converter at the given time.

7. The converter control method according to any one of claims 1 to 6, characterized in that, The objective function includes: The constraints include: in, The switching period is... The phase shift duty cycle is... Average output power, The power deviation is... For target power command, for The resonant cavity port voltage at time t. This refers to the full-cycle resonant current.

8. The converter control method according to any one of claims 1 to 6, characterized in that, Solving the control optimization model to obtain the target control parameters includes: The target control parameters are obtained by solving the control optimization model using a nonlinear programming algorithm or a lookup table interpolation method.

9. A control terminal, characterized in that, It includes a memory and a processor, the memory storing a computer program, and the processor executing the computer program to implement the method as described in any one of claims 1 to 8.

10. A computer-readable storage medium, characterized in that, The computer-readable storage medium stores a computer program that, when executed by a processor, implements the method as described in any one of claims 1 to 8.

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