Transmitting circuit CIM3 and CIM5 elimination method and transmitting circuit

Abstract

The application discloses a transmitting circuit CIM3 and CIM5 elimination method and a transmitting circuit. The method comprises the following steps: in a first processing path, performing zero-degree phase shift processing on a baseband signal and performing zero-degree phase shift processing on a local oscillator signal, performing up-conversion mixing on the processed baseband signal and the processed local oscillator signal, and generating a first radio frequency signal; in a second processing path, performing 45-degree phase shift processing on the baseband signal and performing negative 45-degree phase shift processing on the local oscillator signal, performing up-conversion mixing on the processed baseband signal and the processed local oscillator signal, and generating a second radio frequency signal; performing power synthesis on the first radio frequency signal and the second radio frequency signal, and outputting a radio frequency signal; wherein the CIM3 component and the CIM5 component in the first radio frequency signal and the second radio frequency signal are opposite in phase and equal in amplitude when being synthesized, and thus are mutually cancelled. Some embodiments provided by the application can make the signal distortion components in the transmitting circuit be opposite and cancelled, and avoid the signal attenuation problem that may be introduced in the traditional method.

Description

Technical Field

[0001] This invention belongs to the field of wireless communication technology, specifically relating to a method for suppressing nonlinear distortion in a transmitter, and more particularly to a method and corresponding transmitting circuit capable of simultaneously eliminating inverse third-order intermodulation distortion CIM3 (counter intermodulation 3) and inverse fifth-order intermodulation distortion CIM5 (counter intermodulation 5). Background Technology

[0002] In a wireless transmitter, the baseband signal is up-converted to radio frequency (RF) by a mixer after passing through the baseband signal processing circuit. Due to the nonlinear characteristics of the baseband signal processing circuit, the mixer, and subsequent amplifiers, various nonlinear distortion components are generated. Among them, CIM3 and CIM5 are important factors affecting the signal quality of the transmit link (such as error vector magnitude EVM). CIM3 is mainly manifested at a frequency of ω c The component of -3ω0, CIM5 is mainly characterized by a frequency of ω. c The +5ω0 components typically fall within the useful signal band or adjacent to the channel, making them difficult to filter out using traditional filters.

[0003] In existing technologies, a common method is to actively generate components in the radio frequency domain with equal amplitude but opposite phase to the baseband third harmonic for cancellation. This method typically requires three channels with specific phase differences (e.g., 0°, 45°, 90°) and specific amplitude weights (e.g., 1, 2, 3, 4, 5, 6, 7, 8, 9, 1 ... The baseband signal of 1) is used, and three sets of mixer circuits are employed. Although this method can simplify phase generation by utilizing the characteristics of a 25% duty cycle mixer, the overall circuit complexity is high, the amplitude matching requirements are stringent, and it cannot effectively handle the CIM5 component.

[0004] Therefore, there is an urgent need for a method to suppress nonlinear distortion of the transmitter circuit that can simultaneously eliminate CIM3 and CIM5 and has a simpler circuit structure.

[0005] It should be noted that the above introduction to the technical background is only for the purpose of providing a clear and complete explanation of the technical solutions of this application and facilitating understanding by those skilled in the art. It should not be assumed that these technical solutions are known to those skilled in the art simply because they have been described in the background section of this application. Summary of the Invention

[0006] In order to solve at least one of the above problems, and one or more other potential problems, for example, the present invention aims to provide a method for eliminating CIM3 and CIM5 in a transmitting circuit, so as to solve the problem that the prior art has complex circuits and cannot eliminate CIM3 and CIM5 at the same time.

[0007] To achieve the above objectives, the present invention provides the following technical solution:

[0008] In a first aspect, a method for eliminating CIM3 and CIM5 in a transmitting circuit is provided, comprising: providing a first processing path and a second processing path; in the first processing path, performing zero-degree phase shift processing on a baseband signal and zero-degree phase shift processing on a local oscillator signal, and up-mixing the processed baseband signal and the processed local oscillator signal to generate a first radio frequency (RF) signal; in the second processing path, performing 45-degree phase shift processing on a baseband signal and negative 45-degree phase shift processing on a local oscillator signal, and up-mixing the processed baseband signal and the processed local oscillator signal to generate a second RF signal; and combining the first RF signal and the second RF signal to output an RF signal; wherein the CIM3 and CIM5 components in the first RF signal and the second RF signal are opposite in phase and equal in amplitude during the combining process, thereby canceling each other out.

[0009] Furthermore, in some embodiments, the first radio frequency signal generated by the first processing path described above has a frequency of ω. c The CIM3 component and frequency of -3ω0 are ω c The phase of the CIM5 component with +5ω0 is the first phase; in the second radio frequency signal generated by the second processing path mentioned above, the frequency is ω c The CIM3 component and frequency of -3ω0 are ω c The phase of the CIM5 component of +5ω0 is the second phase, and the first phase and the second phase are π apart.

[0010] Furthermore, in some embodiments, the first radio frequency signal generated by the first processing path described above has a frequency of ω. c The phase of the useful signal component with +ω0 is the third phase; in the second radio frequency signal generated by the second processing path above, the frequency is ω c The phase of the useful signal component +ω0 is the fourth phase, and the difference between the third phase and the fourth phase is 0.

[0011] Furthermore, in some embodiments, the baseband signal of the first processing path described above includes an in-phase component I. a (t) and orthogonal component Q a (t), and the above baseband signal is: I a (t)=Acos(ω0t+φ), Q a (t) = Asin(ω0t + φ); where A is the signal amplitude, ω0 is the baseband angular frequency, and φ is the initial phase. It should be understood that the corresponding in-phase component I... a (t) and orthogonal component Q a(t) represents the component corresponding to the baseband signal of the first processing path (PATH A).

[0012] Furthermore, in some embodiments, in the second processing path described above, the baseband signal after the initial 45-degree phase shift processing includes an in-phase component I. a (t) and orthogonal component Q a (t), and the above baseband signal is: I b (t)=Acos(ω0t+φ+π / 4), Q b (t) = Asin(ω0t + φ + π / 4); A is the signal amplitude, ω0 is the baseband angular frequency, and φ is the initial phase. It should also be understood that the corresponding in-phase component I... b (t) and orthogonal component Q b (t) represents the component corresponding to the baseband signal of the second processing path (PATH B).

[0013] Furthermore, in some embodiments, in the first processing path described above, the local oscillator signal, after undergoing zero-degree initial phase shift processing, becomes cos(ω). c t+θ) and sin(ω c t+θ); In the second processing path described above, the local oscillator signal, after undergoing an initial phase shift of -45 degrees, becomes cos(ω). c t+θ-π / 4) and sin(ω c t+θ-π / 4), where ω c Let θ be the carrier angular frequency and θ be the initial phase.

[0014] Furthermore, in some embodiments, the nonlinear circuit model of the above-described baseband signal processing circuit is expressed as follows: y=a1x+a2x 2 +a3x 3 +a4x 4 +a5x 5 +...+a n x n ;Right now Where x is the input signal, y is the output signal, a1, a2, … are coefficients, and n is a positive integer.

[0015] Furthermore, in some embodiments, the aforementioned first radio frequency signal S A (t) and the aforementioned second radio frequency signal S B (t) are respectively represented as: S A (t)=I ad (t)cos(ω c t+θ)-Q ad (t)sin(ω c t+θ); S B (t)=I bd (t)cos(ω c t+θ-π / 4)−Q bd (t)sin(ω c t+θ-π / 4); in, , , , It should be understood that regarding S in the formula... A (t)=I ad (t)cos(ω c t+θ)-Q ad (t)sin(ω c t+θ); I ad (t) and Q ad (t) is expressed as the summation of the baseband signal of the first path (path A) after passing through a Taylor series; similarly, S B (t)=I bd (t)cos(ω c t+θ-π / 4)-Q bd (t)sin(ω c t+θ-π / 4); I bd (t) and Q bd (t) is expressed as the summation of the baseband signal of the second path (PATH B path) after passing through a Taylor series.

[0016] Furthermore, in some embodiments, the CIM3 and CIM5 components in the first processing path and the second processing path satisfy the following relationship: at frequency ω c At -3ω0, the amplitudes of the signals along both paths are... The phase difference is π; and at frequency ω c At +5ω0, the amplitudes of the signals on both paths are... The phase difference is π.

[0017] In a second aspect of this disclosure, a transmitting circuit is also provided that employs the CIM3 and CIM5 elimination methods as described above.

[0018] This disclosure has the following advantages over the prior art:

[0019] In some embodiments, CIM3 and CIM5 are simultaneously eliminated: by mixing and superimposing two baseband signals with a specific phase shift and the local oscillator signal, the simultaneous cancellation of third-order and fifth-order intermodulation distortions is achieved, significantly improving the linearity of the transmit link. In some embodiments, the circuit structure is greatly simplified: only two mixer circuits are required, which greatly reduces circuit complexity, chip area, and power consumption compared to the three mixer circuits required in the prior art. In some embodiments, the useful signals are superimposed without loss: the two useful signals are superimposed in phase at the output, while the distortion components are canceled out of phase, avoiding the signal attenuation problem that may be introduced in traditional methods. Attached Figure Description

[0020] The above and other features, advantages and aspects of the embodiments of this disclosure will become more apparent from the accompanying drawings and the following detailed description, wherein: Figure 1 A schematic diagram of the circuit architecture of the CIM3 and CIM5 elimination methods according to embodiments of the present disclosure is shown; Figure 2 A schematic diagram showing the cancellation of CIM3 and CIM5 under a fixed baseband phase shift α and different local oscillator phase shifts β according to an embodiment of the present disclosure is shown. Figure 3 A schematic diagram showing the cancellation of CIM3 and CIM5 under fixed carrier phase shift β and different baseband phase shift α according to an embodiment of the present disclosure is shown; Figure 4 A schematic flowchart of a method for eliminating CIM3 and CIM5 in a transmitting circuit according to an embodiment of the present disclosure is shown; And in each of the accompanying figures, the same or corresponding reference numerals indicate the same or corresponding parts. Detailed Implementation

[0021] Embodiments of the present disclosure will now be described in more detail with reference to the accompanying drawings. While some embodiments of the present disclosure are shown in the drawings, it should be understood that the present disclosure can be implemented in various forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided to provide a more thorough and complete understanding of the present disclosure. It should be understood that the accompanying drawings and embodiments of the present disclosure are for illustrative purposes only and are not intended to limit the scope of protection of the present disclosure. Similarly, for clarity and brevity, descriptions of well-known functions and structures are omitted in the following description.

[0022] In some of my embodiments, a method for eliminating CIM3 and CIM5 in the transmitting circuit is provided. It should be understood that this method applies different phase shifts to the baseband signal and the local oscillator signal through two independent processing paths, so that the useful signals in the two RF output signals are superimposed in phase, while the distortion components of CIM3 and CIM5 are canceled out in phase.

[0023] Figure 1 A schematic diagram of the circuit architecture for the CIM3 and CIM5 elimination methods of the transmitting circuit according to an embodiment of the present disclosure is shown. In this illustrated example, the transmitting circuit includes: a first processing path (PATH A): comprising a baseband phase shift unit A, a local oscillator phase shift unit A, and a mixer A; and a second processing path (PATH B): comprising a baseband phase shift unit B, a local oscillator phase shift unit B, and a mixer B; and a power combiner for combining the two radio frequency signals into an output.

[0024] It should be understood that, in some embodiments, the input orthogonal baseband signal is assumed to be: I a (t)=Acos(ω0t+φ), Q a (t)=Asin(ω0t+φ).

[0025] Furthermore, the processing of PATH A includes the following steps:

[0026] Step 1110: Baseband phase shift unit A performs a 0° phase shift on the I and Q signals, i.e., keeps them unchanged.

[0027] Step 1120: Local oscillator phase shift unit A performs a 0° phase shift on the local oscillator signal, that is, mixer A uses cos(ω) c t+θ) and sin(ω c t+θ).

[0028] Step 1130, considering the nonlinearity of the baseband processing circuit, its input-output relationship can be expressed by a Taylor series: y = a1x + a2x 2 +a3x 3 +a4x 4 +a5x 5 +... After upmixing, the output RF signal is: S A (t)=I ad (t)cos(ω c t+θ)-Q ad (t)sin(ω c t+θ); in, , .

[0029] Step 1140: Through expansion calculation, the amplitude and phase of each frequency component in the PATH A output can be obtained, among which the key components relevant to this invention are: Useful signal, frequency is (ω) c +ω0): Amplitude is The phase is φ+θ; CIM3, frequency is (ω c -3ω0): Amplitude is The phase is -(θ-3φ); CIM5, frequency is (ω c +5ω0): Amplitude is The phase is θ+5φ.

[0030] Furthermore, the processing of PATH B includes the following steps:

[0031] Step 1210: Baseband phase shift unit B performs a +45° phase shift on the I and Q signals to obtain: I b (t)=Acos(ω0t+φ+π / 4), Q b (t) = Asin(ω0t + φ + π / 4). Step 1220, the local oscillator phase shift unit B performs a -45° phase shift on the local oscillator signal, that is, mixer B uses cos(ω c t+θ-π / 4) and sin(ω c t+θ-π / 4).

[0032] Step 1230, the output radio frequency signal is: S B (t)=I bd (t)cos(ω c t+θ-π / 4)-Q bd (t)sin(ω c t+θ-π / 4).

[0033] Step 1240, through a similar expansion calculation, yields the key components in the output of PATH B as follows: Useful signal, frequency is (ω) c +ω0): Amplitude is The phase is φ+θ; CIM3, frequency is (ω c -3ω0): Amplitude is The phase is -(θ-3φ)+π; CIM5, frequency is (ω c +5ω0): Amplitude is The phase is θ+5φ+π.

[0034] Furthermore, the signal synthesis process includes the following steps: Step 1310, SA(t) and SB(t) are fed into the power combiner for addition.

[0035] For the useful signal: the two channels have equal amplitudes and the same phase, therefore they are in phase and superimposed, doubling the amplitude. For the CIM3 component: the two channels have equal amplitudes and a phase difference of π, therefore they completely cancel each other out of phase. For the CIM5 component: the two channels have equal amplitudes and a phase difference of π, therefore they completely cancel each other out of phase.

[0036] Therefore, by using the two specific phase shift configurations described above, the above embodiment successfully achieved the complete elimination of CIM3 and CIM5, while ensuring the lossless superposition of useful signals.

[0037] Furthermore, regarding the complete elimination of CIM3 and CIM5 while ensuring lossless superposition of useful signals, please refer to [reference needed]. Figure 1 The following explanation is provided.

[0038] In one embodiment, the initial phase shift of the input baseband signal is 0, that is, the baseband signal is I. a (t)=Acos(ω0t) and Q a (t)=Asin(ω0t), the initial phase shift of the local oscillator signal is 0, that is, the upmixer uses cos(ωt). c t) and sin(ω c t); The baseband nonlinear circuit model is y = a1x + a2x 2 +a3x 3 +a4x 4 +a5x 5 This applies to the I and Q paths respectively. It should be understood that the power of sixth-order and higher-order harmonics is relatively small and has no impact on key calculation results. To simplify the manual, the nonlinear circuit model formula here uses harmonic terms with n less than or equal to 5. The output RF signal is: S A (t)=I ad (t)cos(ω c t)-Q ad (t)sin(ω c t); Therefore, from this equation, we can further obtain the expression for the output signal under fifth-order nonlinearity as follows: .

[0039] It is important to understand that in the above formula, the amplitude and phase of each frequency component are as follows: Frequency ω c +ω0: Corresponding amplitude is The phase is 0; Frequency ω c The corresponding amplitude is Phase is ; Frequency ω c +2ω0: The corresponding amplitude is Phase is ; Frequency ω c -2ω0: Corresponding amplitude is Phase is ; Frequency ω c -3ω0: Corresponding amplitude is The phase is 0; Frequency ω c +4ω0: The corresponding amplitude is Phase is ; Frequency ω c -4ω0: Corresponding amplitude is Phase is ; Frequency ω c +5ω0: The corresponding amplitude is The phase is 0; It should be understood that the above expression fully describes the effect of fifth-order nonlinearity on orthogonal modulation system under zero initial phase conditions, including fundamental gain compression, carrier feedthrough, and generation of harmonic sidebands.

[0040] Furthermore, in one embodiment, for the sake of simplicity, the initial phases of the baseband signal and the carrier signal are set to 0, i.e., θ and φ are set to 0; the phase shift of the input baseband signal is α, i.e., the baseband signal is I. b (t)=Acos(ω0t+α) and Q b (t) = Asin(ω0t + α), the phase shift of the local oscillator signal is β, that is, the upmixer uses cos(ωt + α). c t+β) and sin(ω c The nonlinear circuit model of the baseband circuit is y = a1x + a2x. 2 +a3x 3 +a4x 4 +a5x 5 These are applied to the I and Q paths respectively. It should be understood that, again, because the power of sixth-order and higher-order harmonics is relatively small and has no impact on the calculation results of key conclusions, for the sake of simplicity in the manual, the nonlinear circuit model formula still uses harmonic terms with n less than or equal to 5. The output RF signal of the nonlinear circuit model is: S B (t)=I bd (t)cos(ω c t+β)−Q bd (t)sin(ω c t+β).

[0041] Therefore, from this equation, we can further obtain the expression for the output signal under fifth-order nonlinearity: .

[0042] It is important to understand that in the above formula, the amplitude and phase of each frequency component are as follows: Frequency ω c +ω0: Amplitude is The phase is β+α; Frequency ω c The corresponding amplitude is Phase β+ ; Frequency ω c +2ω0: The corresponding amplitude is The phase is β+2α ; Frequency is ω c -2ω0: then the amplitude is Phase β-2α ; Frequency is ω c -3ω0: then the amplitude is The phase is β-3α; Frequency is ω c +4ω0: then the amplitude is The phase is β+4α+ ; Frequency is ω c -4ω0: then the amplitude is The phase is β-4α+ ; Frequency is ω c +5ω0: The amplitude is The phase is β+5α; It should be understood that the above expression fully describes the effect of fifth-order nonlinearity on orthogonal modulation system under zero initial phase conditions, including fundamental gain compression, carrier feedthrough, and generation of harmonic sidebands.

[0043] Furthermore, for S A With S B The synthesized output is: signal frequency ω c +ω0 ​​component: ; CIM3 frequency ω c -3ω0 component: ; CIM5 frequency ω c +5ω0 component: ; Therefore, it should be understood that the final synthesized signal cos((ω) c +ω0)t) has the largest amplitude, cos((ω c -3ω0)t) has the smallest amplitude, cos((ω c +5ω0)t) has the smallest amplitude.

[0044] Therefore, the following steps are required:

[0045] 1) First, consider the superposition of two cosine signals with the same frequency: cosA + cos(A + ϕ) = 2 ⋅ ; Its envelope amplitude is ; It should be understood that the maximum amplitude is 2: when =±1, that is, ϕ=2mπ∈Z, where m is an integer; It should also be understood that the minimum amplitude is 0: when =0, that is, ϕ=π+2nπ∈Z, where n is an integer.

[0046] 2) Signal frequency ω c For the +ω0 component to have the maximum amplitude, the following condition must be met: β+α=2mπ (m∈Z); CIM3 frequency ω c The -3ω0 component amplitude is minimized, i.e., it is 0, and the following condition must be met: β-3α=π+2nπ (n∈Z); It should be understood that subtracting the two equations above yields: (β-3α)-(β+α)=(π+2nπ)-(2mπ); Therefore, -4α = π + 2(nm)π; Therefore, there is ; Then, if k=nm∈Z, we can obtain: α= = ; Since k can be positive or negative, the above expression is equivalent to: α= , p∈Z; that is, α is an odd multiple of π / 4; Furthermore, substituting the signal frequency ω c The +ω0 component has the largest amplitude and needs to satisfy the condition (β+α=2mπ (m∈Z)), therefore we have: β=2mπ-α; Since m can be any integer, β can be expressed as β = -α + 2mπ, where m ∈ Z; Therefore, setting p=m, the final result is: α= , β=-α+2pπ; (k, p ∈Z)); where k and p are arbitrary integers.

[0047] Furthermore, since the phase difference between the two paths of CIM5 is β+5α, from the calculated α and β values ​​above, we get β+5α=4α+2pπ=(2k+1)π+2pπ=π+2(k+p)π; (k, p ∈ Z); thus, we can conclude that the two paths of CIM5 are completely canceled out by their opposite phases; therefore, we can verify the result based on the final formula above: 1) Take k=0, p=0: then α=π / 4, β=-π / 4; 2) Take k=1, p=0: then α=3π / 4, β=-3π / 4; 3) Take k=-1, p=0: then α=-π / 4, β=π / 4; Furthermore, for k=0, p=0: that is, when α=π / 4, β=-π / 4, as... Figure 2 The figure illustrates the signal power synthesis and CIM3 and CIM5 cancellation when the baseband phase shift α is fixed at π / 4 and the local oscillator phase shift β varies from -π to π. It can be seen from the figure that when β is -π / 4, CIM3 and CIM5 completely cancel each other out, resulting in the maximum signal power synthesis. Furthermore, for k=0, p=0: that is, when α=π / 4, β=-π / 4, such as Figure 3 The figure illustrates the signal power synthesis and CIM3 and CIM5 cancellation when the local oscillator phase shift β is fixed at -π / 4 and the baseband phase shift α varies from -π to π. It can be seen from the figure that when α is π / 4, CIM3 and CIM5 completely cancel each other out, resulting in the maximum signal synthesis power.

[0048] In an alternative embodiment, the embodiments are essentially the same as those described above, except that the phase shifts of the baseband signal and the local oscillator signal can be equivalently transformed within a certain range. For example, the baseband phase shift of PATH B can be changed to -45°, while the local oscillator phase shift can be changed to +45°, as long as the relative phase difference between the two paths satisfies the conditions of CIM3 / CIM5 being out of phase and the useful signals being in phase. Any equivalent transformation that does not depart from the core idea of ​​this invention falls within the protection scope of this invention.

[0049] In some embodiments, such as Figure 4The diagram illustrates a method for eliminating CIM3 and CIM5 in a transmitting circuit, specifically comprising the following steps: Step 210, providing a first processing path and a second processing path; Step 220, in the first processing path, performing a zero-degree phase shift on the baseband signal and a zero-degree phase shift on the local oscillator signal, and up-mixing the processed baseband signal and the processed local oscillator signal to generate a first radio frequency (RF) signal; Step 230, in the second processing path, performing a 45-degree phase shift on the baseband signal and a negative 45-degree phase shift on the local oscillator signal, and up-mixing the processed baseband signal and the processed local oscillator signal to generate a second RF signal; Step 240, combining the first RF signal and the second RF signal to output an RF signal; wherein, the CIM3 and CIM5 components in the first RF signal and the second RF signal have opposite phases and equal amplitudes during the combining process, thus canceling each other out.

[0050] In another alternative embodiment, a transmitting circuit employing the above method is provided. This transmitting circuit includes: a baseband signal generator, a first mixer, a second mixer, a first local oscillator phase shifter, a second local oscillator phase shifter, a first baseband phase shifter, a second baseband phase shifter, and a power combiner. The phase shift of the first baseband phase shifter and the first local oscillator phase shifter are both 0°; the phase shift of the second baseband phase shifter is +45°, and the phase shift of the second local oscillator phase shifter is -45°. This circuit has a simple structure, requiring only two mixers to achieve simultaneous suppression of CIM3 and CIM5, making it particularly suitable for high-linearity, low-power wireless transmitters.

[0051] The various embodiments of this disclosure have been described above. These descriptions are exemplary and not exhaustive, nor are they limited to the disclosed embodiments. Many modifications and variations will be apparent to those skilled in the art without departing from the scope and spirit of the described embodiments. The terminology used herein is chosen to best explain the principles, practical application, or technical improvements to the embodiments in the market, or to enable others skilled in the art to understand the embodiments disclosed herein.

[0052] The above description is merely an optional embodiment of this disclosure and is not intended to limit this disclosure. Various modifications and variations can be made to this disclosure by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of this disclosure should be included within the scope of protection of this disclosure.

Claims

1. A method for eliminating CIM3 and CIM5 in a transmitting circuit, characterized in that, include: Provide a first processing path and a second processing path; In the first processing path, the baseband signal is subjected to zero-degree phase shift processing, and the local oscillator signal is subjected to zero-degree phase shift processing. The processed baseband signal and the processed local oscillator signal are then up-mixed to generate the first radio frequency signal. In the second processing path, the baseband signal is phase-shifted by 45 degrees and the local oscillator signal is phase-shifted by -45 degrees. The processed baseband signal and the processed local oscillator signal are then up-mixed to generate the second radio frequency signal. The first radio frequency signal and the second radio frequency signal are combined to output a radio frequency signal; In this process, the CIM3 and CIM5 components in the first radio frequency signal and the second radio frequency signal are opposite in phase and equal in amplitude when synthesized, thus canceling each other out.

2. The method according to claim 1, characterized in that, In the first radio frequency signal generated by the first processing path, the frequency is ω c The CIM3 component and frequency of -3ω0 are ω c The phase of the CIM5 component with +5ω0 is the first phase; in the second radio frequency signal generated by the second processing path, the frequency is ω c The CIM3 component and frequency of -3ω0 are ω c The phase of the CIM5 component with +5ω0 is the second phase, and the first phase differs from the second phase by π.

3. The method according to claim 1, characterized in that, In the first radio frequency signal generated by the first processing path, the frequency is ω c The phase of the useful signal component with +ω0 is the third phase; in the second radio frequency signal generated by the second processing path, the frequency is ω c The phase of the useful signal component +ω0 is the fourth phase, and the third phase and the fourth phase are 0 phase apart.

4. The method according to claim 1, characterized in that, In the first processing path, the baseband signal includes an in-phase component I. a (t) and orthogonal component Q a (t), and the baseband signal is: I a (t)=Acos(ω0t+φ),Q a (t)=Asin(ω0t+φ); Where A is the signal amplitude, ω0 is the baseband angular frequency, and φ is the initial phase.

5. The method according to claim 4, characterized in that, In the second processing path, the baseband signal after the 45-degree initial phase shift processing includes the in-phase component I. b (t) and orthogonal component Q b (t), and the baseband signal is: I b (t)=Acos(ω0t+φ+π / 4),Q b (t)=Asin(ω0t+φ+π / 4).

6. The method according to claim 1, characterized in that, In the first processing path, the local oscillator signal, after undergoing zero-degree initial phase shift processing, becomes cos(ω). c t+θ) and sin(ω c t+θ); In the second processing path, the local oscillator signal, after undergoing an initial phase shift of -45 degrees, becomes cos(ω). c t+θ-π / 4) and sin(ω c t+θ-π / 4), where ω c Let θ be the carrier angular frequency and θ be the initial phase.

7. The method according to claim 1, characterized in that, The nonlinear circuit model of the baseband signal processing circuit is expressed as follows: ； Where x is the input signal, y is the output signal, and a i Let n be the coefficient, where n is a positive integer.

8. The method according to claim 5, characterized in that, The first radio frequency signal S A (t) and the second radio frequency signal S B (t) are respectively represented as: S A (t)=I ad (t)cos(ω c t+θ)-Q ad (t)sin(ω c t+θ); S B (t)=I bd (t)cos(ω c t+θ-π / 4)-Q bd (t)sin(ω c t+θ-π / 4); in, , , , .

9. The method according to claim 1, characterized in that, The CIM3 and CIM5 components in the first processing path and the second processing path satisfy the following relationship: At frequency ω c At -3ω0, the amplitudes of the signals along both paths are... The phase difference is π; as well as At frequency ω c At +5ω0, the amplitudes of the signals on both paths are... The phase difference is π.

10. A transmitting circuit, characterized in that, The CIM3 and CIM5 elimination methods as described in any one of claims 1 to 9 are employed.

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