A power filter insertion loss optimization design method

Abstract

This invention relates to the field of power filter technology and discloses a method for optimizing the insertion loss design of a power filter. The method includes: firstly, collecting the temperature difference at each monitoring point and correcting the characteristic length of the resonant cavity using the thermal expansion coefficient; secondly, introducing a temperature sensitivity coefficient to construct a dynamic dielectric constant distribution model and correcting the electromagnetic wave phase velocity accordingly; thirdly, calculating the characteristic impedance deviation and interface scattering loss density caused by the dielectric gradient; fourthly, calculating the dynamic skin depth under high-frequency pulses based on the current density benchmark determined by the input power density and establishing a transient conductivity correction factor; and finally, integrating the preceding physical parameters to output a target impedance compensation control sequence. This invention effectively compensates for the insertion loss caused by thermal distortion, dielectric degradation, and the skin effect through closed-loop coupling calculation of multi-dimensional physical parameters, improving the impedance matching accuracy and power transmission efficiency of the filter under extreme operating conditions.

Description

Technical Field

[0001] This invention relates to the field of power filter technology, specifically to a method for optimizing the insertion loss design of power filters. Background Technology

[0002] In high-frequency, high-power power transmission systems, waveguide power filters are core frequency selection devices that ensure system electromagnetic compatibility and transmission stability. Conventional filter design processes primarily rely on ideal physical models. These models typically assume that the geometric dimensions and dielectric parameters of the resonant cavity remain constant during operation. However, in practical applications such as satellite communication, radar transmission, and high-power microwave processing, filters are often subjected to extreme dynamic loads. Existing technologies often neglect the localized temperature rise effect induced by high-power loads. When a strong current passes through the waveguide wall, the continuously accumulated heat energy induces anisotropic geometric thermal distortion. This deformation leads to center frequency shift and worsens insertion loss. Simultaneously, the dielectric constant of the dielectric material filling the filter is extremely sensitive to the thermal field. The generation of local hot spots results in a non-uniform dielectric gradient distribution within the dielectric. Existing static algorithms cannot map the real-time interference of this dielectric gradient on the electromagnetic wave phase velocity. This leads to unpredictable impedance discontinuities within the waveguide, resulting in severe interface scattering losses. Furthermore, when dealing with high-frequency pulsed currents, traditional skin effect models only provide a steady-state reference for the conductive thickness. In transient high-power pulse environments, the charge distribution pattern on the conductor surface deviates significantly with drastic fluctuations in current density. This transient characteristic causes a dynamic contraction of the equivalent conductive area of ​​the metal inner wall, directly leading to a loss in conductivity. Existing compensation schemes cannot establish a logical connection between macroscopic thermal expansion and microscopic skin depth attenuation. This insertion loss failure caused by multidimensional physical field coupling not only reduces power transmission efficiency but also seriously threatens the thermal safety of the system. Due to the lack of a deep analytical mechanism for geometric distortion, dielectric degradation, and transient conduction characteristics, existing technologies struggle to achieve accurate impedance matching under extreme conditions.

[0003] In summary, there is a need to provide a filter insertion loss optimization scheme that can integrate thermal distortion, dielectric gradient, and transient conductivity correction. Summary of the Invention

[0004] This invention provides a method for optimizing the insertion loss of a power filter, which helps to solve the problems mentioned in the background art.

[0005] This invention provides the following technical solution: a method for optimizing the insertion loss of a power supply filter, comprising:

[0006] The real-time temperature difference is calculated based on the real-time absolute temperature and the reference ambient temperature. The compensated cavity characteristic length is then calculated by combining the initial cavity characteristic length and the cavity thermal expansion coefficient.

[0007] The dynamic dielectric constant is calculated based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference.

[0008] The corrected phase velocity is calculated based on the absolute physical constant of magnetic permeability and the dynamic dielectric constant.

[0009] The dynamic characteristic impedance is calculated based on the permeability, the corrected phase velocity, and the cavity length change ratio. The gradient boundary reflection coefficient is then calculated in combination with the initial characteristic impedance.

[0010] The interface scattering loss density is calculated based on the initial input microwave power density and the reflection coefficient, and the reference current density threshold is calculated in combination with the cavity cross-sectional area.

[0011] The dynamic skin depth is calculated based on the power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold.

[0012] The transient equivalent conductivity is calculated based on the initial conductivity, dynamic skin depth and initial cavity characteristic length, and then the conductivity correction factor is calculated.

[0013] The compensated matching impedance is calculated based on the dynamic characteristic impedance, correction factor, and loss density, and an impedance compensation control sequence is generated based on the difference between the compensated and initial characteristic impedances.

[0014] Optionally, the step of calculating the real-time temperature difference based on the real-time absolute temperature and the reference ambient temperature, and calculating the compensated cavity characteristic length by combining the initial cavity characteristic length and the cavity thermal expansion coefficient, includes:

[0015] A temperature sensor is deployed inside the resonant cavity of a waveguide-type power filter.

[0016] Acquire real-time absolute temperature data and ambient baseline temperature data for each monitoring point;

[0017] The real-time temperature difference is obtained by performing a differential calculation between the monitored temperature and the reference temperature.

[0018] The length change is determined by the product of the initial cavity length, thermal expansion parameters, and the real-time temperature difference;

[0019] The initial cavity length is summed with the change to obtain the compensated characteristic length.

[0020] Optionally, the calculation of the dynamic dielectric constant based on the initial dielectric constant, the dielectric constant temperature sensitivity coefficient, and the real-time temperature difference includes:

[0021] Obtain the initial dielectric parameters and temperature sensitivity constant of the dielectric material;

[0022] The product of the sensitivity constant and the real-time temperature difference is used to calculate the medium property drift rate.

[0023] Subtract the product term of the initial dielectric parameter and the drift rate;

[0024] Determine the dynamic medium parameters under the current thermal environment.

[0025] Optionally, the calculation of the corrected phase velocity based on the absolute physical constant permeability and the dynamic permittivity includes:

[0026] Retrieve the system's stored permeability constant;

[0027] Multiply the constant with the dynamic medium parameter;

[0028] Perform the square root reciprocal calculation on the product result;

[0029] The corrected propagation velocity within the waveguide cavity is obtained.

[0030] Optionally, the step of calculating the dynamic characteristic impedance based on permeability, corrected phase velocity, and cavity length change ratio, and calculating the gradient boundary reflection coefficient in combination with the initial characteristic impedance, includes:

[0031] Multiply the permeability by the corrected propagation speed;

[0032] The dynamic characteristic impedance is obtained by weighting the multiplication result using the ratio of the compensated characteristic length to the initial characteristic length.

[0033] Calculate the deviation between the dynamic characteristic impedance and the initial impedance;

[0034] Divide the deviation value by the sum of the impedances of the two to obtain the interface reflection coefficient.

[0035] Optionally, the step of calculating the interface scattering loss density based on the initial input microwave power density and the reflection coefficient, and calculating the reference current density threshold based on the cavity cross-sectional area, includes:

[0036] Read the input microwave energy density;

[0037] Calculate the product of the energy density and the square of the reflection coefficient to determine the scattering loss component;

[0038] The square root is calculated based on the ratio of the input energy density to the product of the initial impedance and the cavity area.

[0039] Generate a current density reference per unit area.

[0040] Optionally, the calculation of dynamic skin depth based on power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold includes:

[0041] The standard conductive thickness is determined by multiplying the pulse angular frequency, permeability, and initial conductivity parameters.

[0042] Monitor real-time high-frequency pulse current and calculate the real-time current intensity per unit area;

[0043] Calculate the offset ratio between the real-time current intensity and the current density reference.

[0044] By performing a nonlinear reduction calculation on the standard conductive thickness in conjunction with the interface reflection coefficient, the dynamic skin depth is obtained.

[0045] Optionally, the step of calculating the transient equivalent conductivity based on the initial conductivity, dynamic skin depth, and initial cavity characteristic length, and then calculating the conductivity correction factor, includes:

[0046] The equivalent conductivity parameter is obtained by multiplying the initial conductivity parameter by the geometric ratio of the dynamic skin depth to the initial cavity length.

[0047] Divide the equivalent conductivity parameter by the initial conductivity parameter; output a dimensionless correction factor characterizing the degree of conductivity degradation.

[0048] Optionally, the step of calculating the compensated matching impedance based on the dynamic characteristic impedance, correction factor, and loss density, and generating an impedance compensation control sequence based on the difference between the compensated and initial characteristic impedance, includes:

[0049] Calculate the product of the dynamic characteristic impedance and the dimensionless correction factor;

[0050] The energy transfer ratio is determined by the ratio of the scattering loss component to the input energy density;

[0051] The product is weighted twice by the energy transfer ratio to obtain the target matching impedance value;

[0052] The deviation vector of the target matching impedance value relative to the initial reference impedance is calculated and output as the adjustment sequence.

[0053] The present invention has the following beneficial effects:

[0054] 1. By acquiring real-time absolute temperature and transient high-frequency current inside a waveguide-type power filter, this technical solution derives the characteristic length of thermal expansion and contraction of the resonant cavity, dynamic dielectric constant, corrected phase velocity, interface scattering loss density, and dynamic skin depth, thereby obtaining a conductivity correction factor and ultimately generating an impedance compensation control sequence to offset errors. This solution is designed because power filters are often used in specific extreme operating environments for high-power microwave and high-frequency pulse transmission. Under these conditions, continuous high-load operation causes localized high temperatures inside the filter, resulting in physical thermal expansion of the metal cavity and degradation of the properties of the internal insulating dielectric material. These multi-dimensional physical changes directly alter the propagation speed of electromagnetic waves within the filter, leading to impedance mismatch and severe electromagnetic wave interface reflection and loss. Simultaneously, transient high-frequency alternating current induces a strong skin effect, causing a sharp reduction in the usable conductive cross-sectional area and resulting in a decrease in overall equivalent conductivity. Conventional steady-state impedance matching methods are completely inadequate to cope with this dynamic impedance shift caused by the superposition of extreme high temperatures and high-frequency pulses, leading to severe insertion loss and power waste. This technical solution offers unique and beneficial effects for this specific high-power, high-frequency thermally coupled environment. It employs deep multi-physics coupling quantification to deeply couple macroscopic thermodynamic geometric deformation, material dielectric parameter drift, and microscopic high-frequency skin depth loss, accurately calculating the impedance bias caused by temperature gradients and pulse current compression on the internal electromagnetic transmission path of the filter. This real-time adaptive correction mechanism for extreme conditions dynamically outputs matching commands to compensate for multiple physical distortions, effectively suppressing microwave scattering caused by sudden changes in internal impedance. This solution enables the power filter to maintain highly accurate impedance matching even under harsh conditions of severe temperature rise and transient strong pulse current impacts, significantly reducing dynamic insertion loss during operation and effectively ensuring the electromagnetic energy transmission efficiency and overall operational stability of the high-frequency, high-power system.

[0055] 2. By deploying high-frequency fiber optic temperature sensors and Hall current sensors at key nodes inside the resonant cavity of the waveguide-type power filter, the real-time absolute temperature of the monitoring point and the reference ambient temperature under the current environment are obtained. The real-time temperature difference is calculated by differential calculation between the obtained absolute temperature and the reference temperature. Then, the initial cavity length is linearly quantized using the inherent thermal expansion parameters of the waveguide substrate to obtain the length change under thermal expansion. Finally, the initial length and the change are added to obtain the accurately compensated cavity characteristic length. This processing method directly addresses the unavoidable thermodynamic deformation phenomenon of the cavity metal under high-power and high-current transmission conditions, accurately mapping the purely physical temperature rise phenomenon into a substantial bias in geometric dimensions, completely overcoming the limitations of traditional power filter design. The static idealization assumption that the internal physical form is an absolutely rigid body eliminates the defects of the control system. This allows the system to perceive and accurately quantify the microscopic elongation or contraction effects of the cavity induced by continuous high-load operation or harsh external thermal environment in real time. This provides an extremely rigorous and highly realistic dynamic spatial geometric benchmark for subsequent wave impedance matching and high-frequency microwave electromagnetic transmission characteristic analysis. It effectively and completely eliminates the step-by-step cumulative calculation deviation caused by distortion of the basic physical scale. This ensures that all subsequent calculation steps of complex electromagnetic networks that are highly dependent on the waveguide cavity size are based on the real, objective and dynamically updated physical boundary morphology. From the source of the underlying physical architecture, it curbs the trend of microwave frequency drift and transmission insertion loss deterioration caused by high-temperature thermal distortion.

[0056] 3. By extracting the initial dielectric parameters and inherent temperature-sensitive constant of the waveguide dielectric material, and performing a rigorous product operation between this sensitive constant and the real-time temperature difference obtained in the previous step, the attenuation drift rate of the insulating dielectric properties under the continuous action of a high-frequency strong thermal field is accurately extracted. Furthermore, the dielectric performance loss due to this drift rate is subtracted from the initial dielectric parameters, thereby accurately establishing the dynamic dielectric parameters under the current extreme thermal environment. This computational mechanism deeply analyzes the physical mechanism of electrical performance degradation of the insulating filling medium inside the waveguide filter when it approaches its critical physical point at high temperatures. It breaks the rigid and idealistic setting of the dielectric constant of the insulating material as constant in conventional high-frequency microwave electromagnetic analysis, fully considering that local hot spots not only cause expansion of the surrounding metal structure but also lead to core... The nonlinear weakening of the internal charge distribution energy storage and dielectric polarization capability of microwave transmission media is addressed by a dynamic multiphysics mapping method that directly transforms macroscopic temperature field parameters into microscopic dielectric polarization characteristic attenuation. This method can track in real time the spatial distribution distortion of internal permittivity caused by long-term thermal aging or transient drastic temperature rise in the insulating substrate. It can lock in advance and accurately quantify the potential broadband electromagnetic field distribution disorder caused by the deterioration of the internal microscopic properties of the material. This allows the filter adaptive control system to grasp the real electrical operating state of the internal core dielectric layer in real time, providing indispensable material-level underlying physical parameters for subsequent phase velocity shift assessment and accurate correction of wave impedance mismatch. This greatly improves the environmental dynamic adaptive capability and physical state simulation fidelity of the high-frequency impedance matching model under extremely complex thermodynamic boundary conditions.

[0057] 4. By calling the preset absolute physical constant permeability in the system's underlying control chip, and performing a rigorous product fusion calculation with the dynamic medium parameters accurately obtained based on the real-time thermal field distribution, and then solving the square root reciprocal equation of the physical product result, the true corrected propagation phase velocity of the microwave signal in the waveguide resonant cavity under the current severe thermal interference environment is derived. This calculation process closely follows the basic principles of classical microwave electromagnetic wave dynamics, directly linking and projecting the microscopic degradation effect of the insulating medium properties caused by extreme high-temperature thermodynamic conditions onto the core physical kinematic characteristics of macroscopic electromagnetic waves. It accurately captures and quantifies the nonlinear shift in microwave high-frequency transmission phase velocity that is inevitably caused by the decrease in dielectric constant, effectively avoiding the problem of adhering to vacuum or room-temperature standard microwaves under actual complex high-power high-frequency full-load conditions. The serious misjudgment of the center frequency of broadband high-frequency resonance caused by wave velocity enables the impedance compensation and control system to accurately and in real time grasp the true dynamic propagation speed of high-frequency electromagnetic waves as they travel through non-uniform media subjected to physical distortion. This real-time calibration and correction mechanism for the underlying wave velocity parameters not only reflects the direct physical influence of the accelerated aging of insulating dielectric materials on the hysteresis effect of high-frequency microwave beam transmission in a realistic and detailed manner, but also provides a decisive transmission speed fundamental variable for the subsequent construction of dynamic multidimensional electromagnetic boundary and impedance abrupt change scattering analysis models. It completely eliminates the underlying static error of phase velocity when performing complex microwave network node analysis under dynamic alternating strong thermal field conditions, and significantly enhances the energy transmission phase stability and overall frequency response accuracy of high-power power supply filters when processing high-frequency broadband pulse signals under complex and severe temperature variation conditions.

[0058] 5. By multiplying the absolute physical constant permeability by the precisely corrected microwave propagation speed, and innovatively introducing the geometric ratio of the compensated dynamic characteristic length (including thermodynamic deformation factors) to the initial system length, a spatial proportional weighting correction is applied to the velocity product. This yields a dynamic characteristic impedance that takes into account both microscopic dielectric material degradation and macroscopic metal physical expansion distortion factors. Subsequently, the deviation between this disturbed dynamic impedance and the filter's factory standard initial impedance is compared and analyzed. The algebraic difference between the two is divided by the sum of their physical impedances to obtain the high-frequency gradient boundary reflection coefficient. This logical design deeply and thoroughly analyzes the spatial abrupt change mechanism of microwave transmission path impedance induced by non-uniform and drastic temperature rise, restoring the originally textual and homogenized internal physical waveguide cavity of the filter to a true reflection of the objective existence of numerous non-uniform dielectric gradients and microwave wave velocity folding phenomena. By accurately quantifying the relative intensity of high-frequency wave reflections that inevitably occur when high-energy electromagnetic waves cross these illusory medium boundaries formed by the aggregation of local non-uniform hot spots at complex physical boundaries, this method completely overcomes the theoretical blind spots of the traditional one-size-fits-all impedance blind matching method. This allows the closed-loop control system to accurately predict and control how much microwave load energy will be mercilessly bounced back due to the slight mismatch of the internal transmission impedance chain. This quantitative assessment and model deconstruction of the internal electromagnetic microwave spatial scattering intensity establishes a solid and irrefutable theoretical microwave network basis for subsequent system-level high-frequency energy dissipation calculations and prediction of deterioration trends. It fundamentally reveals the internal hidden wave dynamics that cause the sharp and abnormal increase in the total insertion loss of the system under extreme high temperature and high pressure environments, and points out a deterministic feedback control direction for achieving adaptive and precise high-frequency impedance correction command output.

[0059] 6. By real-time reading of the initial external microwave total power density input to the acquisition system under the current operating conditions, and directly multiplying it with the square term of the high-frequency interface reflection coefficient derived from the previous core logic, the interface scattering loss density dissipated per unit physical cross-sectional area due to discontinuous changes in internal microwave impedance is quantified. Simultaneously, by dividing the acquired input microwave power density by the physical ratio of the product of the initial standard characteristic impedance and the waveguide cavity cross-sectional area, and performing a square root algebraic calculation, a dynamic reference current density physical threshold for the system at the current high-intensity microwave energy input level is rigorously established. This calculation process effectively transforms the highly abstract electromagnetic wave interface reflection coefficient into a visible and controllable macroscopic high-frequency heat dissipation power index for engineering applications, intuitively revealing the attenuation of high-frequency ineffective microwave energy caused by physical distortions in extreme environments. The accurate scale allows the subsequent impedance compensation network to clearly determine the precise size of the microwave energy gap that the current system urgently needs to compensate for. The establishment of the dynamic reference current density threshold based on power calculation defines a strict physical safety red line for the surge of extreme high-frequency pulse current on the inner surface of the waveguide. It cleverly binds high-frequency power parameters, characteristic impedance, and physical spatial cross-sectional area depth together, constructing a current safety criterion reference system that can dynamically follow the fluctuation of input power. This not only provides a unified evaluation scale for accurately measuring the microscopic destructive force of transient surge current, but also significantly improves the dynamic adaptation range of the entire control system when facing irregular fluctuations in pulse power. It effectively avoids the problem of low-frequency compensation sensitivity mismatch or high-frequency overcompensation oscillation caused by using a single fixed and rigid alarm threshold.

[0060] 7. By comprehensively utilizing externally input pulse alternating angular frequency, constant permeability, and initial conductivity parameters, the normal standard conductive thickness under classical high-frequency electromagnetic theory is calculated and determined. The absolute intensity of transient high-frequency pulse current is captured in real time, and its instantaneous density on the waveguide cross-sectional area is rigorously calculated. Then, the instantaneous current density is compared with a previously established dynamic current threshold using bias ratio analysis. Finally, a nonlinear reduction operation is performed on the standard conductive thickness using the interface microwave reflection coefficient to obtain the dynamic skin depth under harsh operating conditions. This core strategy creatively extends the steady-state skin effect to complex extreme transient high-frequency pulse strong coupling conditions, keenly capturing the strong squeezing effect of the ultra-high-frequency alternating magnetic field on the charge distribution on the metal surface. This squeezing directly leads to… The actual metal cross-section that effectively participates in conduction undergoes a precipitous shrinkage under transient high-power pulse impact. This step not only quantitatively analyzes the geometric attenuation degree of this skin depth, but also deeply correlates it with the severe internal alternating wave reflection interference phenomenon. It truly reflects the secondary deterioration effect of the complex standing wave field formed by the superposition of reflected microwaves and traveling waves on the geometry of the microscopic conductive channel. It effectively fills the technical gap that has long been ignored in the microscopic effects of transient current distortion in conventional filter design. This enables the control system to accurately perceive the real-time reduction of the electron transmission path at the microscopic level, providing a precise calculation basis at the microscopic physical level for the accurate adjustment of the overall macroscopic conductivity performance. It greatly enhances the physical limit of the communication system's ability to withstand destructive surge pulse impacts.

[0061] 8. By extracting the initial macroscopic set dimensions of the physical waveguide cavity, the dynamic skin depth obtained by the strong compression of the microscopic high-frequency pulse is compared with the macroscopic feature length using a cross-physical-level geometric proportional normalization. This geometric discount attenuation rate is then directly applied to the initial conductivity parameters of the material itself to calculate the severely weakened transient equivalent conductivity parameters under the current extreme conditions. Furthermore, this damaged equivalent conductivity parameter is divided by the initial reference parameter to precisely extract the pure proportional characteristic conductivity correction factor, completely stripped of specific physical dimensions. This conversion process cleverly bridges the characterization gap between the microscopic high-frequency charge skin distribution and the macroscopic material conductivity properties, enabling the direct reading of microscopic skin values ​​that are difficult for external hardware to access. The deep variation value is smoothly transformed into a scalar of material conductivity degradation coefficient with universal engineering control significance. This thorough physical dimensionality reduction and dimensionless processing method not only greatly simplifies the concurrent calculation burden of downstream impedance matching compensation network chips, so that the control system no longer needs to be deeply involved in the analysis of complex microscopic three-dimensional electromagnetic fields, but also the pure digital correction factor intuitively and accurately reflects the percentage of comprehensive conductivity loss caused by high-frequency transient high current impact. It provides an extremely effective standardized interface for the overall microwave control system to find a fast linear compensation fulcrum in the intricate nonlinear power attenuation model, and significantly improves the cross-domain computation efficiency and interface compatibility of the transition from the underlying physical model to the hardware digital control sequence.

[0062] 9. By comprehensively aggregating the dynamic characteristic impedances and dimensionless conductivity correction factors derived progressively from various physical dimensions in the early stages, and performing characteristic product fusion, while simultaneously introducing the actual energy transfer residual ratio established based on the ratio of scattering loss components to the original input energy density, the aforementioned product results are subjected to deep secondary global weighting processing to obtain the target matching impedance value urgently needed to be achieved under ideal conditions. Finally, the target matching impedance value is subtracted from the initial reference impedance to obtain the precise absolute deviation vector between the two, which is then output as the unique control sequence. This comprehensive step completes the final convergence from multi-physics discrete quantitative analysis to a single deterministic control command at the end, and addresses the physical displacement caused by thermodynamic geometric deformation and the material aging due to heat. The degradation of dielectric constant caused by dielectric constant changes, the impedance wave reflection dissipation induced by dielectric gradient, and the loss of skin conductivity caused by transient pulse surge current compression are all packaged into a rigorous ultimate compensation equation system. This system completely connects the entire control chain from abnormal parameter acquisition to physical mechanism analysis to execution sequence generation, ensuring that the output impedance compensation control sequence contains the counterbalancing components of all extreme deterioration factors. This allows the digital synthesized impedance network to perform reverse impedance compensation actions with this unique and precise control command, forcibly flattening the impedance bias curve in complex extreme conditions. This perfectly achieves the ultimate design goal of minimizing the dynamic insertion loss of high-frequency power filters and approaching the ideal microwave energy transmission efficiency. Attached Figure Description

[0063] Figure 1 This is a schematic diagram of the basic process of the present invention.

[0064] Figure 2 This is a flowchart of the overall multiphysics field closed-loop compensation process of the present invention.

[0065] Figure 3 This is a flowchart illustrating the calculation of geometric thermal distortion and medium degradation deviation in this invention.

[0066] Figure 4 This is a flowchart illustrating the correction process for high-frequency transient surge current squeezing conductive area in this invention. Detailed Implementation

[0067] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0068] Example 1, refer to Figure 1A method for optimizing the insertion loss of a power supply filter, comprising:

[0069] The real-time temperature difference is calculated based on the real-time absolute temperature and the reference ambient temperature. The compensated cavity characteristic length is then calculated by combining the initial cavity characteristic length and the cavity thermal expansion coefficient.

[0070] The dynamic dielectric constant is calculated based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference.

[0071] The corrected phase velocity is calculated based on the absolute physical constant of magnetic permeability and the dynamic dielectric constant.

[0072] The dynamic characteristic impedance is calculated based on the permeability, the corrected phase velocity, and the cavity length change ratio. The gradient boundary reflection coefficient is then calculated in combination with the initial characteristic impedance.

[0073] The interface scattering loss density is calculated based on the initial input microwave power density and the reflection coefficient, and the reference current density threshold is calculated in combination with the cavity cross-sectional area.

[0074] The dynamic skin depth is calculated based on the power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold.

[0075] The transient equivalent conductivity is calculated based on the initial conductivity, dynamic skin depth and initial cavity characteristic length, and then the conductivity correction factor is calculated.

[0076] The compensated matching impedance is calculated based on the dynamic characteristic impedance, correction factor, and loss density, and an impedance compensation control sequence is generated based on the difference between the compensated and initial characteristic impedances.

[0077] The calculation of the real-time temperature difference based on the real-time absolute temperature and the reference ambient temperature, combined with the initial cavity characteristic length and the cavity thermal expansion coefficient, to calculate the compensated cavity characteristic length includes:

[0078] A temperature sensor is deployed inside the resonant cavity of a waveguide-type power filter.

[0079] Acquire real-time absolute temperature data and ambient baseline temperature data for each monitoring point;

[0080] The real-time temperature difference is obtained by performing a differential calculation between the monitored temperature and the reference temperature.

[0081] The length change is determined by the product of the initial cavity length, thermal expansion parameters, and the real-time temperature difference;

[0082] The initial cavity length is summed with the change to obtain the compensated characteristic length.

[0083] The calculation of the dynamic dielectric constant based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference includes:

[0084] Obtain the initial dielectric parameters and temperature sensitivity constant of the dielectric material;

[0085] The product of the sensitivity constant and the real-time temperature difference is used to calculate the medium property drift rate.

[0086] Subtract the product term of the initial dielectric parameter and the drift rate;

[0087] Determine the dynamic medium parameters under the current thermal environment.

[0088] The calculation of the corrected phase velocity based on the absolute physical constant of permeability and the dynamic dielectric constant includes:

[0089] Retrieve the system's stored permeability constant;

[0090] Multiply the constant with the dynamic medium parameter;

[0091] Perform the square root reciprocal calculation on the product result;

[0092] The corrected propagation velocity within the waveguide cavity is obtained.

[0093] The calculation of dynamic characteristic impedance based on permeability, corrected phase velocity, and cavity length change ratio, combined with the initial characteristic impedance to calculate gradient boundary reflection coefficient, includes:

[0094] Multiply the permeability by the corrected propagation speed;

[0095] The dynamic characteristic impedance is obtained by weighting the multiplication result using the ratio of the compensated characteristic length to the initial characteristic length.

[0096] Calculate the deviation between the dynamic characteristic impedance and the initial impedance;

[0097] Divide the deviation value by the sum of the impedances of the two to obtain the interface reflection coefficient.

[0098] The calculation of interface scattering loss density based on the initial input microwave power density and the reflection coefficient, combined with the calculation of the reference current density threshold based on the cavity cross-sectional area, includes:

[0099] Read the input microwave energy density;

[0100] Calculate the product of the energy density and the square of the reflection coefficient to determine the scattering loss component;

[0101] The square root is calculated based on the ratio of the input energy density to the product of the initial impedance and the cavity area.

[0102] Generate a current density reference per unit area.

[0103] The calculation of dynamic skin depth based on power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold includes:

[0104] The standard conductive thickness is determined by multiplying the pulse angular frequency, permeability, and initial conductivity parameters.

[0105] Monitor real-time high-frequency pulse current and calculate the real-time current intensity per unit area;

[0106] Calculate the offset ratio between the real-time current intensity and the current density reference.

[0107] By performing a nonlinear reduction calculation on the standard conductive thickness in conjunction with the interface reflection coefficient, the dynamic skin depth is obtained.

[0108] The calculation of transient equivalent conductivity based on initial conductivity, dynamic skin depth, and initial cavity characteristic length, followed by the calculation of conductivity correction factors, includes:

[0109] The equivalent conductivity parameter is obtained by multiplying the initial conductivity parameter by the geometric ratio of the dynamic skin depth to the initial cavity length.

[0110] Divide the equivalent conductivity parameter by the initial conductivity parameter; output a dimensionless correction factor characterizing the degree of conductivity degradation.

[0111] The process of calculating the compensated matching impedance based on the dynamic characteristic impedance, correction factor, and loss density, and generating an impedance compensation control sequence based on the difference between the compensated and initial characteristic impedance, includes:

[0112] Calculate the product of the dynamic characteristic impedance and the dimensionless correction factor;

[0113] The energy transfer ratio is determined by the ratio of the scattering loss component to the input energy density;

[0114] The product is weighted twice by the energy transfer ratio to obtain the target matching impedance value;

[0115] The deviation vector of the target matching impedance value relative to the initial reference impedance is calculated and output as the adjustment sequence.

[0116] Example 2, based on Example 1, refers to... Figures 2 to 4 This paper presents a more specific method for optimizing the insertion loss of power supply filters, which includes the derivation of underlying physical quantities. Specifically, it includes:

[0117] Reference Figure 3 The calculation of the real-time temperature difference based on the real-time absolute temperature and the reference ambient temperature, combined with the initial cavity characteristic length and the cavity thermal expansion coefficient, to calculate the compensated cavity characteristic length includes:

[0118] The purpose of this step is to quantify the effect of temperature rise on the physical dimensions of the waveguide cavity.

[0119] High-frequency fiber optic temperature sensors and Hall current sensors are deployed at key nodes inside the resonant cavity of the waveguide-type power filter, and a network analyzer is connected to the control system.

[0120] At the start of the current step, the system first drives the high-frequency fiber optic temperature sensor to acquire the first... Real-time absolute temperature at each monitoring point And obtain the reference ambient temperature under the current environment. ;

[0121] The system calculates the temperature offset at each monitoring point by the difference between the acquired real-time absolute temperature and the reference temperature. The calculation formula is as follows:

[0122] In the formula, For the first Real-time temperature difference at each monitoring point; For the first Real-time absolute temperature at each monitoring point; The reference ambient temperature;

[0123] Next, the system reads the initial cavity feature length from the register. And the coefficient of thermal expansion of the cavity of the material The value of the cavity thermal expansion coefficient is based on the standard physical property parameter manual of the selected waveguide substrate. When the value is too large, it will cause the system to over-complicate the geometric compensation calculation for slight temperature fluctuations. When the value is too small, it will not be able to accurately cover the actual thermal deformation deviation under extremely high load.

[0124] Based on the calculated temperature difference and the thermal expansion characteristics of the material itself, the system derives the actual dimensions of the cavity after thermal distortion. This process involves linear compensation for the initial length through the introduction of the coefficient of thermal expansion, and the calculation formula is as follows:

[0125] In the formula, The compensated cavity feature length; The initial cavity feature length; The coefficient of thermal expansion of the cavity; For the first Real-time temperature difference at each monitoring point.

[0126] By deploying high-frequency fiber optic temperature sensors and Hall current sensors at key nodes inside the resonant cavity of a waveguide-type power filter, the real-time absolute temperature of the monitoring point and the reference ambient temperature under the current environment are obtained. The real-time temperature difference is calculated by differential calculation between the obtained absolute temperature and the reference temperature. Then, the initial cavity length is linearly quantized using the inherent thermal expansion parameters of the waveguide substrate to obtain the length change under thermal expansion. Finally, the initial length and the change are added together to obtain the accurately compensated cavity characteristic length. This processing method directly addresses the unavoidable thermodynamic deformation phenomenon of the cavity metal under high-power and high-current transmission conditions, accurately mapping the purely physical temperature rise phenomenon into a substantial geometrical bias, completely overcoming the limitations of traditional power filter designs that rely on internal... The static idealization assumption that the physical form of the cavity is an absolutely rigid body eliminates the defects of the control system. This allows the control system to sense and accurately quantify the microscopic elongation or contraction effect of the cavity induced by continuous high-load operation or harsh external thermal environment in real time. This provides an extremely rigorous and highly realistic dynamic spatial geometric benchmark for subsequent wave impedance matching and high-frequency microwave electromagnetic transmission characteristic analysis. It effectively and completely eliminates the step-by-step cumulative calculation deviation caused by distortion of the basic physical scale. This ensures that all subsequent calculation steps of complex electromagnetic networks that are highly dependent on the waveguide cavity size are based on the real, objective and dynamically updated physical boundary morphology. From the source of the underlying physical architecture, it curbs the trend of microwave frequency drift and transmission insertion loss deterioration caused by high temperature thermal distortion.

[0127] The calculation of the dynamic dielectric constant based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference includes:

[0128] Compensation based solely on physical geometry cannot cover the degradation of dielectric properties of high-frequency substrates caused by temperature. The purpose of this step is to introduce a temperature sensitivity coefficient for dielectric constant to quantify the effect of thermal gradient on the material's permittivity distribution.

[0129] At the start of the system steps, the initial dielectric constant of the filter is obtained. Set the temperature sensitivity coefficient of the dielectric constant. The value is based on the temperature coefficient specified by the manufacturer for the microwave dielectric substrate. This value directly determines the sensitivity weight of the subsequent electromagnetic response to the temperature field. The larger the value, the more severe the aging and deterioration of the dielectric.

[0130] The system uses the thermal field distribution calculated in the previous step to directly calculate the actual dielectric constant of each monitoring point. The calculation formula is as follows:

[0131] In the formula, For the first The dynamic dielectric constant of each monitoring point; The initial dielectric constant; The dielectric constant is the temperature sensitivity coefficient. For the first Real-time temperature difference at each monitoring point.

[0132] By extracting the initial dielectric parameters and inherent temperature-sensitive constant of the waveguide dielectric material, and performing a rigorous product operation between this sensitive constant and the real-time temperature difference obtained in the previous step, the attenuation drift rate of the insulating dielectric properties under the continuous action of a high-frequency strong thermal field is accurately extracted. Furthermore, the dielectric performance degradation caused by this drift rate is subtracted from the initial dielectric parameters, thereby precisely establishing the dynamic dielectric parameters under the current extreme thermal environment. This computational mechanism deeply analyzes the physical mechanism of electrical performance degradation of the insulating filling medium inside the waveguide filter when it approaches its critical physical point at high temperatures. It breaks the rigid and idealistic setting of the dielectric constant of the insulating material as constant in conventional high-frequency microwave electromagnetic analysis, fully considering that local hot spots not only cause expansion of the surrounding metal structure but also lead to core micro-... The nonlinear weakening of the charge distribution energy storage and dielectric polarization capability within the wave transmission medium is addressed by a dynamic multiphysics mapping method that directly transforms macroscopic temperature field parameters into microscopic dielectric polarization characteristic attenuation. This method can track in real time the spatial distribution distortion of internal permittivity caused by long-term thermal aging or transient drastic temperature rise in the insulating substrate. It can lock in advance and accurately quantify the potential broadband electromagnetic field distribution disorder caused by the deterioration of the material's internal microscopic properties. This allows the filter adaptive control system to grasp the real electrical operating state of the internal core dielectric layer in real time, providing indispensable material-level underlying physical parameters for subsequent phase velocity shift assessment and accurate correction of wave impedance mismatch. This greatly improves the environmental dynamic adaptive capability and physical state simulation fidelity of the high-frequency impedance matching model under extremely complex thermodynamic boundary conditions.

[0133] The calculation of the corrected phase velocity based on the absolute physical constant of permeability and the dynamic dielectric constant includes:

[0134] The change in dielectric constant directly affects the phase velocity of electromagnetic waves propagating within the filter, thus causing a drift in the resonant frequency. This step combines the absolute permeability with the dynamic dielectric constant calculated in the previous step to determine the actual microwave propagation phase velocity under the current environment.

[0135] Obtaining the vacuum constant and permeability ;

[0136] The corrected phase velocity is calculated using the following formula:

[0137] In the formula, To correct the phase velocity; For the first The dynamic dielectric constant of each monitoring point; is the magnetic permeability.

[0138] By calling the preset absolute physical constant permeability in the system's underlying control chip, and performing a rigorous product fusion calculation with the dynamic medium parameters accurately obtained based on the real-time thermal field distribution, and then solving the square root reciprocal equation of the physical product result, the true corrected propagation phase velocity of the microwave signal in the waveguide resonant cavity under the current severe thermal interference environment is derived. This calculation process closely follows the basic principles of classical microwave electromagnetic wave dynamics, directly linking and projecting the microscopic degradation effect of the insulating medium properties caused by extreme high-temperature thermodynamic conditions onto the core physical kinematic characteristics of macroscopic electromagnetic waves. It accurately captures and quantifies the nonlinear shift in microwave high-frequency transmission phase velocity that is inevitably caused by the decrease in dielectric constant, effectively avoiding the need to adhere to vacuum or room-temperature standard microwave waves under actual complex high-power high-frequency full-load conditions. The problem of severe misjudgment of the center frequency of broadband high-frequency resonance caused by high speed enables the impedance compensation and control system to accurately grasp the true dynamic propagation speed of high-frequency electromagnetic waves when they travel through non-uniform media subjected to physical distortion. This real-time calibration and correction mechanism for the underlying wave dynamics velocity parameters not only reflects the direct physical influence of the accelerated aging of insulating dielectric materials on the hysteresis effect of high-frequency microwave beam transmission in a realistic and detailed manner, but also provides a decisive transmission speed fundamental variable for the subsequent construction of dynamic multidimensional electromagnetic boundary and impedance abrupt change scattering analysis model. It completely eliminates the underlying static error of phase velocity when performing complex microwave network node analysis under dynamic alternating strong thermal field conditions, and significantly enhances the energy transmission phase stability and overall frequency response accuracy of high-power power supply filters when processing high-frequency broadband pulse signals under complex and severe temperature variation conditions.

[0139] The calculation of dynamic characteristic impedance based on permeability, corrected phase velocity, and cavity length change ratio, combined with the initial characteristic impedance to calculate gradient boundary reflection coefficient, includes:

[0140] After the phase velocity shifts, the non-uniform dielectric distribution inside the medium will form a false electromagnetic boundary, leading to microwave reflection and scattering, and increasing insertion loss.

[0141] At this point, the system first obtains the initial cavity feature length. ;

[0142] The current dynamic characteristic impedance is then calculated using the following formula:

[0143] In the formula, It is the dynamic characteristic impedance; Permeability; To correct the phase velocity; The compensated cavity feature length; The initial cavity feature length;

[0144] Subsequently, the system obtains the initial characteristic impedance of the filter. ;

[0145] By utilizing the mismatch between the dynamic characteristic impedance and the initial steady-state impedance, the reflection intensity of electromagnetic waves at the interface is quantified, and the gradient boundary reflection coefficient is calculated. The calculation formula is as follows:

[0146] In the formula, The gradient boundary reflection coefficient; It is the dynamic characteristic impedance; This is the initial characteristic impedance.

[0147] By multiplying the absolute physical constant permeability by the precisely corrected microwave propagation speed, and innovatively introducing a geometric ratio between the compensated dynamic characteristic length (incorporating thermodynamic deformation factors) and the initial system length, a spatial proportional weighting correction is applied to the velocity product. This yields a dynamic characteristic impedance that balances the distortion caused by both microscopic dielectric material degradation and macroscopic metal expansion. Subsequently, a deviation analysis is performed between this disturbed dynamic impedance and the filter's factory-standard initial impedance. The algebraic difference between the two is divided by the sum of their physical impedances to obtain the high-frequency gradient boundary reflection coefficient. This logical design deeply and thoroughly analyzes the spatial abrupt change mechanism of microwave transmission path impedance induced by non-uniform and drastic temperature rise. It restores the originally textual and homogenized internal physical waveguide cavity of the filter to a reality where a large number of non-uniform dielectric gradients and microwave wave velocity folding phenomena objectively exist. By precisely quantifying the relative intensity of high-frequency wave reflections that inevitably occur when high-energy electromagnetic waves cross these illusory medium boundaries formed by the aggregation of local non-uniform hot spots at complex physical boundaries, this method completely overcomes the theoretical blind spots of the traditional one-size-fits-all impedance blind matching method. This allows the closed-loop control system to accurately predict and control how much microwave load energy will be mercilessly bounced back due to the slight mismatch in the internal transmission impedance chain. This quantitative assessment and model deconstruction of the internal electromagnetic microwave spatial scattering intensity establishes a solid and irrefutable theoretical microwave network basis for subsequent system-level high-frequency energy dissipation calculations and predictions of deterioration trends. It fundamentally reveals the internal hidden wave dynamics that cause the sharp and abnormal increase in the total insertion loss of the system under extreme high temperature and high pressure environments, and points to a deterministic feedback control direction for achieving adaptive and precise high-frequency impedance correction command output.

[0148] Reference Figure 4 The step of calculating the interface scattering loss density based on the initial input microwave power density and the reflection coefficient, and calculating the reference current density threshold based on the cavity cross-sectional area, includes:

[0149] This step is to clarify the direct microwave energy loss caused by the abrupt change in wave impedance.

[0150] Obtaining microwave power density ;

[0151] The electromagnetic wave scattering loss per unit area is evaluated by directly multiplying the power density of the input microwave by the square of the reflection coefficient. The calculation formula is as follows:

[0152] In the formula, This represents the interface scattering loss density. Microwave power density; The gradient boundary reflection coefficient;

[0153] To subsequently determine the extent to which the transient high-frequency pulse current disrupts the skin effect, the system at this point acquires the cross-sectional area of ​​the cavity. ;

[0154] This step requires establishing a safety threshold. The reference current density threshold of the system is calculated using the power characteristic impedance equation. The calculation formula is as follows:

[0155] In the formula, The reference current density threshold; Microwave power density; The initial characteristic impedance; This is the cross-sectional area of ​​the cavity.

[0156] By real-time reading of the initial external microwave total power density input to the acquisition system under current operating conditions, and directly multiplying it by the square term of the high-frequency interface reflection coefficient derived from the previous core logic, the interface scattering loss density dissipated per unit physical cross-sectional area due to discontinuous changes in internal microwave impedance is quantified. Simultaneously, by dividing the acquired input microwave power density by the physical ratio of the product of the initial standard characteristic impedance and the waveguide cavity cross-sectional area, and performing a square root algebraic calculation, a dynamic reference current density physical threshold for the system at the current high-intensity microwave energy input level is rigorously established. This calculation process effectively transforms the highly abstract electromagnetic wave interface reflection coefficient into a visible and controllable macroscopic high-frequency heat dissipation power index for engineering applications, intuitively revealing the true attenuation of high-frequency ineffective microwave energy caused by physical distortions in extreme environments. The precise scale allows the subsequent impedance compensation network to accurately determine the exact size of the microwave energy gap that the system urgently needs to compensate for. The establishment of the dynamic reference current density threshold based on power calculation defines a stringent physical safety red line for the surge of extreme high-frequency pulse current on the inner surface of the waveguide. It cleverly binds high-frequency power parameters, characteristic impedance, and physical spatial cross-sectional area depth together, constructing a current safety criterion reference system that can dynamically follow the fluctuation of input power. This not only provides a unified evaluation scale for accurately measuring the microscopic destructive force of transient surge current, but also significantly improves the dynamic adaptation range of the entire control system when facing irregular fluctuations in pulse power. It effectively avoids the problem of low-frequency compensation sensitivity mismatch or high-frequency overcompensation oscillation caused by using a single fixed and rigid alarm threshold.

[0157] The calculation of dynamic skin depth based on power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold includes:

[0158] The above-mentioned equivalent matching of scattering loss is based on steady-state current; however, high-frequency pulsed current will squeeze the distribution of charge on the conductor surface, resulting in a sharp reduction in the conductive cross-sectional area.

[0159] At the start of this step, the system acquires real-time transient high-frequency current through sensors. And simultaneously acquire the externally set power pulse angular frequency. and the initial conductivity of the substrate itself ;

[0160] Then, combining the obtained threshold and scattering reflection coefficient, the dynamic skin depth under the current high-frequency alternating magnetic field is derived, and the calculation formula is as follows:

[0161] In the formula, For dynamic skin depth; The power pulse angular frequency; Permeability; The initial conductivity; The gradient boundary reflection coefficient; It is a transient high-frequency current; The cross-sectional area of ​​the cavity; The reference current density threshold is used.

[0162] By comprehensively utilizing externally input pulse alternating angular frequency, constant permeability, and initial conductivity parameters, the normal standard conductive thickness under classical high-frequency electromagnetic theory is determined. The absolute intensity of transient high-frequency pulse currents is captured in real time, and their instantaneous density on the waveguide cross-sectional area is rigorously calculated. Then, this instantaneous current density is compared with a previously established dynamic current threshold using bias ratio analysis. Finally, a nonlinear reduction operation is performed on the standard conductive thickness using the interface microwave reflection coefficient to obtain the dynamic skin depth under harsh operating conditions. This core strategy creatively extends the steady-state skin effect to complex extreme transient high-frequency pulse strong coupling conditions, keenly capturing the strong squeezing effect of ultra-high-frequency alternating magnetic fields on the charge distribution on the metal surface. This squeezing directly leads to the actual... The metal cross-section that effectively participates in conduction undergoes a precipitous shrinkage under transient high-power pulse impact. This step not only quantitatively analyzes the geometric attenuation degree of this skin depth, but also deeply correlates it with the severe alternating wave reflection interference phenomenon inside. It truly reflects the secondary deterioration effect of the complex standing wave field formed by the superposition of reflected microwaves and traveling waves on the geometry of the microscopic conductive channel. It effectively fills the technical gap that has long been ignored in the microscopic effect of transient current distortion in conventional filter design. It enables the control system to accurately perceive the real-time reduction of the electron transmission path at the microscopic level, providing a precise calculation basis at the microscopic physical level for the accurate adjustment of the overall macroscopic conductivity performance, and greatly enhancing the physical limit of the communication system's ability to withstand destructive surge pulse impacts.

[0163] The calculation of transient equivalent conductivity based on initial conductivity, dynamic skin depth, and initial cavity characteristic length, followed by the calculation of conductivity correction factors, includes:

[0164] Changes in skin depth directly lead to a loss in macroscopic conductivity during transient processes. This step aims to establish an effective conductivity mapping using geometric scale ratios;

[0165] The transient equivalent conductivity is obtained by comparing the dynamic skin depth with the initial cavity feature length and normalizing the calculation. The calculation formula is as follows:

[0166] In the formula, Transient equivalent conductivity; The initial conductivity; For dynamic skin depth; The initial cavity feature length;

[0167] Next, in order to output to the impedance compensation control network, the system needs to convert the transient equivalent conductivity into a dimensionless control factor, that is, to obtain the conductivity correction factor, calculated as follows:

[0168] In the formula, This is the conductivity correction factor; Transient equivalent conductivity; The initial conductivity is given.

[0169] By extracting the initial macroscopic set dimensions of the physical waveguide cavity, the dynamic skin depth obtained by the strong compression of microscopic high-frequency pulses is compared with the macroscopic feature length using a cross-physical-level geometric proportional normalization. This geometric discount attenuation rate is then directly applied to the initial conductivity parameters of the material itself to calculate the severely weakened transient equivalent conductivity parameters under the current extreme conditions. Furthermore, this damaged equivalent conductivity parameter is divided by the initial reference parameter to precisely extract a pure proportional characteristic conductivity correction factor completely stripped of specific physical dimensions. This conversion process cleverly bridges the characterization gap between the microscopic high-frequency charge skin distribution and the macroscopic material conductivity properties, allowing the microscopic skin depth, which is difficult to directly access and read by external hardware, to be accurately determined. The degree variation value is smoothly transformed into a scalar of material conductivity degradation coefficient with universal engineering control significance. This thorough physical dimensionality reduction and dimensionless processing method not only greatly simplifies the concurrent calculation burden of downstream impedance matching compensation network chips, so that the control system no longer needs to be deeply involved in the analysis of complex microscopic three-dimensional electromagnetic fields, but also the pure digital correction factor intuitively and accurately reflects the percentage of comprehensive conductivity loss caused by high-frequency transient high current impact. It provides an extremely effective standardized interface for the overall microwave control system to find a fast linear compensation fulcrum in the intricate nonlinear power attenuation model, and significantly improves the cross-domain computation efficiency and interface compatibility of the transition from the underlying physical model to the hardware digital control sequence.

[0170] The process of calculating the compensated matching impedance based on the dynamic characteristic impedance, correction factor, and loss density, and generating an impedance compensation control sequence based on the difference between the compensated and initial characteristic impedance, includes:

[0171] This step completes the closed-loop control output of the technical solution; at this point, all basic data has been collected. The system integrates the previously generated dynamic impedance bias, interface scattering loss ratio, and transient conductivity correction factor to calculate the ultimate matching impedance that can offset the distortion of the above multidimensional physical quantities. The calculation formula is as follows:

[0172] In the formula, The compensated matching impedance; It is the dynamic characteristic impedance; This is the conductivity correction factor; This represents the interface scattering loss density. The initial input microwave power density;

[0173] Finally, the system performs direct differential processing on the theoretical matching impedance and the steady-state initial characteristic impedance buffered in the system's preceding steps, outputting a direct command sequence to control the digital synthesized impedance network's actions, thereby achieving adaptive optimization of the insertion loss output; the formula for this final sequence is as follows:

[0174] In the formula, The final target is to output a data impedance compensation control sequence; The compensated matching impedance; This is the initial characteristic impedance.

[0175] By comprehensively integrating the dynamic characteristic impedances derived progressively from various physical dimensions and the dimensionless conductivity correction factor, and simultaneously introducing the actual energy transfer residual ratio established based on the ratio of scattering loss components to the original input energy density, the aforementioned product results are subjected to deep secondary global weighting to obtain the target matching impedance value urgently needed under ideal conditions. Finally, the target matching impedance value is subtracted from the initial reference impedance to obtain the precise absolute deviation vector between the two, which is then output as the unique control sequence. This comprehensive step completes the final convergence from multi-physics discrete quantitative analysis to a single deterministic control command at the end, addressing physical displacements caused by thermodynamic geometric deformation and material thermal aging. The resulting dielectric constant degradation, impedance wave reflection dissipation induced by dielectric gradient, and skin layer conductivity loss caused by transient pulse surge current compression are all packaged into a rigorous ultimate compensation equation system. This system completely connects the entire control chain from abnormal parameter acquisition to physical mechanism analysis to execution sequence generation, ensuring that the output impedance compensation control sequence contains the counterbalancing components of all extreme deterioration factors. This allows the digital synthesized impedance network to perform reverse impedance compensation actions with this unique and precise control command, forcibly flattening the impedance bias curve in complex extreme conditions. This perfectly achieves the ultimate design goal of minimizing the dynamic insertion loss of high-frequency power filters and approaching the ideal microwave energy transmission efficiency.

[0176] It should be noted that, in this document, relational terms such as "first" and "second" are used only to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such process, method, article, or apparatus.

[0177] The above description is only a preferred embodiment of the present invention. It should be noted that for those skilled in the art, several improvements and modifications can be made without departing from the technical principles of the present invention, and these improvements and modifications should also be considered within the scope of protection of the present invention.

Claims

1. A method of power filter insertion loss optimization design, characterized by, include: The real-time temperature difference is calculated based on the real-time absolute temperature and the reference ambient temperature. The compensated cavity characteristic length is then calculated by combining the initial cavity characteristic length and the cavity thermal expansion coefficient. The dynamic dielectric constant is calculated based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference. The corrected phase velocity is calculated based on the absolute physical constant of magnetic permeability and the dynamic dielectric constant. The dynamic characteristic impedance is calculated based on the permeability, the corrected phase velocity, and the cavity length change ratio. The gradient boundary reflection coefficient is then calculated in conjunction with the initial characteristic impedance. The interface scattering loss density is calculated based on the initial input microwave power density and the reflection coefficient, and the reference current density threshold is calculated in combination with the cavity cross-sectional area. The dynamic skin depth is calculated based on the power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold. The transient equivalent conductivity is calculated based on the initial conductivity, dynamic skin depth and initial cavity characteristic length, and then the conductivity correction factor is calculated. The compensated matching impedance is calculated based on the dynamic characteristic impedance, correction factor, and loss density, and an impedance compensation control sequence is generated based on the difference between the compensated and initial characteristic impedances.

2. The power filter insertion loss optimization design method of claim 1, wherein, The calculation of the real-time temperature difference based on the real-time absolute temperature and the reference ambient temperature, combined with the initial cavity characteristic length and the cavity thermal expansion coefficient, to calculate the compensated cavity characteristic length includes: A temperature sensor is deployed inside the resonant cavity of a waveguide-type power filter. Acquire real-time absolute temperature data and ambient baseline temperature data for each monitoring point; The real-time temperature difference is obtained by performing a differential calculation between the monitored temperature and the reference temperature. The length change is determined by the product of the initial cavity length, thermal expansion parameters, and the real-time temperature difference; The initial cavity length is summed with the change to obtain the compensated characteristic length.

3. The power filter insertion loss optimization design method according to claim 2, characterized in that, The calculation of the dynamic dielectric constant based on the initial dielectric constant, the temperature sensitivity coefficient of the dielectric constant, and the real-time temperature difference includes: Obtain the initial dielectric parameters and temperature sensitivity constant of the dielectric material; The product of the sensitivity constant and the real-time temperature difference is used to calculate the medium property drift rate. Subtract the product term of the initial dielectric parameter and the drift rate; Determine the dynamic medium parameters under the current thermal environment.

4. The power filter insertion loss optimization design method according to claim 3, characterized in that, The calculation of the corrected phase velocity based on the absolute physical constant of permeability and the dynamic dielectric constant includes: Retrieve the system's stored permeability constant; The constant is multiplied by the dynamic medium parameter; Perform the square root reciprocal calculation on the product result; The corrected propagation velocity within the waveguide cavity is obtained.

5. The power filter insertion loss optimization design method according to claim 4, characterized in that, The calculation of dynamic characteristic impedance based on permeability, corrected phase velocity, and cavity length change ratio, combined with the initial characteristic impedance to calculate gradient boundary reflection coefficient, includes: Multiply the permeability by the corrected propagation speed; The dynamic characteristic impedance is obtained by weighting the multiplication result using the ratio of the compensated characteristic length to the initial characteristic length. Calculate the deviation between the dynamic characteristic impedance and the initial impedance; Divide the deviation value by the sum of the impedances of the two to obtain the interface reflection coefficient.

6. The power filter insertion loss optimization design method according to claim 5, characterized in that, The calculation of interface scattering loss density based on the initial input microwave power density and the reflection coefficient, combined with the calculation of the reference current density threshold based on the cavity cross-sectional area, includes: Read the input microwave energy density; Calculate the product of the energy density and the square of the reflection coefficient to determine the scattering loss component; The square root is calculated based on the ratio of the input energy density to the product of the initial impedance and the cavity area. Generate a current density reference per unit area.

7. The power filter insertion loss optimization design method according to claim 6, characterized in that, The calculation of dynamic skin depth based on power pulse angular frequency, initial conductivity, reflection coefficient, transient high-frequency current, and the threshold includes: The standard conductive thickness is determined by multiplying the pulse angular frequency, permeability, and initial conductivity parameters. Monitor real-time high-frequency pulse current and calculate the real-time current intensity per unit area; Calculate the offset ratio between the real-time current intensity and the current density reference. By performing a nonlinear reduction calculation on the standard conductive thickness in conjunction with the interface reflection coefficient, the dynamic skin depth is obtained.

8. The power filter insertion loss optimization design method according to claim 7, characterized in that, The calculation of transient equivalent conductivity based on initial conductivity, dynamic skin depth, and initial cavity characteristic length, followed by the calculation of conductivity correction factors, includes: The equivalent conductivity parameter is obtained by multiplying the initial conductivity parameter by the geometric ratio of the dynamic skin depth to the initial cavity length. Divide the equivalent conductivity parameter by the initial conductivity parameter; output a dimensionless correction factor characterizing the degree of conductivity degradation.

9. The power filter insertion loss optimization design method according to claim 8, characterized in that, The process of calculating the compensated matching impedance based on the dynamic characteristic impedance, correction factor, and loss density, and generating an impedance compensation control sequence based on the difference between the compensated and initial characteristic impedance, includes: Calculate the product of the dynamic characteristic impedance and the dimensionless correction factor; The energy transfer ratio is determined by the ratio of the scattering loss component to the input energy density; The product is weighted twice by the energy transfer ratio to obtain the target matching impedance value; The deviation vector of the target matching impedance value relative to the initial reference impedance is calculated and output as the adjustment sequence.

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