A low-power modulation adaptive method and system based on data priority

Abstract

This invention proposes a low-power modulation adaptive method and system based on data priority, relating to the field of wireless communication technology. The method includes: performing priority analysis and extraction on the data block to be transmitted to obtain a priority tag; embedding a time-domain amplitude concave window into the data block to obtain a concave modulation frame; adding time-domain phase jitter features to the data block to obtain a jitter modulation frame; transmitting the concave modulation frame; analyzing and extracting the obtained airborne signal to obtain distortion feature parameters; scaling the base time slot increment proportionally based on the ratio of the concave distortion degree to the jitter distortion degree to obtain a time slot increment; and superimposing the time slot increment with the current waiting time slot to obtain a waiting time slot; when the waiting time slot is completed, granting retransmission authorization to the jitter modulation frame and concatenating and encapsulating it with the data block of the next data block to obtain a backoff modulation output frame. This invention can improve the efficiency of a low-power modulation adaptive method based on data priority.

Description

Technical Field

[0001] This invention relates to the field of wireless communication technology, and in particular to a low-power modulation adaptive method and system based on data priority. Background Technology

[0002] In current wireless communication systems, carrier sensing-based channel contention mechanisms can avoid collisions by detecting the busy / idle state of the channel under normal signal-to-noise ratio (SNR) conditions. However, when the communication link is in an environment with extremely low SNR, the background noise power is much higher than the signal power, and the false alarm rate and missed detection rate of energy detection and signal analysis increase sharply, causing the carrier sensing mechanism to essentially fail. Furthermore, traditional fixed-priority-based differentiated scheduling methods typically rely on pre-configured priority tags from upper-layer protocols or the core network, lacking effective utilization of real-time data content characteristics, making it difficult to achieve adaptive and efficient channel allocation in distributed multi-node access scenarios. When multiple nodes simultaneously compete for the same channel resource, most existing differentiated modulation methods employ a pre-set priority queue with fixed backoff parameters. The backoff strategy is often a discrete, random mechanism with jumps, lacking closed-loop coupling between collision feedback and priority control. This makes it difficult to simultaneously achieve the timeliness and low power consumption goals of high-priority data transmission, resulting in a significant decrease in overall system throughput and energy efficiency under extreme conditions.

[0003] On the other hand, existing collision backoff mechanisms generally employ truncated binary exponential backoff algorithms and their variants. These methods exponentially expand the backoff window based on the number of collisions and randomly select the delay duration within the window. The backoff step size is only related to the number of collisions and not to the actual collision intensity. When the channel load remains high, a polarization occurs: either the backoff window is too large, leading to prolonged channel idleness, or the backoff window is too small, resulting in repeated collisions. This leads to drastic fluctuations in channel utilization. Some improved schemes introduce priority factors to simply weight and adjust the backoff parameters, but the backoff process still relies on a random number generator. The backoff time decisions of each node are independent, lacking quantitative utilization of channel feedback information and a memory-based update mechanism. Under extreme communication conditions of high dynamics and high collision intensity, it is difficult to quickly converge to a stable state. Using high-order modulation to improve spectral efficiency places extremely high demands on the quality of the received signal. The bit error rate rises sharply under extremely low signal-to-noise ratio, leading to frequent retransmissions, which in turn significantly increases power consumption. Existing technologies lack a complete solution that can simultaneously achieve accurate collision intensity sensing, priority-driven differentiated backoff control, and a low-overhead retransmission mechanism. Therefore, how to improve channel access efficiency has become an urgent problem to be solved. Summary of the Invention

[0004] This invention provides a low-power modulation adaptive method and system based on data priority to solve the problems mentioned in the background art.

[0005] To achieve the above objectives, the present invention provides a low-power modulation adaptive method based on data priority, comprising: S01. Perform priority analysis and extraction on the data block to be sent to obtain the priority label of the data block to be sent; S02. Embed a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and add time-domain phase jitter features to the data block to obtain a jitter modulation frame of the data block. S03. Transmit the concave modulation frame, analyze and extract the transmitted airborne signal to obtain the distortion characteristic parameters of the airborne signal; S04. Based on the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameter, the base time slot increment is mapped. The base time slot increment is scaled proportionally according to the priority label to obtain the time slot increment of the data block. The time slot increment is then superimposed with the current waiting time slot to obtain the waiting time slot of the data block to be sent. S05. When the waiting time slot is completed, the jitter modulation frame is granted retransmission authorization and concatenated with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted.

[0006] In a preferred embodiment, the step of performing priority analysis and extraction on the data block to be sent to obtain the priority tag of the data block to be sent includes: The service type is parsed from the header field of the data block to be sent to obtain the service level indication of the data block to be sent. The redundancy ratio of the data block to be sent is obtained by performing a duplicate count on the payload of the data block to be sent. The remaining lifetime of the embedded timestamp of the data block to be sent is determined to obtain the urgency level of the data block to be sent; Based on the service level indication, information redundancy ratio, and urgency level, a comprehensive evaluation is performed on the service level indication, information redundancy ratio, and urgency level to obtain the priority label of the data block to be sent.

[0007] In a preferred embodiment, embedding a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block includes: The data block is subjected to an energy envelope scan. The obtained envelope trough position is marked as the indentation window start mark, and the obtained trough duration length is marked as the indentation window duration width. Based on the indentation window start mark and the indentation window duration, the data block is amplitude shaped according to the raised cosine roll-off profile to obtain the time-domain amplitude decay window function of the data block. Based on the time-domain amplitude attenuation window function, the amplitude of each sampling point of the data block is reduced by a multiplier to obtain the concave modulation frame of the data block.

[0008] In a preferred embodiment, the step of adding time-domain phase jitter features to the data block to obtain a jitter-modulated frame of the data block includes: The linear feedback shift register of the data block is subjected to tap XOR feedback, and the resulting shift register is injected into bits and shifted out bit by bit to obtain the binary phase perturbation sequence of the linear feedback shift register; The phase step size of the binary phase perturbation sequence is obtained by performing a Gray mapping on the binary phase perturbation sequence. Based on a preset phase quantization modulus, the phase step value is accumulated and then truncated cyclically to obtain the phase rotation angle value of the phase step value. Based on the phase rotation angle value, a rotation matrix multiplication is performed on the in-phase and quadrature components of each constellation symbol of the data block using a complex multiplier to obtain the jitter modulation frame of the data block.

[0009] In a preferred embodiment, the phase rotation angle value is calculated using the following formula: ; In the formula, For the first The phase rotation angle value of each constellation symbol, The number of phase quantization bits for the phase rotation angle value. The first of the binary phase perturbation sequence The output of the clock cycle 1 The value of the bit, For bitwise XOR operation, This is a function to count the number of 1s in a binary vector. For the first linear feedback shift register A state vector for each clock cycle. The first of the binary phase perturbation sequence A vector composed of outputs over clock cycles. This is a bitwise AND operation.

[0010] In a preferred embodiment, the step of transmitting the concave modulation frame and analyzing and extracting the transmitted airborne signal to obtain the distortion characteristic parameters of the airborne signal includes: The concave modulation frame is up-converted and pushed to obtain the radio frequency transmission waveform of the concave modulation frame; In the shared channel, the transmitting antenna is excited and radiated based on the radio frequency bearer waveform to obtain the air propagation signal of the concave modulation frame; The air-borne signal is down-converted and demodulated to obtain the baseband recovered waveform of the air-borne signal; Energy sampling is performed on the concave regions in the baseband recovery waveform to obtain the local power sequence of the baseband recovery waveform; The amplitude of the concave regions in the local power sequence is compared to obtain the degree of concave distortion of the air-propagating signal; The jitter distortion degree of the airborne signal is obtained by performing phase differential correlation on the local power sequence.

[0011] In a preferred embodiment, the ratio of concave distortion to jitter distortion in the distortion feature parameters is mapped to a base time slot increment. This base time slot increment is then scaled proportionally according to the priority label to obtain the time slot increment for the data block. Finally, this time slot increment is superimposed with the current waiting time slot to obtain the waiting time slot for the data block to be sent. This process includes: Based on a preset ratio threshold group, the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters are successively approximated and compared to obtain the distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion. Based on the distortion ratio interval index, the address of the read-only memory is decoded, and the pre-fixed time slot step value is extracted from the read-only memory to obtain the basic time slot increment of the data block to be sent. Based on the priority label, the basic time slot increment is proportionally weighted to obtain the time slot increment of the data block; The current waiting time slot is accumulated and merged with the time slot increment to obtain the waiting time slot of the data block to be sent.

[0012] In a preferred embodiment, the step of successively approximating and comparing the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters based on a preset ratio threshold group to obtain a distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion includes: The degree of concave distortion is compared with the degree of jitter distortion by a fixed multiple, and the multiple level at which the inequality first holds true is recorded to obtain the coarse adjustment ratio level of the degree of concave distortion to the degree of jitter distortion. Based on a preset ratio threshold group, the two adjacent boundary thresholds of the coarse adjustment ratio level are extracted to obtain the upper boundary threshold and lower boundary threshold of the coarse adjustment ratio level. The difference between the degree of concave distortion and the lower boundary threshold is amplified, and the amplified result is compared with the difference to obtain the fine-tuning offset level between the degree of concave distortion and the degree of jitter distortion. By bit-joining the coarse adjustment ratio level and the fine adjustment offset level, an index of the distortion ratio range between the degree of concave distortion and the degree of jitter distortion is obtained.

[0013] In a preferred embodiment, when the waiting time slot is completed, granting retransmission authorization to the jitter modulation frame and concatenating it with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted includes: The waiting time slot is monitored by a decreasing count to obtain the zero-trigger pulse of the waiting time slot; Based on the zero-trigger pulse, the transmit permission is set for the jitter modulation frame to obtain the repeat transmit authorization signal for the jitter modulation frame; The data sequence of the next data block is buffered and read to obtain the symbol stream to be concatenated for the next data block; Based on the repeated transmission authorization signal, the jitter modulation frame is joined end-to-end with the symbol stream to be spliced ​​to obtain the backoff modulation output frame of the data block to be transmitted.

[0014] To address the above problems, the present invention also provides a low-power modulation adaptive system based on data priority, the system comprising: The priority parsing module is used to perform priority analysis and extraction on the data block to be sent, and obtain the priority label of the data block to be sent; A dual-feature embedding module is used to embed a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and to add time-domain phase jitter features to the data block to obtain a jitter modulation frame of the data block. The channel collision sensing module is used to transmit the concave modulation frame, analyze and extract the transmitted air propagation signal, and obtain the distortion feature parameters of the air propagation signal. The backoff decision and threshold evolution module is used to map the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameters to the basic time slot increment, scale the basic time slot increment according to the priority label to obtain the time slot increment of the data block, and superimpose the time slot increment with the current waiting time slot to obtain the waiting time slot of the data block to be sent. The retransmission merging and frame assembly module is used to grant retransmission authorization to the jitter modulation frame when the waiting time slot is completed, and to splice and encapsulate it with the data block of the next data block to obtain the backoff modulation output frame of the data block to be sent.

[0015] Compared with the prior art, the present invention has the following beneficial effects: 1. This technical solution is specifically designed for distributed communication scenarios where multiple devices share the same wireless channel under extremely low signal-to-noise ratio conditions and where carrier sensing or centralized scheduling is not feasible. Within this application scope, by actively embedding time-domain amplitude concave windows and time-domain phase jitter features into the data blocks to be transmitted, and extracting the ratio of concave distortion to jitter distortion based on the air propagation echo of the transmitted signal, accurate collision intensity perception without channel estimation is achieved. This fundamentally eliminates the dependence on carrier sensing and network signaling interaction, enabling communication nodes to autonomously acquire channel contention state information even in extreme environments where the signal is submerged in noise.

[0016] 2. This scheme uses collision intensity ratio mapping as the base time slot increment, and then scales the base time slot increment proportionally according to the priority label of the data block. The scaled increment is then accumulated and updated with the current waiting time slot, forming a closed-loop backoff decision mechanism with collision memory effect, avoiding channel utilization fluctuations caused by traditional random backoff. Simultaneously, after backoff, only the pre-constructed jitter-modulated frame is retransmitted and directly concatenated with the data sequence of the next data block for output, hiding the retransmission overhead in the initial transmission process, significantly reducing the average transmit power required for a single effective data transmission. This mechanism enables the system to maintain low-latency access for high-priority data while ensuring stable overall channel throughput and substantially improving energy efficiency. Attached Figure Description

[0017] Figure 1 A flowchart illustrating a low-power modulation adaptive method based on data priority, provided in an embodiment of the present invention; Figure 2 A functional block diagram of a low-power modulation adaptive system based on data priority is provided in an embodiment of the present invention; The objectives, features, and advantages of this invention will be further explained in conjunction with the embodiments and with reference to the accompanying drawings. Detailed Implementation

[0018] It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention.

[0019] This application provides a low-power modulation adaptive method based on data priority. The executing entity of this low-power modulation adaptive method based on data priority includes, but is not limited to, at least one of the following electronic devices that can be configured to execute the method provided in this application: a server, a terminal, etc. In other words, the low-power modulation adaptive method based on data priority can be executed by software or hardware installed on a terminal device or a server device. The server includes, but is not limited to, a single server, a server cluster, a cloud server, or a cloud server cluster. The server can be an independent server or a cloud server that provides basic cloud computing services such as cloud services, cloud databases, cloud computing, cloud functions, cloud storage, network services, cloud communication, middleware services, domain name services, security services, content delivery networks (CDNs), and big data and wireless communication platforms.

[0020] Reference Figure 1 The diagram shown is a flowchart illustrating a low-power modulation adaptive method based on data priority according to an embodiment of the present invention. In this embodiment, the low-power modulation adaptive method based on data priority includes: S01. Perform priority analysis and extraction on the data block to be sent to obtain the priority label of the data block to be sent.

[0021] In this embodiment of the invention, the step of performing priority analysis and extraction on the data block to be sent to obtain the priority tag of the data block to be sent includes: The service type is parsed from the header field of the data block to be sent to obtain the service level indication of the data block to be sent. The redundancy ratio of the data block to be sent is obtained by performing a duplicate count on the payload of the data block to be sent. The remaining lifetime of the embedded timestamp of the data block to be sent is determined to obtain the urgency level of the data block to be sent; Based on the service level indication, information redundancy ratio, and urgency level, a comprehensive evaluation is performed on the service level indication, information redundancy ratio, and urgency level to obtain the priority label of the data block to be sent.

[0022] The specific implementation process for parsing the service type of the header field of the data block to be sent to obtain the service level indication (SSI) of the data block is as follows: The differential service code point field located at a fixed offset position in the data block to be sent, which occupies six bits, is read. The binary value of these six bits is directly used as the SSI, which is an integer ranging from 0 to 63. A larger value indicates a higher forwarding priority for the data block in regular network scheduling. After parsing, the SSI is stored in a local register.

[0023] The specific implementation process for calculating the information redundancy ratio of the payload of the data block to be sent is as follows: The payload portion of the data block to be sent is expanded bit by bit, and scanned bit by bit starting from the first bit. A counter for the current longest consecutive identical bit string length and a global longest consecutive identical bit string length register are set up. Each time a bit identical to the previous bit is encountered, the current counter is incremented. When a different bit is encountered, the value of the current counter is compared with the global register. If the current counter is larger, the global register is updated. Then, the current counter is cleared and a new consecutive string is counted. After scanning the entire payload, the length of the longest consecutive identical bit string stored in the global register is divided by the total number of bits in the payload, and the resulting ratio is used as the information redundancy ratio.

[0024] The specific implementation process for determining the urgency level of the data block to be sent by judging its remaining lifetime using the embedded timestamp is as follows: Extract the embedded timestamp field from the data block to be sent. This field records the last deadline at which the data block is allowed to be successfully received. Obtain the current time indicated by the system clock of the current device, and subtract the current time from the deadline to obtain the remaining lifetime. Compare the remaining lifetime with three preset thresholds: the first threshold is the high urgency threshold, the second threshold is the medium urgency threshold, and the third threshold is the low urgency threshold. If the remaining lifetime is less than the high urgency threshold, output an urgency level value of 0; if the remaining lifetime is between the high and medium urgency thresholds, output an urgency level value of 1; if the remaining lifetime is between the medium and low urgency thresholds, output an urgency level value of 2; if the remaining lifetime is greater than the low urgency threshold, output an urgency level value of 3.

[0025] The specific implementation process for obtaining the priority label of the data block to be sent by comprehensively evaluating the service level indication, information redundancy ratio, and urgency level is as follows: The service level indication, information redundancy ratio, and urgency level are respectively input into three multipliers. The first multiplier multiplies the service level indication by a preset first weighting coefficient and outputs a first weighted value. The second multiplier multiplies the information redundancy ratio by a preset second weighting coefficient and outputs a second weighted value, where the second weighting coefficient is negative, so that a higher information redundancy ratio results in a smaller weighted value. The third multiplier multiplies the urgency level by a preset third weighting coefficient and outputs a third weighted value. The outputs of the three multipliers are fed into an accumulator for summation to obtain a comprehensive score. The comprehensive score is sequentially compared with multiple threshold values ​​in the priority level table, which contains four threshold values ​​corresponding to priority labels 0, 1, 2, and 3. The comparison proceeds upwards from the lowest threshold. When the comprehensive score is less than the current threshold value, the comparison stops, and the level index corresponding to the current threshold value is output as the priority label. If the overall score is greater than or equal to the highest threshold, then the highest level index is output.

[0026] The beneficial effects of this priority analysis and extraction method are that, in distributed communication scenarios with extremely low signal-to-noise ratios, it comprehensively evaluates priority tags through three dimensions: service type parsing, payload duplication statistics, and embedded timestamp remaining lifetime determination. This allows the priority to simultaneously reflect network scheduling level, data redundancy, and timeliness. The introduction of information redundancy ratio allows highly redundant data to receive lower priority, allowing for longer backoff and thus saving power. The embedded timestamp remaining lifetime determination ensures that critical data about to expire receives higher priority to reduce the drop rate, thereby providing accurate decision-making basis for subsequent backoff control and improving the rationality of channel resource allocation and data delivery success rate.

[0027] S02. Embed a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and add time-domain phase jitter features to the data block to obtain a jitter modulation frame of the data block.

[0028] In this embodiment of the invention, the embedding of a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block includes: The data block is subjected to an energy envelope scan. The obtained envelope trough position is marked as the indentation window start mark, and the obtained trough duration length is marked as the indentation window duration width. Based on the indentation window start mark and the indentation window duration, the data block is amplitude shaped according to the raised cosine roll-off profile to obtain the time-domain amplitude decay window function of the data block. Based on the time-domain amplitude attenuation window function, the amplitude of each sampling point of the data block is reduced by a multiplier to obtain the concave modulation frame of the data block.

[0029] The step of adding time-domain phase jitter features to the data block to obtain the jitter-modulated frame of the data block includes: The linear feedback shift register of the data block is subjected to tap XOR feedback, and the resulting shift register is injected into bits and shifted out bit by bit to obtain the binary phase perturbation sequence of the linear feedback shift register; The phase step size of the binary phase perturbation sequence is obtained by performing a Gray mapping on the binary phase perturbation sequence. Based on a preset phase quantization modulus, the phase step value is accumulated and then truncated cyclically to obtain the phase rotation angle value of the phase step value. Based on the phase rotation angle value, a rotation matrix multiplication is performed on the in-phase and quadrature components of each constellation symbol of the data block using a complex multiplier to obtain the jitter modulation frame of the data block.

[0030] The phase rotation angle value is calculated using the following formula: ; In the formula, For the first The phase rotation angle value of each constellation symbol, The number of phase quantization bits for the phase rotation angle value. The first of the binary phase perturbation sequence The output of the clock cycle 1 The value of the bit, For bitwise XOR operation, This is a function to count the number of 1s in a binary vector. For the first linear feedback shift register A state vector for each clock cycle. The first of the binary phase perturbation sequence A vector composed of outputs over clock cycles. This is a bitwise AND operation.

[0031] The specific implementation process for performing energy envelope scanning on the data block, marking the obtained envelope trough position as the indentation window start marker, and marking the obtained trough duration as the indentation window duration width is as follows: Calculate the absolute value of the amplitude of each sampling point of the data block to form an amplitude curve that changes over time. Slide a fixed-length observation window from left to right on this curve, calculating the average amplitude value within the window each time, and determining the center position of the window with the smallest average value as the envelope trough position. Record the sequence number of the envelope trough position in the sampling point sequence, and use this sequence number as the indentation window start marker. Simultaneously record the duration of the trough region, i.e., the total number of sampling points spanned within the sliding observation window from the sampling point where the amplitude begins to decrease at the local peak to the sampling point where the amplitude leaves the lowest point and continues to rise; where the amplitude begins to decrease is the local peak immediately to the left of the local trough window, and the recovery and rise point is the starting sampling point where the amplitude begins to rise from the bottom of the trough, and the interval completely includes the decreasing segment, the bottom segment, and the rising transition segment.

[0032] Within the selected local trough sliding observation window, the left local peak is taken as the left boundary of the concave interval. After the amplitude drops to the minimum value at the bottom of the trough, sampling points are traversed along the time axis one by one. Sampling points that meet the following dual conditions are prioritized as the right boundary of the rising segment of the concave window: the amplitude of the sampling point rises back to be equal to the amplitude of the local peak at the left boundary, and the continuous sampling points after this sampling point maintain this amplitude without any further amplitude increase. The determination logic of this boundary relies on the inherent characteristics of the raised cosine roll-off window function. The raised cosine profile requires that the amplitude at both ends of the concave interval be uniformly normalized to 1, the local peak at the left boundary is the highest point of amplitude in the interval, and the bottom of the trough is the lowest point of amplitude. The signal smoothly drops from the left boundary to the bottom of the trough and then rises symmetrically. In an ideal continuous time domain, there must exist sampling positions where the amplitude is completely equal to the peak at the left boundary.

[0033] To address extreme conditions such as amplitude deviation in discrete sampling after analog-to-digital conversion and the absence of strictly equal amplitude sampling points within the window, a two-level fault tolerance judgment rule is added: The first level is amplitude tolerance matching judgment, where the system pre-configures a minimum power tolerance threshold. Sampling points whose amplitude difference between the first amplitude and the left boundary local peak amplitude is less than this tolerance threshold, and whose subsequent continuous sampling points have stable amplitudes without significant increases, are equivalently identified as the right boundary of the rising segment of the concave window. The second level is a window bottom-out truncation rule. If no sampling point meets the conditions after tolerance matching, the last sampling point of the current local trough sliding observation window is directly forced as the right boundary of the rising segment of the concave window. The bottom-out truncation only slightly widens the concave window interval, does not lose the concave amplitude attenuation characteristics, and can still complete the extraction of distortion feature parameters of subsequent air propagation signals normally, without affecting all core functions such as channel collision intensity perception and backoff time slot calculation. The above complete determination process covers both ideal continuous signals and extreme scenarios of discrete sampling, ensuring that any local low-level sliding observation window can uniquely determine the complete time domain interval of the concave window, and the complete interval includes the falling segment, the valley segment, and the rising transition segment.

[0034] Based on the indentation window start mark and the indentation window duration, the data block is shaped according to the raised cosine roll-off profile to obtain the time-domain amplitude attenuation window function of the data block. The specific implementation process is as follows: Starting from the indentation window start mark and using the indentation window duration as the total length, an amplitude attenuation profile with a raised cosine roll-off shape is constructed. The amplitude value of this profile is 1 at its starting point, then smoothly decreases to the attenuation coefficient at the lowest point, and then smoothly increases to the amplitude value of 1 at the end point. The roll-off coefficient uses a preset smooth transition value, making the entire profile shape resemble a cosine curve with a central indentation. The amplitude value of the profile corresponding to each sampling point is calculated to form an amplitude sequence of the same length as the data block. In this sequence, the amplitude values ​​of sampling points outside the indentation window are all 1, and the amplitude values ​​of sampling points inside the indentation window decrease from 1 to the lowest point and then rise back to 1 according to the raised cosine roll-off law. This amplitude sequence is used as the time-domain amplitude attenuation window function.

[0035] The specific implementation process for reducing the amplitude of each sampling point of the data block to obtain the concave modulation frame based on the time-domain amplitude attenuation window function is as follows: The value of each sampling point of the data block and the amplitude value at the corresponding position in the time-domain amplitude attenuation window function are fed into the multiplier. The multiplier performs real-number multiplication and outputs the reduced sampling point value. For sampling points outside the concave window, since the window function value is 1, the multiplication output is equal to the original sampling point value. For sampling points within the concave window, the multiplication output is equal to the original sampling point value multiplied by an attenuation coefficient less than 1, thereby achieving amplitude reduction. All reduced sampling points are arranged in their original order to form a new sampling point sequence, which is used as the concave modulation frame.

[0036] The specific implementation process for obtaining the binary phase perturbation sequence of the linear feedback shift register by performing tap XOR feedback on the linear feedback shift register of the data block and shifting out the resulting shift register bits one by one is as follows: Initialize a linear feedback shift register consisting of eight storage units connected in series, with each storage unit storing one binary bit, and the initial value is set to zero. Read a preset seed value from the data block and write the eight bits of the seed value into the eight storage units of the register respectively. Determine the tap positions of the register as the fourth and seventh storage units. At each clock cycle, take out the bits in the fourth and seventh storage units and perform an XOR operation. The rule of the XOR operation is that if the two bits are the same, output zero; if they are different, output one. Inject the result of the XOR operation as the feedback bit into the highest bit of the register, i.e., the first storage unit. At the same time, shift out the bit in the lowest bit of the register, i.e., the eighth storage unit, as the output bit of the current clock cycle. Then, shift the bits in the first to seventh storage units of the register one bit to the right in sequence, discard the shifted-out bit in the eighth storage unit, and fill the first storage unit with the feedback bit. The above process is repeated for eight clock cycles, with one shift-out bit output in each cycle. The bits output in the eight clock cycles are arranged in chronological order to form an eight-bit binary sequence, which is used as the binary phase perturbation sequence of the linear feedback shift register.

[0037] The specific implementation process for obtaining the phase step value of the binary phase perturbation sequence by performing Gray mapping on the binary phase perturbation sequence is as follows: The eight bits in the binary phase perturbation sequence are grouped into four bit pairs in adjacent order. The first bit pair consists of the first and second bits, the second bit pair consists of the third and fourth bits, the third bit pair consists of the fifth and sixth bits, and the fourth bit pair consists of the seventh and eighth bits. A Gray mapping table is pre-established, containing four entries. The first entry corresponds to a bit pair of 0-0, which maps to a phase step value of 0; the second entry corresponds to a bit pair of 0-1, which maps to a phase step value of 1; the third entry corresponds to a bit pair of 1-1, which maps to a phase step value of 2; and the fourth entry corresponds to a bit pair of 1-0, which maps to a phase step value of 3. For each bit pair, the corresponding phase step value is searched in the Gray mapping table according to its binary combination. The four found phase step values ​​are arranged in sequence, and this sequence is used as the phase step value of the binary phase perturbation sequence.

[0038] The specific implementation process for obtaining the phase rotation angle value of the phase step value by accumulating the phase step value based on a preset phase quantization modulus is as follows: A fixed positive integer of four is preset as the phase quantization modulus. An accumulator is set with an initial value of zero. The first phase step value is read sequentially from the phase step value sequence and added to the value in the accumulator to obtain the sum. It is determined whether the sum is greater than or equal to four. If the sum is greater than or equal to four, four is continuously subtracted from the sum, checking the remaining value after each subtraction to ensure it is still greater than or equal to four, until the remaining value is less than four. This remaining value is then used as the new accumulator value. If the sum is less than four, the sum is directly used as the new accumulator value. The updated accumulator value is output as the phase rotation angle value corresponding to the current step value. Then, the second phase step value is read, and the above accumulating and truncating process is repeated to obtain the second phase rotation angle value. Similarly, after processing all four phase step values, the four phase rotation angle values ​​are arranged in order to obtain the phase rotation angle value sequence of the phase step values.

[0039] Based on the phase rotation angle values, the specific implementation process of obtaining the jitter modulation frame of the data block by performing rotation matrix multiplication on the in-phase and quadrature components of each constellation symbol in the data block using a complex multiplier is as follows: Each constellation symbol is sequentially extracted from the data block. Each constellation symbol consists of two values: the first value is called the in-phase component, and the second value is called the quadrature component. The corresponding phase rotation angle values ​​are sequentially extracted from the phase rotation angle value sequence. These angle values ​​are zero, one, two, or three, corresponding to 0 degrees, 90 degrees, 180 degrees, and 270 degrees, respectively. The phase rotation angle values ​​are converted into coefficients required for the rotation operation. The specific conversion rules are: when the angle value is zero, the cosine coefficient is one and the sine coefficient is zero; when the angle value is one, the cosine coefficient is zero and the sine coefficient is one; when the angle value is two, the cosine coefficient is negative one and the sine coefficient is zero; when the angle value is three, the cosine coefficient is zero and the sine coefficient is negative one. The in-phase component of the constellation symbol is multiplied by the cosine coefficient to obtain the first product. The quadrature component of the constellation symbol is multiplied by the sine coefficient to obtain the second product. Subtracting the second product from the first product yields the rotated in-phase component. Multiplying the in-phase component of the constellation symbol by a sine coefficient yields the third product. Multiplying the quadrature component of the constellation symbol by a cosine coefficient yields the fourth product. Adding the third product to the fourth product yields the rotated quadrature component. Combining the new in-phase component and the new quadrature component forms the rotated constellation symbol. Performing the above rotation matrix multiplication operation on all constellation symbols, arranging all rotated constellation symbols in their original order, forms a new constellation symbol sequence, which is used as the jitter modulation frame of the data block.

[0040] The parameters in the formula for calculating the phase rotation angle are derived from the following sources: (The...) The phase rotation angle value of the constellation symbol is taken from the sequence of phase rotation angle values ​​of the phase step value. The number of phase quantization bits is a fixed positive integer preset by the system, which determines the quantization fineness of the phase rotation angle value. The first bit of the binary phase perturbation sequence... The output of the clock cycle 1 The value of the bit comes from the binary phase perturbation sequence of the linear feedback shift register, where Indicates the first An eight-bit sequence output in one clock cycle. This indicates the 8th bit in the octet sequence. Each bit position The value ranges from zero to one less than the number of phase quantization bits. The first... The state vector in the nth clock cycle refers to the state vector in the nth clock cycle. At the start of the first clock cycle, the vector consists of eight bits stored in the eight memory cells of the linear feedback shift register. The first binary phase perturbation sequence... The vector formed by subtracting one clock cycle from the output refers to the vector formed by arranging each bit sequentially in the eight-bit sequence shifted out in the previous clock cycle.

[0041] The significance of the phase rotation angle calculation formula lies in performing a bitwise AND operation between the current state vector of the linear feedback shift register and the output vector of the previous cycle, counting the number of bits with a value of one in the bitwise AND result to obtain a non-negative integer. This non-negative integer is used as a modulation factor and XORed with the weighted value of each bit in the binary phase perturbation sequence output in the current clock cycle. Then, the modulus is taken with the number of phase quantization bits, finally outputting an integer in the range of zero to the phase quantization modulus minus one. Multiplying this integer by the basic angle step size yields the phase rotation angle value. This formula ensures that the generation of the current phase rotation angle value depends on both the current register state and the historical output sequence, thereby generating a phase rotation amount in each clock cycle that is non-linearly correlated with both the preceding and following cycles, enhancing the randomness and detectability of phase jitter characteristics.

[0042] The trend of the phase rotation angle calculation formula is as follows: the more bits with a value of one in the bitwise AND result of the current state vector of the linear feedback shift register and the output vector of the previous cycle, the larger the modulation factor. The higher the probability of bit flipping in the XOR result of the modulation factor and the current output bit weighting value, the more uniform the distribution of the modulo operation result within its range. When the number of bits with a value of one in the bitwise AND result is zero, the modulation factor is zero. In this case, the formula degenerates into simply accumulating the current output bit weighting value and taking the modulus, and the phase rotation angle value exhibits a regular change. Therefore, this formula can adaptively adjust the distribution characteristics of the phase rotation angle value under different register state combinations, maintaining the statistical distinguishability of phase jitter characteristics in low signal-to-noise ratio environments.

[0043] The beneficial effects of this technical solution are that, in distributed communication scenarios with extremely low signal-to-noise ratios, by actively embedding time-domain amplitude concave windows and phase jitter features, and utilizing a nonlinear formula based on the state and historical output of a linear feedback shift register to generate phase rotation angle values, the transmitter can autonomously sense collision intensity and generate jitter modulation frames with high randomness and detectability. The raised cosine roll-off shaping of the amplitude concave window and the reduction of the multiplier amplitude ensure the reproducibility of the concave features, while the cumulative cyclic truncation of the phase jitter features and the rotation matrix multiplication ensure uniform distribution of phase disturbances and low-overhead retransmission compatibility. The entire solution achieves closed-loop backoff control without carrier sensing and channel estimation, effectively reducing retransmission power consumption and improving channel access efficiency and data delivery reliability.

[0044] S03. Transmit the concave modulation frame, analyze and extract the transmitted airborne signal to obtain the distortion characteristic parameters of the airborne signal.

[0045] In this embodiment of the invention, the step of transmitting the concave modulation frame and analyzing and extracting the transmitted airborne signal to obtain the distortion feature parameters of the airborne signal includes: The concave modulation frame is up-converted and pushed to obtain the radio frequency transmission waveform of the concave modulation frame; In the shared channel, the transmitting antenna is excited and radiated based on the radio frequency bearer waveform to obtain the air propagation signal of the concave modulation frame; The air-borne signal is down-converted and demodulated to obtain the baseband recovered waveform of the air-borne signal; Energy sampling is performed on the concave regions in the baseband recovery waveform to obtain the local power sequence of the baseband recovery waveform; The amplitude of the concave regions in the local power sequence is compared to obtain the degree of concave distortion of the air-propagating signal; The jitter distortion degree of the airborne signal is obtained by performing phase differential correlation on the local power sequence.

[0046] The specific implementation process for up-converting the concave modulation frame to obtain its radio frequency transmission waveform is as follows: The baseband sampling point sequence of the concave modulation frame is fed into an upconverter. The upconverter contains a local oscillator that generates a sinusoidal carrier signal at a fixed frequency. The value of each baseband sampling point is multiplied by the instantaneous amplitude of the sinusoidal carrier signal to obtain the modulated signal. All modulated signals are continuously output in time sequence, forming a high-frequency continuous waveform, which is used as the radio frequency transmission waveform of the concave modulation frame.

[0047] In a shared channel, the specific implementation process for exciting and radiating the transmitting antenna based on the radio frequency bearer waveform to obtain the airborne propagation signal of the concave modulation frame is as follows: The radio frequency transmission waveform is transmitted to the input port of the transmitting antenna through a feeder. The transmitting antenna converts the input electrical signal into an electromagnetic wave and radiates it into free space. The radiated electromagnetic wave propagates in the shared channel and reaches the receiving end after path attenuation and multipath effects. The electromagnetic wave induced by the receiving antenna is converted into an electrical signal, and this electrical signal is used as the airborne propagation signal of the concave modulation frame.

[0048] The specific implementation process for down-converting and demodulating the airborne signal to obtain the baseband recovered waveform is as follows: The airborne signal is fed into a downconverter. The downconverter contains a local oscillator that generates a sinusoidal carrier signal with the same frequency as the transmitter. The received airborne signal is multiplied by this sinusoidal carrier signal to obtain a mixed signal containing baseband and high-frequency components. This mixed signal is passed through a low-pass filter to remove the high-frequency components and retain the low-frequency baseband components. The filtered continuous waveform is synchronized with timing, and the sampling points within each symbol period are extracted to obtain a discrete baseband sampling point sequence. This sequence is used as the baseband recovered waveform of the airborne signal.

[0049] The specific implementation process for sampling energy in the recessed region of the baseband recovered waveform to obtain the local power sequence of the baseband recovered waveform is as follows: Based on the preset recessed window start mark and recessed window duration width at the transmitter, the sampling point interval corresponding to the recessed window in the baseband recovered waveform is determined. Within this interval, the complex value of each sampling point is read sequentially from the first sampling point to the last sampling point. The square of the magnitude of the complex value of each sampling point is calculated, i.e., the square of the in-phase component plus the square of the quadrature component, to obtain the instantaneous power of that sampling point. The instantaneous powers of all sampling points within the recessed window are arranged in the sampling order to form a power value sequence, which is used as the local power sequence of the baseband recovered waveform.

[0050] The specific implementation process for comparing the amplitude of the concave intervals in the local power sequence to obtain the degree of concave distortion of the airborne signal is as follows: Find the minimum value in the local power sequence, which corresponds to the power value at the bottom of the concave window. Use the original concave depth value preset by the transmitter as a reference benchmark; this original concave depth value is the theoretical power value at the bottom of the concave window under collision-free conditions. Calculate the deviation between the minimum value and the original concave depth value; specifically, subtract the minimum value from the original concave depth value to obtain the power difference. Normalize this power difference to the range of zero to one; the normalization method is to divide the power difference by the original concave depth value, and the resulting ratio is the degree of concave distortion. Use the finally calculated ratio as the degree of concave distortion of the airborne signal.

[0051] The specific implementation process for obtaining the jitter distortion degree of the air-borne signal by performing phase differential correlation on the local power sequence is as follows: Sampling points in the non-concave interval outside the concave window are extracted from the baseband recovered waveform. These sampling points carry phase jitter characteristics. The phase angle of each sampling point is calculated, and the in-phase and quadrature components are converted into phase angles using a coordinate rotation digital conversion method. The phase angles of two adjacent sampling points are subtracted to obtain the phase difference value. This operation is performed on all adjacent sampling points to obtain the measured phase difference sequence. A reference phase difference sequence corresponding to the phase jitter characteristics attached at the transmitter is pre-stored locally. The measured phase difference sequence is aligned with the reference phase difference sequence, and the two difference values ​​are multiplied at each corresponding position to obtain a product sequence. All values ​​in the product sequence are summed to obtain the correlation accumulation value. This correlation accumulation value is divided by the square root of the energy product of the two sequences to obtain the normalized correlation coefficient. The correlation coefficient is subtracted from one to obtain the jitter distortion degree. The finally calculated value is used as the jitter distortion degree of the air-borne signal.

[0052] The beneficial effects of this transmission and parsing extraction process are that, in distributed communication scenarios with extremely low signal-to-noise ratios, by actively transmitting modulated frames embedded with concave windows and self-monitoring the propagating signals in the air, the energy and phase difference features of the concave interval are accurately extracted from the baseband recovered waveform, achieving collision intensity sensing without carrier sensing and channel estimation. The separation and extraction of concave distortion and jitter distortion eliminates the influence of channel gain on the detection results, enabling the system to stably acquire collision feedback in environments where the signal is submerged in noise. This provides a reliable basis for subsequent closed-loop backoff decisions, effectively reducing retransmission power consumption and improving channel access efficiency and transmission reliability.

[0053] S04. Based on the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameters, the base time slot increment is mapped. The base time slot increment is scaled proportionally according to the priority label to obtain the time slot increment of the data block. The time slot increment is then superimposed with the current waiting time slot to obtain the waiting time slot of the data block to be sent.

[0054] In this embodiment of the invention, the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameters is mapped to the basic time slot increment. The basic time slot increment is scaled proportionally according to the priority label to obtain the time slot increment of the data block. The time slot increment is then superimposed with the current waiting time slot to obtain the waiting time slot of the data block to be sent. This includes: Based on a preset ratio threshold group, the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters are successively approximated and compared to obtain the distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion. Based on the distortion ratio interval index, the address of the read-only memory is decoded, and the pre-fixed time slot step value is extracted from the read-only memory to obtain the basic time slot increment of the data block to be sent. Based on the priority label, the basic time slot increment is proportionally weighted to obtain the time slot increment of the data block; The current waiting time slot is accumulated and merged with the time slot increment to obtain the waiting time slot of the data block to be sent.

[0055] The method involves successively approximating and comparing the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters based on a preset ratio threshold group, to obtain an index of the distortion ratio range between the degree of concave distortion and the degree of jitter distortion, including: The degree of concave distortion is compared with the degree of jitter distortion by a fixed multiple, and the multiple level at which the inequality first holds true is recorded to obtain the coarse adjustment ratio level of the degree of concave distortion to the degree of jitter distortion. Based on a preset ratio threshold group, the two adjacent boundary thresholds of the coarse adjustment ratio level are extracted to obtain the upper boundary threshold and lower boundary threshold of the coarse adjustment ratio level. The difference between the degree of concave distortion and the lower boundary threshold is amplified, and the amplified result is compared with the difference to obtain the fine-tuning offset level between the degree of concave distortion and the degree of jitter distortion. By bit-joining the coarse adjustment ratio level and the fine adjustment offset level, an index of the distortion ratio range between the degree of concave distortion and the degree of jitter distortion is obtained.

[0056] The specific implementation process for obtaining the basic time slot increment of the data block to be transmitted by decoding the address of the read-only memory (ROM) based on the distortion ratio interval index and extracting the pre-fixed time slot step size value from the ROM is as follows: The distortion ratio interval index is input as an address into the ROM. The ROM internally stores a mapping table, where each address corresponds to a pre-fixed time slot step size value. Address zero corresponds to a time slot step size of one, address one to two, address two to four, address three to eight, and address four to sixteen. The address decoder converts the index value into a physical address of the ROM and selects the corresponding storage unit. The binary value stored in that storage unit is read out and used as the basic time slot increment of the data block to be transmitted.

[0057] The specific implementation process for obtaining the time slot increment of the data block by proportionally weighting the basic time slot increment based on the priority tag is as follows: Obtain the priority tag, which is an integer value ranging from zero to two, corresponding to high priority, medium priority, and low priority, respectively. Pre-set three weighting coefficients: priority tag zero corresponds to a weighting coefficient of 0.5, priority tag one corresponds to a weighting coefficient of 1.0, and priority tag two corresponds to a weighting coefficient of 2.0. Input the basic time slot increment and the corresponding weighting coefficients into a multiplier, which performs real number multiplication and outputs the product. Round the product to the nearest integer value, and use this integer value as the time slot increment of the data block.

[0058] The specific implementation process for accumulating and merging the current waiting time slot and the time slot increment to obtain the waiting time slot of the data block to be sent is as follows: Read the currently stored waiting time slot value from the local register. Feed the time slot increment and the current waiting time slot value together into an adder, which performs integer addition and outputs the sum. Write this sum back to the local register, overwriting the original current waiting time slot value. Use the written register value as the waiting time slot of the data block to be sent.

[0059] The specific implementation process for comparing the degree of concave distortion with a fixed multiple of the degree of jitter distortion at progressively higher ratios, and recording the first multiple level that makes the inequality true, to obtain the coarse adjustment ratio level of the degree of concave distortion to the degree of jitter distortion, is as follows: The degree of jitter distortion is multiplied by a preset multiple sequence, which is 2, 4, 8, and 16. First, the result of multiplying the degree of jitter distortion by 2 is calculated and compared with the degree of concave distortion. If the degree of concave distortion is less than the product, the comparison stops, and the level of the current multiple is recorded as one. If not, the result of multiplying by 4 is calculated and compared. If the degree of concave distortion is less than the product, the level is recorded as two. This continues until multiplying by 16, where the result still does not make the inequality true, and the level is recorded as five. The final recorded level value is used as the coarse adjustment ratio level of the degree of concave distortion to the degree of jitter distortion.

[0060] The specific implementation process for extracting the upper and lower boundary thresholds of the coarse adjustment ratio level based on the preset ratio threshold group is as follows: The preset ratio threshold group contains five boundary thresholds, namely 0.5, 1.0, 2.0, 4.0, and 8.0, each boundary threshold corresponding to a level boundary. When the coarse adjustment ratio level is one, the corresponding lower boundary threshold is 0.0, and the upper boundary threshold is 0.5. When the coarse adjustment ratio level is two, the lower boundary threshold is 0.5, and the upper boundary threshold is 1.0. When the coarse adjustment ratio level is three, the lower boundary threshold is 1.0, and the upper boundary threshold is 2.0. When the coarse adjustment ratio level is four, the lower boundary threshold is 2.0, and the upper boundary threshold is 4.0. When the coarse adjustment ratio level is five, the lower boundary threshold is 4.0, and the upper boundary threshold is 8.0. Based on the coarse adjustment ratio level, the corresponding two boundary values ​​are extracted from this group, with the smaller value used as the lower boundary threshold and the larger value used as the upper boundary threshold.

[0061] The specific implementation process for amplifying the difference between the degree of concave distortion and the lower boundary threshold, and comparing the amplified result with the difference between the upper boundary threshold and the lower boundary threshold to obtain the fine-tuning offset level of the degree of concave distortion and the degree of jitter distortion is as follows: Subtract the lower boundary threshold from the degree of concave distortion to obtain a difference. Multiply this difference by a preset amplification factor, which is equal to two, four, or eight, with the specific value determined according to the coarse adjustment ratio level. Simultaneously calculate the difference between the upper boundary threshold and the lower boundary threshold to obtain an interval width. Compare the amplified difference with the interval width to determine the proportion of the amplified difference falling within the interval width. Divide this proportion into four equally spaced fine-tuning levels. If the amplified difference is less than one-quarter of the interval width, the fine-tuning offset level is zero; if it is between one-quarter and one-half, the level is one; if it is between one-half and three-quarters, the level is two; and if it is greater than three-quarters, the level is three. This level value is used as a fine-tuning offset level between the degree of concave distortion and the degree of jitter distortion.

[0062] The specific implementation process for combining the coarse adjustment ratio level and the fine adjustment offset level to obtain the distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion is as follows: The coarse adjustment ratio level is converted into a three-bit binary code, where level 1 corresponds to 001, level 2 to 010, level 3 to 011, level 4 to 100, and level 5 to 101. The fine adjustment offset level is converted into a two-bit binary code, where level 0 corresponds to 00, level 1 to 01, level 2 to 10, and level 3 to 11. The three-bit code of the coarse adjustment ratio level is used as the high-order bit, and the two-bit code of the fine adjustment offset level is used as the low-order bit, concatenating them into a five-bit binary number. This five-bit binary number is used as the final index value and output as the distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion.

[0063] The beneficial effect is that this mapping and scaling process uses two levels of approximation comparison—coarse and fine—to accurately quantify the ratio of concave distortion to jitter distortion into a distortion ratio interval index. The base time slot increment is then obtained by looking up a table in read-only memory and weighted proportionally by combining it with priority tags, ultimately accumulating and updating the waiting time slots. This mechanism achieves joint closed-loop backoff control of collision intensity and data priority, enabling the backoff step size to continuously and adaptively adjust according to the actual collision intensity. This avoids the channel fluctuation and priority scheduling failure problems of traditional discrete random backoff, thus significantly improving channel access efficiency and reducing the transmission delay and overall power consumption of high-priority data in distributed scenarios with extremely low signal-to-noise ratios.

[0064] S05. When the waiting time slot is completed, the jitter modulation frame is granted retransmission authorization and concatenated with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted.

[0065] In this embodiment of the invention, the step of granting retransmission authorization to the jitter modulation frame when the waiting time slot is completed, and concatenating and encapsulating it with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted, includes: The waiting time slot is monitored by a decreasing count to obtain the zero-trigger pulse of the waiting time slot; Based on the zero-trigger pulse, the transmit permission is set for the jitter modulation frame to obtain the repeat transmit authorization signal for the jitter modulation frame; The data sequence of the next data block is buffered and read to obtain the symbol stream to be concatenated for the next data block; Based on the repeated transmission authorization signal, the jitter modulation frame is joined end-to-end with the symbol stream to be spliced ​​to obtain the backoff modulation output frame of the data block to be transmitted.

[0066] The specific implementation process for monitoring the decrementing count of the waiting time slot to obtain the zero-trigger pulse for the waiting time slot is as follows: A decrementing counter is set up, and the value of the waiting time slot is loaded into the counter as its initial value. At each clock cycle, the counter decrements its current value by one and saves the result back to the counter. After each clock cycle decrement, the current value of the counter is checked to see if it equals zero. The moment the counter value decreases from one to zero, the detection circuit outputs a high-level pulse signal, which lasts for one clock cycle before returning to a low level. This high-level pulse signal is used as the zero-trigger pulse for the waiting time slot.

[0067] The specific implementation process for setting the transmit enable flag of the jitter-modulated frame based on the zero-reset trigger pulse to obtain the repeat transmit authorization signal of the jitter-modulated frame is as follows: The zero-reset trigger pulse is sent to a set trigger, the initial state of which is zero. When the high level of the zero-reset trigger pulse arrives, the trigger sets the output state to one and maintains this state until an external reset signal is received. The output signal of the trigger is used as a transmit enable flag, which is used to control the opening of subsequent transmit channels. This transmit enable flag signal is named the repeat transmit authorization signal of the jitter-modulated frame.

[0068] The specific implementation process for reading the data sequence of the next data block from the cache to obtain the symbol stream to be concatenated for the next data block is as follows: The system maintains a first-in-first-out (FIFO) cache queue to temporarily store the data sequence of the next data block that has undergone symbol periodic expansion. When concatenation is required, the value of each symbol is read sequentially from the head of the cache queue, and retrieved one by one according to the storage order to form a continuous symbol output sequence. The entire sequence of symbols read is used as the symbol stream to be concatenated for the next data block.

[0069] The specific implementation process of concatenating the jitter modulation frame with the symbol stream to be spliced ​​based on the repeat transmit grant signal to obtain the backoff modulation output frame of the data block to be transmitted is as follows: When the repeat transmit grant signal is valid, the splicer is started. The splicer first outputs all the sample points of the jitter modulation frame, one by one in the original order in the frame. After outputting the last sample point of the jitter modulation frame, it immediately outputs the first sample point of the symbol stream to be spliced, and then continues to output all subsequent sample points until the last sample point of the symbol stream to be spliced ​​is output. The splicer does not insert any additional sample points or padding symbols between the two sequences, and directly connects the end of the previous sequence with the beginning of the next sequence. The complete sample point sequence formed after splicing is used as the backoff modulation output frame of the data block to be transmitted.

[0070] The beneficial effects are that this process precisely triggers the zero-trigger pulse through decremental counting monitoring, ensuring that retransmission is authorized immediately after the waiting time slot is completed, avoiding conflicts between invalid waiting and premature transmission. The transmit-allow setting mechanism ensures that jitter-modulated frames are output only when authorized, guaranteeing the uniqueness and reliability of the retransmission operation. The symbol stream of the next data block is read from the buffer and seamlessly connected with the retransmission frame, merging the retransmission and the initial transmission into a single output frame, completely hiding the time slot overhead occupied by the retransmission. Thus, in distributed scenarios with extremely low signal-to-noise ratios, the system achieves a one-time combined transmission of retransmission and the initial transmission, significantly reducing the average power consumption per bit and improving channel utilization and data throughput.

[0071] like Figure 2 The diagram shown is a functional block diagram of a low-power modulation adaptive system based on data priority, provided in an embodiment of the present invention.

[0072] The low-power modulation adaptive system 100 based on data priority described in this invention can be installed in an electronic device. Depending on the functions implemented, the low-power modulation adaptive system 100 based on data priority may include a priority parsing module 101, a dual-feature embedding module 102, a channel collision sensing module 103, a backoff decision and threshold evolution module 104, and a retransmission merging and frame assembly module 105. The modules described in this invention can also be referred to as units, which are a series of computer program segments that can be executed by the processor of an electronic device and can perform a fixed function, stored in the memory of the electronic device.

[0073] In this embodiment, the functions of each module / unit are as follows: The priority parsing module 101 performs priority analysis and extraction on the data block to be sent to obtain the priority tag of the data block to be sent; The dual-feature embedding module 102 embeds a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and adds a time-domain phase jitter feature to the data block to obtain a jitter modulation frame of the data block. The channel collision sensing module 103 transmits the concave modulation frame, analyzes and extracts the transmitted airborne signal, and obtains the distortion feature parameters of the airborne signal. The backoff decision and threshold evolution module 104 is used to map the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameter to the basic time slot increment, scale the basic time slot increment according to the priority label to obtain the time slot increment of the data block, and superimpose the time slot increment with the current waiting time slot to obtain the waiting time slot of the data block to be sent. When the waiting time slot is completed, the retransmission merging and frame assembly module 105 grants retransmission authorization to the jitter modulation frame and splices and encapsulates it with the data block of the next data block to obtain the backoff modulation output frame of the data block to be sent.

[0074] In the several embodiments provided by this invention, it should be understood that the disclosed methods and systems can be implemented in other ways. For example, the system embodiments described above are merely illustrative; for instance, the division of modules is only a logical functional division, and other division methods may be used in actual implementation.

[0075] The modules described as separate components may or may not be physically separate. The components shown as modules may or may not be physical units; that is, they may be located in one place or distributed across multiple network units. Some or all of the modules can be selected to achieve the purpose of this embodiment according to actual needs.

[0076] Furthermore, the functional modules in the various embodiments of the present invention can be integrated into one processing unit, or each unit can exist physically separately, or two or more units can be integrated into one unit. The integrated unit can be implemented in hardware or in the form of hardware plus software functional modules.

[0077] It will be apparent to those skilled in the art that the present invention is not limited to the details of the exemplary embodiments described above, and that the present invention can be implemented in other specific forms without departing from the spirit or essential characteristics of the present invention.

[0078] This application embodiment describes a method for acquiring and processing relevant data based on wireless communication technology. Wireless communication utilizes digital computers or machines controlled by digital computers to simulate, extend, and expand human intelligence, perceive the environment, acquire knowledge, and use that knowledge to obtain optimal results—a theory, method, technology, and application system.

[0079] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit it. Although the present invention has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can be made to the technical solutions of the present invention without departing from the spirit and scope of the technical solutions of the present invention.

Claims

1. A low-power modulation adaptive method based on data priority, characterized in that, The method includes: S01. Perform priority analysis and extraction on the data block to be sent to obtain the priority label of the data block to be sent; S02. Embed a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and add time-domain phase jitter features to the data block to obtain a jitter modulation frame of the data block. S03. Transmit the concave modulation frame, analyze and extract the transmitted airborne signal to obtain the distortion characteristic parameters of the airborne signal; S04. Based on the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameter, the base time slot increment is mapped. The base time slot increment is scaled proportionally according to the priority label to obtain the time slot increment of the data block. The time slot increment is then superimposed with the current waiting time slot to obtain the waiting time slot of the data block to be sent. S05. When the waiting time slot is completed, the jitter modulation frame is granted retransmission authorization and concatenated with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted.

2. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, The step of performing priority analysis and extraction on the data block to be sent to obtain the priority tag of the data block to be sent includes: The service type is parsed from the header field of the data block to be sent to obtain the service level indication of the data block to be sent. The redundancy ratio of the data block to be sent is obtained by performing a duplicate count on the payload of the data block to be sent. The remaining lifetime of the embedded timestamp of the data block to be sent is determined to obtain the urgency level of the data block to be sent; Based on the service level indication, information redundancy ratio, and urgency level, a comprehensive evaluation is performed on the service level indication, information redundancy ratio, and urgency level to obtain the priority label of the data block to be sent.

3. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, The step of embedding a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block includes: The data block is subjected to an energy envelope scan. The obtained envelope trough position is marked as the indentation window start mark, and the obtained trough duration length is marked as the indentation window duration width.

4. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, The step of adding time-domain phase jitter features to the data block to obtain the jitter-modulated frame of the data block includes: The linear feedback shift register of the data block is subjected to tap XOR feedback, and the resulting shift register is injected into bits and shifted out bit by bit to obtain the binary phase perturbation sequence of the linear feedback shift register; The phase step size of the binary phase perturbation sequence is obtained by performing a Gray mapping on the binary phase perturbation sequence. Based on a preset phase quantization modulus, the phase step value is accumulated and then truncated cyclically to obtain the phase rotation angle value of the phase step value. Based on the phase rotation angle value, a rotation matrix multiplication is performed on the in-phase and quadrature components of each constellation symbol of the data block using a complex multiplier to obtain the jitter modulation frame of the data block.

5. The low-power modulation adaptive method based on data priority as described in claim 4, characterized in that, The phase rotation angle value is calculated using the following formula: ； In the formula, For the first The phase rotation angle value of each constellation symbol, The number of phase quantization bits for the phase rotation angle value. The first of the binary phase perturbation sequence The output of the clock cycle 1 The value of the bit, For bitwise XOR operation, This is a function to count the number of 1s in a binary vector. For the first linear feedback shift register A state vector for each clock cycle. The first of the binary phase perturbation sequence A vector composed of outputs over clock cycles. This is a bitwise AND operation.

6. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, The process involves transmitting and outputting the concave modulation frame, and then analyzing and extracting the transmitted airborne signal to obtain distortion characteristic parameters of the airborne signal, including: The concave modulation frame is up-converted and pushed to obtain the radio frequency transmission waveform of the concave modulation frame; In the shared channel, the transmitting antenna is excited and radiated based on the radio frequency bearer waveform to obtain the air propagation signal of the concave modulation frame; The air-borne signal is down-converted and demodulated to obtain the baseband recovered waveform of the air-borne signal; Energy sampling is performed on the concave regions in the baseband recovery waveform to obtain the local power sequence of the baseband recovery waveform; The amplitude of the concave regions in the local power sequence is compared to obtain the degree of concave distortion of the air-propagating signal; The jitter distortion degree of the airborne signal is obtained by performing phase differential correlation on the local power sequence.

7. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, The ratio of concave distortion to jitter distortion in the distortion feature parameters is mapped to a base time slot increment. This base time slot increment is scaled proportionally according to the priority label to obtain the time slot increment of the data block. This time slot increment is then superimposed with the current waiting time slot to obtain the waiting time slot for the data block to be sent, including: Based on a preset ratio threshold group, the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters are successively approximated and compared to obtain the distortion ratio interval index of the degree of concave distortion and the degree of jitter distortion. Based on the distortion ratio interval index, the address of the read-only memory is decoded, and the pre-fixed time slot step value is extracted from the read-only memory to obtain the basic time slot increment of the data block to be sent. Based on the priority label, the basic time slot increment is proportionally weighted to obtain the time slot increment of the data block; The current waiting time slot is accumulated and merged with the time slot increment to obtain the waiting time slot of the data block to be sent.

8. The low-power modulation adaptive method based on data priority as described in claim 7, characterized in that, The method involves successively approximating and comparing the degree of concave distortion and the degree of jitter distortion in the distortion feature parameters based on a preset ratio threshold group, to obtain an index of the distortion ratio range between the degree of concave distortion and the degree of jitter distortion, including: The degree of concave distortion is compared with the degree of jitter distortion by a fixed multiple, and the multiple level at which the inequality first holds true is recorded to obtain the coarse adjustment ratio level of the degree of concave distortion to the degree of jitter distortion. Based on a preset ratio threshold group, the two adjacent boundary thresholds of the coarse adjustment ratio level are extracted to obtain the upper boundary threshold and lower boundary threshold of the coarse adjustment ratio level. The difference between the degree of concave distortion and the lower boundary threshold is amplified, and the amplified result is compared with the difference to obtain the fine-tuning offset level between the degree of concave distortion and the degree of jitter distortion. By bit-joining the coarse adjustment ratio level and the fine adjustment offset level, an index of the distortion ratio range between the degree of concave distortion and the degree of jitter distortion is obtained.

9. The low-power modulation adaptive method based on data priority as described in claim 1, characterized in that, When the waiting time slot is completed, the jitter modulation frame is granted retransmission authorization and concatenated with the data block of the next data block to obtain the backoff modulation output frame of the data block to be transmitted, including: The waiting time slot is monitored by a decreasing count to obtain the zero-trigger pulse of the waiting time slot; Based on the zero-trigger pulse, the transmit permission is set for the jitter modulation frame to obtain the repeat transmit authorization signal for the jitter modulation frame; The data sequence of the next data block is buffered and read to obtain the symbol stream to be concatenated for the next data block; Based on the repeated transmission authorization signal, the jitter modulation frame is joined end-to-end with the symbol stream to be spliced ​​to obtain the backoff modulation output frame of the data block to be transmitted.

10. A low-power modulation adaptive system based on data priority, characterized in that, For implementing the low-power modulation adaptive method based on data priority as described in claim 1, the system comprises: The priority parsing module is used to perform priority analysis and extraction on the data block to be sent, and obtain the priority label of the data block to be sent; A dual-feature embedding module is used to embed a time-domain amplitude notch window into the data block to obtain a notch modulation frame of the data block, and to add time-domain phase jitter features to the data block to obtain a jitter modulation frame of the data block. The channel collision sensing module is used to transmit the concave modulation frame, analyze and extract the transmitted air propagation signal, and obtain the distortion feature parameters of the air propagation signal. The backoff decision and threshold evolution module is used to map the ratio of the degree of concave distortion to the degree of jitter distortion in the distortion feature parameters to the basic time slot increment, scale the basic time slot increment according to the priority label to obtain the time slot increment of the data block, and superimpose the time slot increment with the current waiting time slot to obtain the waiting time slot of the data block to be sent. The retransmission merging and frame assembly module is used to grant retransmission authorization to the jitter modulation frame when the waiting time slot is completed, and to splice and encapsulate it with the data block of the next data block to obtain the backoff modulation output frame of the data block to be sent.

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