An absorption terahertz high-speed on-off switch modulator and method

Abstract

This invention discloses an absorption-type terahertz high-speed switching modulator and method, relating to the field of terahertz radio frequency switching technology, and solves the problems of low operating frequency, low switching speed, and drastic port impedance changes in existing radio frequency switches. The invention includes a coupled-line directional coupler, with its through-end and coupled-end connected to a first impedance switching network and a second impedance switching network of identical structure, respectively. One end of the first impedance switching network and one end of the second impedance switching network are connected to a bias circuit via the coupled-line directional coupler, respectively. The other ends of the first and second impedance switching networks are connected to a baseband circuit, respectively. The input end of the coupled-line directional coupler is connected to a radio frequency input, and the isolation end of the coupled-line directional coupler is connected to a radio frequency output. This invention solves the problem of port impedance jumps during switch state switching.

Description

Technical Field

[0001] This invention relates to the field of terahertz radio frequency switching technology, and more specifically to an absorption-type terahertz high-speed switching modulator and method. Background Technology

[0002] Single-pole single-throw (SPST) solid-state RF switches are widely used in RF signal control, OOK modulators, and other fields. They typically employ high-speed RF technologies such as PIN diodes, HBTs, SOI CMOS, and HEMT transistors, and use series, parallel, and series-parallel topologies to control signal transmission and blocking by adjusting port impedance. In addition to the transistor-based topology, RF switches often incorporate appropriate matching structures to reduce bandwidth loss caused by transistor parasitic capacitance.

[0003] The main problems with current solid-state RF switches are: 1. Their operating frequency bands are generally low, making it difficult to support ultra-high-speed Gbps communication rates; 2. Most of their structures are used for low-frequency switching, lacking circuit design optimization for high-frequency switching modulation; 3. Their input impedance will change drastically with switching, especially when turned off, it will reflect the signal to the front-end circuit, affecting the performance of the front-end circuit. Summary of the Invention

[0004] To address the problems existing in the prior art, the present invention provides an absorption-type terahertz high-speed switching modulator and method, which solves the problems of low operating frequency, low switching speed and drastic changes in port impedance of existing RF switches.

[0005] An absorptive terahertz high-speed switching modulator includes: a coupled-line directional coupler, an impedance switching network, a bias circuit, and a baseband circuit; the through end and the coupled end of the coupled-line directional coupler are respectively connected to a first impedance switching network and a second impedance switching network with identical structures; one end of the first impedance switching network and one end of the second impedance switching network are respectively connected to the bias circuit through the coupled-line directional coupler; the other ends of the first impedance switching network and the second impedance switching network are respectively connected to the baseband circuit; the input end of the coupled-line directional coupler is connected to the radio frequency input; and the isolation end of the coupled-line directional coupler is connected to the radio frequency output.

[0006] Furthermore, the first impedance switching network is a port network, and its port input impedance can be switched between different impedances.

[0007] Furthermore, the first impedance switching network is a transistor, the base of which is connected to the through terminal of the directional coupler. The transistor has the following characteristics: its switching impedance can be adjusted by adjusting its size and control voltage; its parasitic capacitance does not affect the high reflection coefficient of the port; and it results in small parasitic capacitance.

[0008] Furthermore, as the size of the transistor increases, both the on and off resistances decrease while the on and off capacitances increase; as the on and off voltages increase, the on and off resistances also decrease while the off capacitance decreases slightly. By controlling these two factors together, the required transistor equivalent parameters can be obtained.

[0009] Furthermore, the directional coupler structure is an asymmetric coupling line, and the target optimization parameters are obtained by modeling and analyzing using asymmetric coupling line theory.

[0010] Furthermore, the baseband circuit includes a first transmission line, an open stub, a second transmission line, and a GSG pin connected in sequence. One end of the first transmission line away from the open stub is connected to the other end of the first impedance switching network and the other end of the second impedance switching network.

[0011] A high-speed terahertz switching modulation method using absorption involves the following steps: When the switch is open, the radio frequency (RF) signals to the through and coupled ends of a coupled-line directional coupler are completely reflected. The signals reflected from the through and coupled ends undergo a 90-degree phase shift, resulting in opposite phases at the input end, leading to further reflection and complete transmission through the isolation end. When the switch is closed, the output end of the switch can be considered the isolation end of the directional coupler. The input RF signal is completely dissipated by the matched impedance of the through and coupled ends, achieving high isolation and low reflection. The 90-degree phase shift is achieved by the transmission characteristics of the coupled-line directional coupler, ensuring a 90-degree phase difference between the outgoing signals from the through and coupled ends after a signal is input from the input end.

[0012] The beneficial effects of this invention include:

[0013] 1. To address the issue that existing terahertz reflective switches are not conducive to OOK modulation, a scheme using an absorptive switch modulation is proposed. The design of the absorptive switch is completed using a coupling line as the core structure. This solves the problem of port impedance switching during switch state transitions.

[0014] 2. To address the issue that traditional symmetrical coupling lines in coupled-line switches cannot achieve matching within dimensional requirements, an asymmetrical coupling line is proposed to complete the matching. This reduces the difficulty of achieving the required coupling lines under different manufacturing processes while maintaining the advantage of a larger coupling line bandwidth.

[0015] 3. To address the issue that traditional switches do not optimize the switching rate, a method is proposed to match the base parasitic parameters of the transistor to increase the baseband bandwidth and prevent reflection, thereby optimizing the theoretical modulation rate to 40Gbps. Attached Figure Description

[0016] Figure 1This is a schematic diagram of the structure of an absorption-type terahertz high-speed switching modulator according to an embodiment of this application.

[0017] Figure 2 This is a schematic diagram of an absorption-type terahertz high-speed switching modulator according to an embodiment of this application.

[0018] Figure 3 This is a schematic diagram illustrating port impedance optimization in an embodiment of this application.

[0019] Figure 4 This refers to the reflection coefficient of the impedance switching network involved in the embodiments of this application.

[0020] Figure 5 This is a schematic diagram of a passive transistor small-signal model according to an embodiment of this application.

[0021] Figure 6 This is a scan image of the resistance and capacitance values ​​at the center frequency, representing the square of the amplitude of the transistor reflection coefficient involved in the embodiments of this application.

[0022] Figure 7 This is a schematic diagram illustrating the matching principle of the baseband circuit using distributed transmission line modeling, as described in an embodiment of this application.

[0023] Figure 8 This is a schematic diagram of the baseband circuit involved in an embodiment of this application.

[0024] Figure 9 This is a circuit topology diagram of an absorption-type terahertz high-speed switching modulator according to an embodiment of this application.

[0025] Figure 10 This is a simulation diagram of an absorption-type terahertz high-speed switching modulator according to an embodiment of this application. Detailed Implementation

[0026] To make the objectives, technical solutions, and advantages of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of this application, and not all of the embodiments. Therefore, the following detailed description of the embodiments of this application provided in the accompanying drawings is not intended to limit the scope of the claimed application, but merely represents selected embodiments of this application. All other embodiments obtained by those skilled in the art based on the embodiments of this application without creative effort are within the scope of protection of this application.

[0027] Example 1

[0028] An absorption-type terahertz high-speed switching modulator, such as Figure 1As shown, it includes: a coupled-line directional coupler, a first impedance switching network, a second impedance switching network, a bias circuit, and a baseband circuit; the through end and the coupled end of the coupled-line directional coupler are respectively connected to the first impedance switching network and the second impedance switching network with the same structure; one end of the first impedance switching network and one end of the second impedance switching network are respectively connected to the bias circuit through the coupled-line directional coupler; the other ends of the first impedance switching network and the second impedance switching network are respectively connected to the baseband circuit; the input end of the coupled-line directional coupler is connected to the RF input end; and the isolation end of the coupled-line directional coupler is connected to the RF output end.

[0029] The function of the coupling line directional coupler is the same as that of the 90-degree directional coupler.

[0030] like Figure 2 As shown, the switch consists of a coupled-line directional coupler and two identical impedance switching networks. The two impedance switching networks are placed at the through and coupled ends of the coupled line. The specific design principle is as follows:

[0031] For a lossless, symmetric, and reciprocal ideal 90-degree directional coupler, the S-matrix can be written in the following four-port form:

[0032]

[0033] In the above formula and Defined as the pass-through and coupling coefficients of this directional coupler. Pass-through coefficient Characterizing the pass-through performance of a coupler, it is numerically equal to the ratio of the pass-through end of the coupled line to the input end when the ports are matched. Parameters, and coupling coefficients Characterizing the degree of coupling of a coupler, it is numerically equal to the coupling degree between the coupled end and the input end of the coupling line when the ports are matched. parameter.

[0034] An impedance switching network is a one-port network whose port input impedance can be switched between different impedances, assuming its current impedance is a fixed value. When the port reference impedance is When the reflection coefficient is viewed from the coupled line port towards the impedance switching network, it can be defined as: As shown in the following formula:

[0035]

[0036] By cascading the port impedance with the coupled-line network, and based on multi-port network analysis, the original four-port network is transformed into the following two-port network:

[0037]

[0038] Assuming the directional coupler has a coupling degree of 3dB, that is... When the expression is transformed, it takes the following form:

[0039]

[0040] Observing this S-parameter reveals the reflection coefficients of port 1 and port 2. , The transmission coefficient that is always 0 and from the input to the isolation terminal Only depends on In other words, regardless of how the input impedance of the impedance switching network changes, the input and output impedances of the switch always remain matched, and the transmission coefficient of the circuit depends on the reflection coefficient of the impedance switching network. The imaginary unit in the equation only causes a 90-degree phase delay and does not affect the transmission power.

[0041] In this way, when the switch is turned on, the reflection coefficient of the impedance switching network is reduced. An amplitude close to 1 is sufficient to guarantee the transmission coefficient of the switch. The amplitude is also close to 1, meaning the switch insertion loss is small; at the same time, when the switch is closed, the reflection coefficient of the port is close to 0, which makes the transmission coefficient amplitude also close to 0, meaning the isolation of the switch is high; when the impedance switching network changes, the overall reflection coefficient of the device remains at 0, meaning the switch has good impedance matching in both closed and open states.

[0042] Intuitively, when the coupling line switch is open, the RF signal to the through and coupled ends is completely reflected. The signal reflected from the through and coupled ends undergoes a 90-degree phase shift and then has an opposite phase at the input end, resulting in complete reflection and transmission through the isolation end. When the switch is closed, the output end of the switch can be regarded as the isolation end of the directional coupler. The input RF signal is completely consumed by the matching impedance of the through and coupled ends, so that the switch achieves high isolation and low reflection.

[0043] The transmission characteristics of the coupled-line switch derived above have a perfect match, which means that in the initial design, device matching and insertion loss and isolation optimization can be separated, and the focus can be placed on designing and optimizing the impedance switching network.

[0044] In another embodiment, to more intuitively represent the input impedance requirements of the impedance switching circuit, an impedance circle diagram is used to represent the impedance position, such as... Figure 3 As shown in the figure, the input impedance of the impedance switching network at different operating frequencies is plotted on this Smith chart. The goal is that when the switch is on, the impedance curve should be in the blue area, and the bluer the better, while when the switch is off, the input impedance curve should be in the orange area, and the orange the better.

[0045] According to this standard, this embodiment directly selects a specially sized grounded transistor as the impedance switching network. An impedance switching network is defined as a single-port network that can switch between different input impedances. Based on the previous derivation of the principle, it is known that it needs to ensure its input impedance meets certain requirements at a high bandwidth. Figure 4 Generally speaking, it is very difficult for a switching network (or reconfigurable circuit network) to meet the requirement of consistent input impedance across a wide frequency band under different states.

[0046] The specific reasons are threefold:

[0047] Firstly, the switching impedance of the transistor itself can be adjusted by modifying its size and control voltage. The main purpose of this is to adjust the transistor's on-resistance to match the port's reference impedance, thereby achieving high isolation during switching.

[0048] Secondly, the parasitic capacitance of the transistor itself does not affect the high reflection coefficient of the port. As mentioned earlier, the reason why traditional parallel switches require inductance matching is to shift the port impedance to a purely resistive line. However, coupled-line switches only require the transistor to produce a large port reflection coefficient, and do not require the impedance to be purely resistive.

[0049] Thirdly, the parasitic effect caused by a single transistor is small. Traditional switches utilize small on-resistance to reflect signals, while coupled-line switches require an on-resistance of 50Ω. Left and right matching impedances (for different coupled-line parameters, the matching impedance is determined by both odd and even-mode impedances). This means that coupled-line switches can use smaller transistors to meet the designed on-resistance requirements, and also means that parasitic capacitances are further reduced, especially in terms of their impact on the circuit in the on-state.

[0050] The following describes transistor selection methods: In RF switching applications, passive transistors (with zero power supply bias) are often used as the core switching device. For example... Figure 5 As shown, ignoring the control terminals, the source and drain terminals (collector and emitter) of the transistor are simplified and equivalent to resistors. With capacitor Parallel connection of resistors. The resistance at which a transistor is turned on is called the turn-on resistance. When closed, the resistor is turned off. Parasitic capacitance The capacitance at which the transistor is turned on is denoted as the turn-on capacitance. When closed, it is recorded as the closed capacitor. Generally speaking, the turn-off resistance of a transistor is much greater than its on-resistance, while the turn-off capacitor is slightly larger than the turn-on capacitor.

[0051] These parameters are affected by both transistor size and switching voltage: as size increases, both on- and off-resistance decrease while both on- and off-capacitance increase; as switching voltage increases, both on- and off-resistance decrease while off-capacitance decreases slightly. By controlling these two factors together, the equivalent transistor parameters suitable for this circuit topology can be combined, thereby improving switching performance.

[0052] To visually represent the range of transistor selection, the formula for the parallel input impedance of the transistor under this equivalent model is listed below:

[0053]

[0054] Substituting the input impedance into the previously mentioned reflection coefficient formula and calculating its amplitude, we can obtain the following after calculation and simplification:

[0055]

[0056] like Figure 6 This allows us to plot the square of the reflection coefficient amplitude and scan the resistance and capacitance values ​​at the center frequency, facilitating adjustments to the transistor size and control voltage. The uppermost image shows the transistor when it's on, revealing the resistance value at this time. The closer to 50 ohms and the conducting capacitance The smaller the value, the smaller the reflection coefficient. Furthermore, as the on-capacitance increases, the on-resistance should decrease slightly to ensure it lies within the valley of the contour image. The lower image shows the transistor when it is off, where the off-resistance can be seen. Larger and closed capacitor The larger the value, the greater the amplitude of the reflection coefficient.

[0057] The optimized transistor resistor adjustment circuit in this embodiment exhibits port reflection coefficient simulation results in the 120GHz to 160GHz frequency band, as shown in Figure 4. To compensate for the degradation of switch isolation caused by the on-capacitance, a 45-ohm on-resistance was selected, and the off-resistance was maximized by reducing the off-voltage. Statistically, throughout the scanning range, this impedance switching circuit maintains a minimum reflection coefficient of 0.89 when the switch is on and a maximum reflection coefficient of 0.08 when it is off. This ensures low insertion loss and high isolation of the switch across the wide bandwidth.

[0058] In another embodiment, due to limitations in process dimensions, the port characteristics of symmetrical coupling lines are not good, so an asymmetrical coupling line is used in this embodiment.

[0059] The specific limitations are quite complex and require analysis based on the principles of coupled lines: the transmission characteristics of symmetrical coupled lines are mainly determined by odd-mode impedance. Even-mode impedance This is certain, and these two impedances are related to the unit self-capacitance inductance and the unit mutual capacitance inductance of an ideal conductor structure, respectively. Generally, to ensure matching of the directional coupler, its matching impedance needs to be adjusted. up to 50 ,Right now At the same time, in order to ensure a coupling degree of 3dB, the square of its coupling degree C needs to be equal to 0.5, that is... Combining these two formulas, we can obtain the following requirements for the odd-mode impedance of the coupled line: This strict requirement necessitates fine-tuning the structure of the coupling lines.

[0060] The distributed capacitance and inductance of a conductor are affected by the minimum conductor size and minimum conductor spacing. When the coupler is symmetrical, the capacitance and inductance on both sides also appear in pairs. This means that the odd-mode impedance cannot be continuously adjusted by changing the conductor size. In this case, the structure can be changed to an asymmetrical coupled line to optimize the odd-mode impedance to the required value.

[0061] The analysis at this point cannot be performed using the same simple odd-even mode analysis as with ordinary symmetrical coupled lines, because the structural asymmetry leads to mode asymmetry. The port excitation can no longer generate a signal of equal amplitude in the circuit. The following is an ideal analysis of the theory of asymmetrical coupled lines:

[0062] For a coupled line of arbitrary transverse dimensions, a universal coupled line equation can be derived from the transmission line equation:

[0063]

[0064]

[0065]

[0066]

[0067] in , , , These represent the impedance and admittance per unit length of the two transmission lines (labeled 1 and 2), respectively. , This represents the mutual impedance and admittance per unit length between two transmission lines. It should be noted that the time factor has been omitted from the above formula. Therefore, the derivation result only applies to the case of sinusoidal steady state.

[0068] Simplifying the above formulas and canceling out the current, we obtain a system of second-order differential equations concerning the voltages on the two coupled lines:

[0069]

[0070]

[0071] Among them are coefficients , , , Written as:

[0072]

[0073]

[0074]

[0075]

[0076] Observation reveals that when the coupling line size parameters are determined, all coefficients in the equation system are related to the position on the line. Regardless, this system of equations is a second-order system of differential equations with constant coefficients. Assume the voltage function on the line has... The voltage waves of the same mode on both lines have the same propagation constant. Then we can obtain the eigenpropagation modes of this system of equations: Model and Rewritten as the following formula:

[0077]

[0078] The above formula yields four solutions:

[0079]

[0080]

[0081] in, Indicating different directions of propagation mold, Indicating different directions of propagation Model. Therefore. Model and The mode constitutes the eigenmodes on any coupled line. Just like the even and odd modes in a symmetric structure, any mode passing through the line can be decomposed and analyzed using these two modes.

[0082] By simplifying, the relationship between the voltages at the same position on the two lines can also be obtained:

[0083]

[0084] for In terms of models, the relationship between the two is as follows:

[0085]

[0086] for In terms of models, the relationship between the two is as follows:

[0087]

[0088] As can be observed from the above formula, when the coupling line is lossless, the coefficient , , , All are negative real numbers, therefore Both are real numbers, and if one is positive, the other must be negative. Specifically, when the structure of the coupled line is symmetrical, ... = , = We can get the current value. and This actually corresponds to the even-mode propagation constant in symmetric coupled-line analysis. With odd mode propagation constant .

[0089] Assuming the coupled line is in an infinite homogeneous medium, then based on the properties of the TEM wave on the transmission line:

[0090]

[0091]

[0092] Substituting the above conditions into the calculation, the propagation constant under uniform lossless medium conditions can be obtained. Relationship with mode voltage The expression is as follows:

[0093]

[0094]

[0095] In the formula and .

[0096] Simultaneously, by combining the mode voltage relationship with the above equation, the elements of the Z matrix of the asymmetric coupling line can be solved as follows:

[0097]

[0098]

[0099]

[0100]

[0101]

[0102]

[0103] In the above formula This represents the electric length, while in the above formula... Written as:

[0104]

[0105]

[0106] Comparing the direct coupled lines in even-odd mode analysis, we can see that the elements of the Z-matrix of asymmetric and symmetric coupled lines are essentially the same. Let the even-mode and odd-mode impedances of one line of the coupled line be denoted as... and Let the even-mode and odd-mode impedances of the other line be denoted as... and Then we can obtain the equivalent conditions for asymmetric coupled lines and symmetric coupled lines:

[0107]

[0108]

[0109]

[0110] Therefore, it can be concluded that the difference between asymmetric and symmetric coupled lines lies in the inconsistent odd-mode impedances of the two lines in an asymmetric coupled line, while the odd-mode impedances of the two lines in a symmetric coupled line are consistent. This inconsistent odd-mode impedance does not affect the coupling form of the coupled line. In fact, an asymmetric coupled line can be regarded as a symmetric coupled line that performs impedance transformation on the ports connecting the two lines. Through simulation optimization and appropriate adjustments, parameters such as the width and spacing of the metal layers can completely transform the asymmetric coupled line into a 90-degree directional coupler network.

[0111] like Figure 9 In this embodiment, a single conductor With two conductors This forms an asymmetric coupling line. Although this coupling line appears symmetrical along the central axis, it cannot be analyzed using odd-even modes because the excitation modes on the two lines are not of equal amplitude. Therefore, the asymmetric coupling line theory described above is still used for modeling and analysis. Finally, after optimizing parameters such as the spacing and width of the three conductors, the equivalent odd-mode and even-mode impedances of the structure at the center of the frequency band of 140GHz were measured to be... and This makes its matching resistor 50 Around 80GHz.

[0112] The purpose of this optimization is to adjust the parameters of the three-segment coupled line to be consistent with those of a normal symmetrical coupled line (i.e., to use a three-segment asymmetrical coupled line to equivalently create a symmetrical coupled line with odd-mode and even-mode impedances of 25Ω and 115Ω, respectively), so that it can be used in the overall switching modulator. It should be emphasized that the derivation of the equivalence conditions between the asymmetrical and symmetrical coupled lines is performed in this embodiment.

[0113] In another embodiment, in conventional RF switch designs, since the signal rate at the control port is typically below GHz, the leads are often directly connected to the base of the transistor. However, in the design of high-speed switches and OOK modulators, the overall bandwidth and matching of the circuit from the GSG input port to the transistor base directly affect the integrity of the baseband signal; therefore, optimizing the bandwidth of this part of the baseband circuit is essential.

[0114] In this embodiment, the schematic diagram of the baseband circuit matching principle using distributed transmission line modeling is as follows: Figure 7 As shown. The overall signal path is from the digital signal source through the transmission line to the transistor. Due to the low input impedance of the transistor at low frequencies... Generally, signals with large magnitudes will mostly reflect towards the signal source at the base. If the impedance continuity of the intermediate transmission line is not guaranteed, the signal will be reflected multiple times along the line, affecting the signal bandwidth and causing a "ringing" phenomenon, which will reduce the modulation depth and speed of the RF modulation signal.

[0115] The schematic diagram of the baseband circuit used in this embodiment is shown below. Figure 8 As shown, its structure mainly consists of Pins, transmission lines , Path-opening branches With transmission line Composition. Theoretically, the largest parasitic parameter in a transmission structure comes from... Since the pins are used for wire bonding or probe placement, their size must not be too small, resulting in significant parasitic capacitance between the signal terminals and the ground plane. To mitigate the effects of this capacitance, firstly... Shift the input impedance from the lower half-capacitive plane to the upper half-inductive plane of the Smith chart, and then through... Pull the impedance of the inductor plane back onto the real axis of the chart, and ensure that Impedance matching is achieved when the characteristic impedance matches the real input impedance. Simplifying this to a lumped circuit, it can be considered... When the wire length is short and the wire width is thin, it is an inductor, while an open circuit branch This can also be simplified to a capacitor. From this perspective, the parasitic capacitance from the pin... Equivalent inductance and The equivalent capacitance forms a segment The matching circuit is a type that ensures that the low-pass transmission characteristics of the baseband circuit are not compromised over a large bandwidth, and "absorbs" parasitic capacitance to ensure impedance matching of the overall circuit.

[0116] In another embodiment, the circuit schematic of an absorptive terahertz high-speed switching modulator is shown in the attached figure. Figure 9 As shown, it includes the following five parts:

[0117] and The two transistors correspond to the first impedance switching network and the second impedance switching network. and The resulting coupling lines correspond to a directional coupler. , and The bias circuit formed , , and The baseband circuit consists of the first transmission line, the open stub, the second transmission line, and the GSG pin. , and , Input-output isolated DC The pin circuits correspond to the RF input and RF output terminals.

[0118] The collectors of T1 and T0 are connected to one end of conductors TL1 and TL0, respectively; the other ends of TL1 and TL0 are connected to wires TL2 and TL3, which are quarter-wavelength wires, respectively. These two wires are connected to the 0-ohm line L0 at VCC level to provide bias for the transistors; at the same time, TL1 and TL0 are connected to the RF output and input GSG pins through capacitors Co and Ci, respectively. The capacitors here are used to provide DC isolation; the bases of T1 and T0 are connected to the baseband GSG pins through TL4, TL5 and TL6.

[0119] The coupler uses a coupled line. This reduces the parasitic attenuation of lumped elements such as inductors at high frequencies, allowing its operating frequency to be increased to the terahertz band; at the same time, the transmission form of the coupled line allows it to transmit terahertz waves with less loss when the switch is open, while when the switch is closed, the coupled line forms a directional coupler, allowing the terahertz waves on the line to be dissipated and absorbed by the matched load presented by the transistor at the through end;

[0120] and This forms a three-wire asymmetric coupled line, and its equivalent odd-mode and even-mode impedances are measured to be... and This makes its matching resistor 50 Around 45 degrees. In the simulation, it was found that with matching termination inductance, reducing the electrical length of the coupling line to approximately 45 degrees maintained switching performance. To compensate for the performance loss, [the following was added]... and Parameter optimization is performed while providing bias to the transistor.

[0121] Transistor terminated by coupling line and The baseband circuit signal operates in the variable resistance region, where its on-resistance needs to be 50Ω. The left and right sides are designed to ensure that the terahertz wave is completely absorbed in the transistor's on-resistance when the modulator is off. Due to the design of the coupling line, the transistor's parasitic capacitance will not affect the switching matching.

[0122] In addition, to further improve the modulation rate, a matching design was implemented for the circuit connected to the base of the transistor (hereinafter referred to as the baseband circuit). By ensuring that its input impedance is equal to the characteristic impedance of the transmission line connected to the GSG, the output resistance is matched with the impedance of the transmission path when an external high-speed signal source is connected, thus preventing ringing in the high-speed baseband signal.

[0123] and This constitutes the baseband circuit described above. It is a short, thin wire that provides a certain inductance value to reduce the impact of the parasitic capacitance of the transistor base on the baseband bandwidth. It is a segment with a characteristic impedance of 50. The transmission line facilitates GSG compatibility.

[0124] Specific effects are as follows Figure 10 As shown, Figure 10 These are the S-parameters of this switch under static conditions, scanning from the RF input to the output port from 120 to 160 GHz. They can be divided into two sets of S21 and S11 curves based on the on and off states. As can be seen from the figure, the S21 (insertion loss) in the on state remains within the 2.2 dB range within the frequency band, and the S21 (isolation) in the off state remains above 21 dB. This indicates that the structure has the advantages of low insertion loss, high isolation, high bandwidth, and high flatness. These advantages are brought about by the coupling line + optimized impedance switching network design. At the same time, the S11 is below -15 dB in both on and off states, indicating that the structure has excellent matching performance in both states, fully achieving the design purpose of an absorptive switch.

[0125] The embodiments described above merely illustrate specific implementation methods of this application, and while the descriptions are detailed and specific, they should not be construed as limiting the scope of protection of this application. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of the technical solution of this application, and these modifications and improvements all fall within the scope of protection of this application.

Claims

1. An absorption terahertz high-speed on-off switch modulator, characterized by, include: Coupled line directional coupler, first impedance switching network, second impedance switching network, bias circuit, baseband circuit; The through end and coupling end of the directional coupler are respectively connected to a first impedance switching network and a second impedance switching network with the same structure. One end of the first impedance switching network and one end of the second impedance switching network are respectively connected to a bias circuit through the directional coupler. The other ends of the first impedance switching network and the second impedance switching network are respectively connected to a baseband circuit. The input end of the directional coupler is connected to the RF input, and the isolation end of the directional coupler is connected to the RF output.

2. The kind of absorption type terahertz high speed switch modulator according to claim 1, its characterized in that, The first impedance switching network is a one-port network, and its port input impedance can be switched between different impedances.

3. The high-speed THz absorption switch modulator according to claim 1, wherein, The first impedance switching network is a transistor, and the base of the transistor is connected to the through terminal of the directional coupler. The transistor has the following characteristics: its switching impedance can be adjusted by adjusting its size and control voltage; its parasitic capacitance does not affect the high reflection coefficient of the port; and it results in small parasitic capacitance.

4. The absorption-type terahertz high-speed switching modulator according to claim 3, characterized in that, As the size of the transistor increases, both the on and off resistances decrease while the on and off capacitances increase. Conversely, as the on and off voltages increase, the on and off resistances decrease while the off capacitance decreases slightly. By controlling these two factors together, the required transistor equivalent parameters are obtained.

5. The absorption type terahertz high-speed on-off switch modulator according to claim 1, wherein The directional coupler includes an asymmetric coupling line.

6. The absorption type terahertz high-speed on-off switch modulator according to claim 1, wherein The baseband circuit includes a first transmission line, an open stub, a second transmission line, and a GSG pin connected in sequence. The end of the first transmission line away from the open stub is connected to the other end of the first impedance switching network and the other end of the second impedance switching network.

7. An absorption type terahertz high-speed on-off switch modulation method characterized by, The method employs an absorptive terahertz high-speed switching modulator as described in any one of claims 1-6, comprising: when the switch is open, the radio frequency signals leading to the through end and the coupling end of the directional coupler are completely reflected; the signals reflected from the through end and the coupling end are phase-shifted by 90 degrees and then phase-opposite at the input end, resulting in complete reflection and transmission entirely through the isolation end; while when the switch is closed, the output end of the switch can be regarded as the isolation end of the directional coupler, and the input radio frequency signal is completely consumed by the matching impedance of the through end and the coupling end, so that the switch achieves high isolation and low reflection.

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