A control method, system, and device for a switchable winding parameter inductor.

Abstract

This invention relates to the field of power electronic conversion control technology, and discloses a control method, system, and device for a switchable winding parameter inductor. The control method includes: when the inductor switching condition is met, locking the trigger window at the absolute valley point of the freewheeling phase; before reaching the absolute valley point, controlling the first and second AC switches to simultaneously conduct to form an overlapping conduction state to dissipate leakage inductance energy; during the overlapping conduction period, calculating the basic feedforward duty cycle according to the equivalent current increment constraint equation, and calculating the compensation duty cycle in combination with the equivalent nonlinear inductor decay trajectory, superimposing to generate a transient target duty cycle sequence; writing this sequence into a shadow register, turning off the first AC switch when reaching the absolute valley point, and synchronously loading the sequence to control the conversion circuit. This invention dissipates leakage inductance through overlapping conduction and calculates the transient duty cycle based on algebraic feedforward, suppressing voltage spikes and current surges during switching, and maintaining the excitation balance of the system.

Description

Technical Field

[0001] This invention relates to the field of power electronic conversion and control technology, specifically to a control method, system, and device for a switchable winding parameter inductor. Background Technology

[0002] Variable parameter inductors dynamically change their inductance by switching the connection state of different winding nodes. They are used in power electronic converters that need to accommodate different load conditions, and can optimize the steady-state operating efficiency and dynamic response capability of the system. In actual operating conditions, the control system usually needs to perform inductance switching operations online according to the changes in output load.

[0003] However, due to the physical characteristic that inductors cannot change current instantaneously and the existence of uncoupled leakage inductance in actual windings, when the switching array directly disconnects the original winding circuit under load, the energy stored in the leakage inductance loses its discharge path and generates high-voltage spikes at both ends of the AC switch, increasing the loss and breakdown risk of the underlying hardware. At the same time, the instantaneous jump in inductance directly breaks the original steady-state operating conditions of the converter. The main switch cannot match the new inductance parameters to maintain excitation balance under the duty cycle before switching, which in turn causes violent fluctuations in the inductor current, resulting in a significant drop or overshoot in the output voltage. Existing control schemes mostly rely on the existing proportional-integral feedback network of the converter to passively suppress such disturbances. However, the digital control loop has inherent sampling and calculation delays and cannot intervene in advance and output matching control commands during the transition phase of hardware parameter changes. This lag in feedback regulation makes it impossible for the system to maintain flux balance at the moment of parameter switching, affecting the dynamic stability of the power electronic converter and the operational reliability of the devices. Summary of the Invention

[0004] To address the shortcomings of existing technologies, this invention provides a control method, system, and device for a switchable winding parameter inductor. This solves the problem of high voltage spikes, current overshoot, and output voltage drops caused by leakage inductance and parameter abrupt changes when the inductance of a variable parameter inductor is switched online in existing control systems, which leads to the loss of excitation balance in the system.

[0005] To achieve the above objectives, the present invention provides the following technical solution: The first aspect of this invention provides a control method for a switchable winding parameter inductor, applied to a control system including a controller, a conversion circuit, and a variable parameter inductor, the control method comprising: The controller collects the electrical parameters of the conversion circuit. When the electrical parameters meet the preset inductance switching conditions, it generates a control command to switch the inductance from the initial inductance to the target inductance and pauses the closed-loop control program. The action trigger window is aligned to the freewheeling phase of the conversion circuit and locked at the absolute valley point of the inductance current. Before the time reaches the absolute valley point during the freewheeling phase, the first AC switch and the second AC switch in the variable parameter inductor are simultaneously turned on to form an overlapping conduction state, and a local closed loop is formed inside the variable parameter inductor to dissipate the energy stored in the leakage inductance. During the control cycle of maintaining the overlapping conduction state, the algebraic feedforward calculation program is started to calculate the basic feedforward duty cycle corresponding to the target inductance according to the equivalent current increment constraint equation, and calculate the compensation duty cycle for compensating the flux deviation according to the equivalent nonlinear inductance decay trajectory during this time period. The basic feedforward duty cycle and the compensation duty cycle are superimposed to generate a transient target duty cycle sequence. The transient target duty cycle sequence is written into the shadow register of the internal pulse width modulation module. When the system clock count reaches the absolute valley point, the first AC switch is turned off, and the transient target duty cycle sequence is loaded into the active register for subsequent conduction control. Then, the execution of the closed-loop control program is resumed.

[0006] Preferably, when the electrical parameters meet the preset inductance switching conditions, generating a control command to switch the inductance from the initial inductance to the target inductance includes: The average load current flowing through the variable parameter inductor is collected; When the average load current collected continuously exceeds the preset high-level static threshold or low-level static threshold, and the duration exceeds the preset anti-disturbance period, it is determined that the inductor switching condition is met, and the corresponding control command is generated.

[0007] Preferably, the pause closed-loop control program includes: The generated control command is written into the internal latch, and the system is switched from the normal closed-loop operation mode to the transient switching waiting mode according to the flip of the state machine flag bit. In the transient switching waiting mode, the proportional-integral-derivative feedback adjustment algorithm in the digital control loop is paused, and the current duty cycle control variable of the main switch is forcibly locked to the initial duty cycle of the previous switching cycle before the control command is received.

[0008] Preferably, aligning the action trigger window to the freewheeling phase of the conversion circuit and locking it at the absolute valley point of the inductor current includes: Read the data from the register in the internal pulse width modulation module, extract the time interval during which the conversion circuit is in the off state, and directly define it as the freewheeling stage; The input voltage and output voltage are collected synchronously, the peak value at the start point of the freewheeling phase is calculated, and the instantaneous value of the inductor current is calculated as a descending trajectory in the freewheeling phase by combining the demagnetizing voltage, initial inductance and time variable. When the conversion circuit is in continuous conduction mode, the moment when the time variable is equal to the switching cycle is determined as the absolute valley point moment; when the instantaneous value of the inductor current is detected to bottom out and return to zero in advance, it is determined to be in intermittent conduction mode and the time of returning to zero is directly locked as the absolute valley point moment.

[0009] Preferably, the control method further includes: aligning the control time base that triggers the AC switch array action to a fixed time offset interval before the absolute valley point time, based on the extracted absolute valley point time.

[0010] Preferably, controlling the first AC switch and the second AC switch to simultaneously conduct, forming an overlapping conduction state, includes: An overlap control sequence is generated based on a preset overlap conduction dead zone compensation logic; The output maintains a level to keep the first AC switch in the ON state, and extracts a preset overlap time as a time advance, and outputs a drive level to turn on the second AC switch before the absolute valley point arrives.

[0011] Preferably, the step of forming a local closed loop inside the variable parameter inductor to dissipate the energy stored in the leakage inductance includes: In the overlapping conduction state where the first AC switch and the second AC switch are simultaneously turned on, the two winding nodes of the variable parameter inductor with different lead-out nodes are simultaneously connected to the common bus, so that the coil segment between the two winding nodes and the two AC switches together form a local closed loop. A differential equation describing the decaying circulating current is established based on the uncoupled leakage inductance. The preset overlap time is calculated based on the time constant of the differential equation, the equivalent series resistance of the local closed loop, and the preset safety margin coefficient, and is used as the time reference for executing the second AC switch opening action.

[0012] Preferably, the step of calculating the basic feedforward duty cycle corresponding to the target inductance based on the equivalent current increment constraint equation includes: Based on the principle that the current increment during the switching period before switching is equal to the current increment after switching, an equivalent current increment constraint equation in a single cycle is constructed. The base feedforward duty cycle, which provides a static reference for maintaining the excitation balance of the system, is calculated by multiplying the initial duty cycle of the previous steady-state cycle by the target inductance and dividing by the initial inductance.

[0013] Preferably, the step of calculating the compensation duty cycle for compensating flux deviation based on the equivalent nonlinear inductance attenuation trajectory within the time period includes: Using the preset overlap time as an input variable, the equivalent nonlinear inductor decay trajectory characterizing the local magnetic field collapse process is calculated based on the mutual inductance in the parallel impedance network of the variable parameter inductor and the damping coefficient of the magnetic core material. A definite integral operation is performed on the attenuation trajectory within the preset overlapping time interval to convert the volt-second area loss caused by the nonlinear decrease of the inductance into the corresponding pulse width compensation amount, which is used as the compensation duty cycle.

[0014] Preferably, the step of superimposing the basic feedforward duty cycle and the compensated duty cycle to generate a transient target duty cycle sequence is performed using a linear superposition algorithm, so that the reconstructed transient target duty cycle sequence includes a correction factor for compensating for hardware switching states.

[0015] Preferably, the step of turning off the first AC switch and loading the transient target duty cycle sequence into the activity register when the system clock count reaches the absolute valley point includes: Configure the hardware counter trigger source of the internal pulse width modulation module, and set the matching count value used to trigger the register data update to the system clock count corresponding to the absolute valley point time; When the count matches, a trigger update action is generated, the first AC switch is turned off through the underlying hardware routing, and the transient target duty cycle sequence in the shadow register is loaded into the active register simultaneously.

[0016] Preferably, the execution of the recovery closed-loop control program includes: When the system enters the next switching cycle, the conversion circuit performs turn-on and turn-off operations according to the transient target duty cycle sequence in the active register; Clear the control command state in the internal latch and reset and initialize the integral history state variables in the digital control loop before restoring the closed-loop control program, resetting them to initial values ​​that match the basic feedforward duty cycle.

[0017] A second aspect of the present invention provides a control system for a switchable winding parameter inductor, comprising a controller, a conversion circuit, and a variable parameter inductor; The conversion circuit includes a main switch transistor, the control terminal of which is connected to the controller and is driven by the pulse width modulation signal output by the controller to perform power conversion. The variable parameter inductor is connected to the conversion circuit and includes a main magnetic core and a multi-tap winding. The multi-tap winding includes a first winding node and a second winding node with different physical lead-out nodes. The variable parameter inductor also includes an AC switch array, which includes a first AC switch and a second AC switch, and the control terminals of the first AC switch and the second AC switch are respectively connected to the independent output pins of the controller. The controller is configured to perform the control method for the switchable winding parameter inductor described in the first aspect above.

[0018] Preferably, both the first AC switch and the second AC switch are constructed using anti-series metal-oxide-semiconductor field-effect transistors; the power supply terminal of the first AC switch is connected to the first winding node to receive the initial inductance, and the power supply terminal of the second AC switch is connected to the second winding node to receive the target inductance.

[0019] A third aspect of the present invention provides an electronic device, comprising: Bus; memory for storing computer program instructions; processor connected to the memory via the bus; and control system for the switchable winding parameter inductor described in the second aspect above; The processor is used to read and execute computer program instructions in the memory to drive the control system to execute control logic, and the electronic device provides a power supply network and heat dissipation structure for the control system.

[0020] This invention provides a control method, system, and device for an inductor with switchable winding parameters. It offers the following advantages: 1. The method provided by the present invention controls the first AC switch and the second AC switch to be turned on simultaneously to form an overlapping conduction state during the freewheeling stage and before the absolute valley point is reached. This operation constructs a local closed loop inside the variable parameter inductor, directly dissipates the energy stored in the uncoupled leakage inductance, avoids the switch array from causing destructive voltage spikes due to current cut-off at the parameter switching node, and reduces the impact loss on the underlying hardware circuitry.

[0021] 2. This invention introduces algebraic feedforward calculation within the control cycle that maintains the overlapping conduction state. The basic feedforward duty cycle is solved based on the equivalent current increment constraint equation. The compensation duty cycle is calculated for the nonlinear degradation characteristics caused by the local closed loop. The transient target duty cycle sequence generated by superimposing the two directly cancels the flux deviation caused by the jump in inductor parameters, thereby suppressing the inductor current overshoot and output voltage drop during the inductor switching transient process.

[0022] 3. This invention utilizes the shadow register of the internal pulse width modulation module to preload the transient target duty cycle sequence, and configures a hardware trigger source to synchronously execute data updates and the cutting-off action of the first AC switch when the system clock matches to the absolute valley point. This underlying execution timing eliminates the software calculation delay of the digital control loop, realizes the precise alignment of transient compensation data and underlying physical hardware actions, thereby ensuring the smoothness of control logic handover and the stability of closed-loop recovery during system state switching. Attached Figure Description

[0023] Figure 1 This is a schematic diagram of the control system of the present invention; Figure 2 This is a schematic diagram of the structural framework of the electronic device of the present invention; Figure 3 This is a flowchart illustrating the control method of the present invention; Figure 4 This is a timing diagram illustrating the operation status monitoring and phase domain locking of the present invention; Figure 5 This is a schematic diagram illustrating the timing control principle of the overlapping conduction microstate injection of the present invention; Figure 6 This is a schematic diagram illustrating the control principle of the volt-second balance feedforward and transitional trajectory mapping of the present invention; Figure 7 This is a schematic diagram illustrating the principle of underlying register synchronization and physical execution in this invention; Figure 8 This is a schematic diagram of the transient comparison waveforms of the inductor current in this invention; Figure 9 This is a schematic diagram of the transient comparison waveforms of the output voltage of the present invention; Figure 10 This is a schematic diagram of the waveform structure for comparing the drain-source voltage of the first AC switch of the present invention.

[0024] Among them, 10 is the controller; 20 is the conversion circuit; 21 is the main switch; 30 is the variable parameter inductor; 31 is the first AC switch; 32 is the second AC switch; 40 is the bus; 50 is the memory; and 60 is the processor. Detailed Implementation

[0025] The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0026] Please see the appendix Figure 1 , Figure 1This is a schematic diagram of a control system according to an embodiment of the present invention. The present invention provides a control system including: a controller 10, a conversion circuit 20, and a variable parameter inductor 30.

[0027] The controller 10 has a floating-point arithmetic unit and a timing trigger circuit, which are used to perform state sampling, feedforward algebraic equation solving and pulse width modulation signal output control.

[0028] The conversion circuit 20 includes a main switch 21. The control terminal of the main switch 21 is connected to the controller 10. The main switch 21 is driven by the pulse width modulation signal output by the controller 10 and is used to perform the power conversion process of the conversion circuit 20.

[0029] The variable parameter inductor 30 is connected in the conversion circuit 20 as an energy storage and filtering element. The variable parameter inductor 30 includes a main magnetic core and a multi-tap winding wound on the main magnetic core. The multi-tap winding has different physical lead-out nodes, specifically including a first winding node and a second winding node.

[0030] The variable parameter inductor 30 includes an AC switch array, which is configured at the lead-out node of the multi-tap winding. The AC switch array includes a first AC switch 31 and a second AC switch 32, both of which are composed of anti-series metal-oxide-semiconductor field-effect transistors.

[0031] The power supply terminal of the first AC switch 31 is connected to the first winding node of the multi-tap winding for inputting the initial inductance. The power supply terminal of the second AC switch 32 is connected to the second winding node of the multi-tap winding for inputting the target inductance. The control terminals of the first AC switch 31 and the second AC switch 32 are respectively connected to the independent output pins of the controller 10.

[0032] Please see the appendix Figure 2 , Figure 2 This is a schematic diagram of the structural framework of an electronic device according to an embodiment of the present invention. The present invention provides an electronic device including: a bus 40, a memory 50, a processor 60, and the aforementioned control system.

[0033] The processor 60 is electrically connected to the memory 50 and the control system via the bus 40. The controller 10 in the control system can be connected to the bus 40 as an independent control chip and called by the processor 60; or the controller 10 can be directly physically integrated into the processor 60.

[0034] The memory 50 is used to store computer program instructions, and the processor 60 is used to read and execute the computer program instructions in the memory 50 to drive the control system to execute control logic. The electronic device, as a complete machine with a physical form, provides a power supply network and heat dissipation structure for the control system to support the physical implementation of the underlying method.

[0035] Please see the appendix Figure 3 , Figure 3 This is a flowchart illustrating a control method according to an embodiment of the present invention. The present invention provides a control method for a switchable winding parameter inductor, based on the aforementioned electronic device and its included control system, comprising the following steps: S1, the controller 10 collects the electrical parameters of the conversion circuit 20. When the electrical parameters meet the preset inductor switching conditions, the controller 10 generates a control command to switch the inductance from the initial inductance to the target inductance, writes the control command into the internal latch, and pauses the execution of the closed-loop control program. The controller 10 analyzes the pulse width modulation carrier state of the main switch 21, aligns the action trigger window to the freewheeling stage of the conversion circuit 20, and locks it at the absolute valley point of the inductor current. S2, before the conversion circuit 20 is in the freewheeling stage and the time reaches the absolute valley point, the controller 10 outputs a drive signal to the first AC switch 31 and the second AC switch 32 to control the first AC switch 31 and the second AC switch 32 to be turned on simultaneously, forming an overlapping conduction state. The duration of the overlapping conduction state is set to a preset overlapping time. In the overlapping conduction state, the first winding node and the second winding node are short-circuited by the first AC switch 31 and the second AC switch 32, forming a local closed loop inside the variable parameter inductor 30 to dissipate the energy stored in the leakage inductance. S3, during the control cycle of maintaining the overlapping conduction state, the controller 10 starts the algebraic feedforward calculation program to calculate the basic feedforward duty cycle of the corresponding target inductance according to the equivalent current increment constraint equation; the controller 10 uses the preset overlap time as the input variable to calculate the equivalent nonlinear inductance attenuation trajectory within this time period, and calculates the compensation duty cycle used to compensate for the flux deviation according to the equivalent nonlinear inductance attenuation trajectory; the controller 10 performs superposition operation on the basic feedforward duty cycle and the compensation duty cycle to generate a transient target duty cycle sequence; S4, the controller 10 writes the transient target duty cycle sequence into the shadow register of the internal pulse width modulation module, configures the hardware trigger source so that a trigger update action is generated when the system clock count corresponds to the absolute valley point; when the system clock count reaches the absolute valley point, the hardware trigger source turns off the first AC switch 31 and simultaneously loads the transient target duty cycle sequence from the shadow register into the active register; the main switch 21 is turned on in the subsequent switching cycle according to the transient target duty cycle sequence in the active register, the controller 10 clears the control instruction state in the internal latch, and resumes the execution of the closed-loop control program.

[0036] Please see the appendix Figure 4 , Figure 4This is a timing diagram for operation status monitoring and phase domain locking according to an embodiment of the present invention. Regarding the execution logic and underlying physical mechanism of the above step S1, the present invention refines it into the following sub-steps in specific implementation.

[0037] The controller 10 collects the average load current flowing through the variable parameter inductor 30 through the internal analog-to-digital conversion module, and uses a preset fixed-width hysteresis comparison logic to determine the inductor switching conditions. In this embodiment, the preset fixed width can be configured according to actual application requirements, and its value range is set to 5% to 10% of the rated load current of the conversion circuit.

[0038] The controller 10 is pre-configured with a high static threshold calibrated according to the peak-to-peak value of the rated ripple current of the conversion circuit and a low static threshold calibrated according to the load drop tolerance.

[0039] As a preferred approach, the high-level static threshold can be flexibly set, with its value range being 110% to 120% of the rated load current of the converter circuit, and the low-level static threshold range being 20% ​​to 30% of the rated load current of the converter circuit.

[0040] When the average load current value collected continuously exceeds the high-level static threshold or the low-level static threshold, and the duration exceeds the preset anti-disturbance period, the controller 10 determines that the inductor switching condition is met and generates the corresponding control command; wherein, the specific value of the preset anti-disturbance period can be set to the switching cycle of 3 to 5 consecutive pulse width modulation signals according to the anti-interference requirements.

[0041] As a preferred approach, after generating the control command, the controller 10 writes it into the internal latch. Based on the flipping of the state machine flag bit, the system is switched from the normal closed-loop operation mode to the transient switching waiting mode. In this transient switching waiting mode, the controller 10 suspends the call to the proportional-integral-derivative feedback adjustment algorithm in the digital control loop and forcibly locks the current duty cycle control variable of the main switch tube 21 to the initial duty cycle of the previous switching cycle before receiving the control command.

[0042] The controller 10 reads the data from the comparison register in the internal pulse width modulation module, uses the comparator logic to extract the time interval during which the main switch 21 is in the off state, and directly defines this time interval as the freewheeling stage of the converter circuit 20. The controller 10 configures the conventional hardware parameters based on the existing power electronic topology for the forward voltage drop characteristics of the diodes in the freewheeling loop of the converter circuit 20 (with values ​​between 0.7V and 1.2V) and the dead time setting of the synchronous rectification network (with a preset value range of 20 nanoseconds to nanoseconds).

[0043] After defining the time boundary of the freewheeling phase, the controller 10 establishes a synchronous mapping relationship between the hardware timer count value and the current time-domain waveform. The controller 10 synchronously acquires the input voltage and output voltage of the conversion circuit 20 through the internal analog-to-digital conversion module to obtain the applied voltage across the time-varying parameter inductor when the main switch is in the conduction phase and the demagnetizing voltage during the freewheeling phase. Based on the superposition of the acquired average load current and half of the system current ripple, the controller 10 calculates the peak value of the inductor current at the beginning of the freewheeling phase. The controller 10 uses circuit equations to calculate the decreasing trajectory of the instantaneous value of the inductor current during the freewheeling phase. ; In the formula, This represents the instantaneous value of the inductor current. This is the peak value of the inductor current at the beginning of the freewheeling phase. This refers to the demagnetizing voltage across the variable parameter inductor during the freewheeling phase. This represents the initial inductance of the current connected conversion circuit; It is a time variable; The on-time of the main switch in the current cycle.

[0044] A switching period representing the pulse width modulation carrier is set, where the value of the time variable is between the conduction time and the switching period. When the conversion circuit is in continuous conduction mode, the controller 10 determines the absolute valley point when the time variable is equal to the switching period. When the controller 10 detects that the instantaneous value of the inductor current reaches zero ahead of time, it determines that the system is in intermittent conduction mode and directly locks the zero point moment as the absolute valley point moment.

[0045] Furthermore, the controller 10 aligns the control time base that triggers the AC switch array operation to a fixed time offset interval before the absolute valley point time based on the extracted absolute valley point time.

[0046] The value range of this fixed-time bias interval can be determined according to the dynamic characteristics of the switching device, and it is set to 1% to 5% of the switching period of the pulse width modulation signal.

[0047] Please see the appendix Figure 5 , Figure 5 This is a timing control principle diagram of overlapping conduction microstate injection according to an embodiment of the present invention. Based on the aforementioned precisely locked phase domain reference, the present invention delves into the underlying hardware timing in step S2 to construct a local energy discharge path, which is specifically refined into the following sub-steps.

[0048] Within the interval where the controller 10 determines that the system operating trajectory has entered the freewheeling phase and the time has not reached the absolute valley point, the controller 10 generates an overlapping control sequence according to the preset overlapping conduction dead zone compensation logic. The value range of the dead zone compensation time can be set according to the actual switching transistor type, and is preset to 10 nanoseconds to 50 nanoseconds. The controller 10 outputs a sustaining level to keep the first AC switch 31 in the conducting state. The controller 10 extracts the preset overlapping time as a time advance and outputs a drive level to turn on the second AC switch 32 before the absolute valley point arrives.

[0049] In the overlapping conduction state where the second AC switch 32 is on and the first AC switch 31 is not off, the first winding node and the second winding node of the variable parameter inductor 30 are simultaneously connected to the common bus by the first AC switch 31 and the second AC switch 32, which are in the conduction state. The coil segment between the first winding node and the second winding node, together with the first AC switch 31 and the second AC switch 32, form a local closed loop.

[0050] In the local closed loop, there exists a damped circulating current generated by the uncoupled leakage inductance of the variable parameter inductor 30. The controller 10 establishes a differential equation describing this damped circulating current: ; In the formula, The non-coupled leakage inductance of the variable parameter inductor between the first winding node and the second winding node is measured offline and pre-configured in the internal storage space of the controller 10. For the symbols of differential operators; The instantaneous value of the decaying circulating current flowing within a local closed loop; It is a time variable; This is the equivalent series resistance of a local closed loop.

[0051] The equivalent series resistance is the sum of the internal resistance of the first AC switch 31, the internal resistance of the second AC switch 32, and the DC resistance of the coil between the first winding node and the second winding node.

[0052] As a preferred method, the controller 10 calculates the preset overlap time based on the time constant of the differential equation: ; In the formula, Preset overlap time; The safety margin factor is set to a value of 3 to 5, which can be selected according to the reliability requirements of the system design. The non-coupled leakage inductance of the variable parameter inductor between the first winding node and the second winding node is measured offline and pre-configured in the internal storage space of the controller 10. This is the equivalent series resistance of a local closed loop.

[0053] The controller 10 uses the calculated preset overlap time as the timing reference for the timing advance of the second AC switch 32 opening action.

[0054] Please see the appendix Figure 6 , Figure 6 This is a control principle diagram of volt-second balance feedforward and transitional trajectory mapping according to an embodiment of the present invention. When the underlying physical network is in a controlled overlapping microstate, the system control is synchronously handed over to the feedforward calculation loop. In implementing step S3, the present invention establishes a multi-dimensional algebraic equation to compensate for dynamic distortion. The specific implementation details are as follows.

[0055] During the current control cycle in which controller 10 maintains the overlapping conduction state, due to the physical changes in the internal parameters of the system, conventional proportional-integral closed-loop calculations often lead to control divergence due to the lag characteristics of the integrator. During this period, controller 10 initiates an algebraic feedforward calculation program. The primary task of this program is to solve for the duty cycle reference under the ideal step model. Based on the current continuity and flux conversion principle of the power electronic converter, in order to prevent drastic changes in system current during the transient period of inductance switching, the current increment of the inductor during the conduction of the switching transistor should remain equal to the current increment before switching. Controller 10 calculates the basic feedforward duty cycle required to fully switch to the target inductance based on the equivalent current increment constraint equation within a single cycle: ; In the formula, The initial duty cycle of the controller in the steady-state cycle before the controller receives the control command; The switching period of the pulse width modulation signal; The applied voltage across the inductor when the main switch is in the on-state; This is the initial inductance. Based on the feedforward duty cycle; The target inductance.

[0056] After algebraic simplification, we can obtain: ; In the formula, Based on the feedforward duty cycle; The initial duty cycle of the controller in the steady-state cycle before the controller receives the control command; The target inductance; This is the initial inductance.

[0057] Through the above equation constraints, the basic feedforward duty cycle can provide a static reference for maintaining the excitation balance of the system.

[0058] Because the controlled overlapping conduction microstate disrupts the ideal step characteristics of inductor switching, the variable parameter inductor 30 exhibits nonlinear degradation characteristics during this stage. The controller 10 uses the preset overlap time as a known input variable and substitutes it into the trajectory solving equation. The controller 10 analyzes the equivalent impedance network when the windings are connected in parallel to calculate the equivalent nonlinear inductor decay trajectory during this time period. ; In the formula, The equivalent transition state inductance during the overlapping time period; This is the initial inductance. The target inductance; The mutual inductance between the coils at the first winding node and the second winding node; is the base of the natural logarithm; The damping coefficient, whose value range is pre-tuned to 10 by offline fitting of the loss characteristics of the magnetic core material. 3 Up to 10 5 s -1 Between these, the loss characteristics of the magnetic core material are pre-tuned by offline fitting; It is a time variable.

[0059] This attenuation trajectory reflects the local magnetic field collapse process caused by the physical short circuit. Since the decrease in equivalent inductance will cause the slope of the freewheeling current to decrease to be distorted, the controller 10 further calculates the compensation duty cycle for compensating for the flux linkage deviation based on the attenuation trajectory of the equivalent nonlinear inductance: ; In the formula, To compensate for the duty cycle; For constant molecules; The switching period of the pulse width modulation signal; The applied voltage across the inductor when the main switch is in the on-state; The lower limit of the definite integral represents the start time of the overlapping time interval; The preset overlap time serves as the upper limit for the definite integral; This refers to the demagnetizing voltage across the variable parameter inductor during the freewheeling phase. The target inductance; The equivalent transition state inductance during the overlapping time period; The time derivative element of the integral variable; It is a time variable.

[0060] The controller 10 converts the volt-second area loss caused by the nonlinear decrease of the inductance into the corresponding pulse width compensation amount by performing definite integral calculation on the preset overlapping time interval.

[0061] After obtaining the static reference base and the transient compensation amount respectively, the controller 10 performs a duty cycle sequence reconstruction operation. In this embodiment, as a preferred method, the controller 10 uses a linear superposition algorithm to superimpose the basic feedforward duty cycle and the compensation duty cycle to generate the transient target duty cycle sequence: ; In the formula, This is a transient target duty cycle sequence; Based on the feedforward duty cycle; To compensate for the duty cycle.

[0062] The transient target duty cycle sequence contains a special correction factor to compensate for the micro-state of hardware switching, which enables the reconstructed digital waveform to closely match the underlying physical changes of the current cycle, thereby providing driving data support for navigating the transient region.

[0063] Please see the appendix Figure 7 , Figure 7 This is a schematic diagram of the underlying register synchronization and physical execution according to an embodiment of the present invention. After completing the transient duty cycle reconstruction, the present invention completes the handover of control through register preloading and hardware synchronization triggering mechanism in step S4, which is specifically detailed as the following sub-steps.

[0064] After the controller 10 completes the generation of the transient target duty cycle sequence, the controller 10 writes the transient target duty cycle sequence into the shadow register of the internal pulse width modulation module through the internal data bus. The controller 10 configures the trigger source of the hardware counter of the internal pulse width modulation module. The controller 10 sets the matching count value used to trigger the update of the register data to the system clock count corresponding to the absolute valley point. The quantization step size resolution of the system clock count is preset to between 5 nanoseconds and 10 nanoseconds, which can be preset according to the processing performance of the controller.

[0065] When the system clock count matches the matching count value corresponding to the absolute valley point, the hardware trigger source generates a trigger update action. At the moment the trigger update action is generated, the underlying hardware router outputs a control level to turn off the first AC switch 31. The underlying hardware router synchronously loads the transient target duty cycle sequence loaded in the shadow register into the active register of the internal pulse width modulation module.

[0066] When the system enters the next switching cycle, the main switch 21 in the converter circuit 20 performs turn-on and turn-off operations according to the transient target duty cycle sequence loaded in the active register. The controller 10 clears the control instruction state in the internal latch. Before restoring the closed-loop control program, the controller 10 resets and initializes the integral historical state variables in the digital control loop. The controller 10 resets the integral historical state variables to the initial values ​​that match the basic feedforward duty cycle, and the system resumes the closed-loop operation mode.

[0067] To further aid in understanding the technical solution of this invention and to verify the effectiveness of the control method for the switchable winding parameter inductor provided by this invention, the following provides an application example, experimental verification, and effect comparison, in conjunction with a specific application scenario (server motherboard core power supply module, VRM).

[0068] Application Examples: In this embodiment, the control system is applied to the core power supply module of the server's motherboard. The input voltage of the converter is set to 12V, the output voltage of the converter is set to 1V, the initial inductance of the variable parameter inductor is set to 1.0 microhenries, and the target inductance of the variable parameter inductor is set to 0.2 microhenries.

[0069] When the load current increases, the controller continuously collects the average load current. When the collected average load current exceeds the preset high static threshold for three consecutive switching cycles, the controller generates a control command to switch the inductance from 1.0 microhenries to 0.2 microhenries. The controller then switches the system into transient switching waiting mode and suspends the call to the proportional-integral-derivative feedback regulation algorithm in the digital control loop.

[0070] The controller extracts the freewheeling phase of the current pulse width modulation signal. Based on the calculated moment when the inductor current drops to the absolute valley point, within a 30-nanosecond time interval before the arrival of the absolute valley point, the controller outputs a drive level to turn on the second AC switch, while simultaneously outputting a sustain level to keep the first AC switch in the conducting state. The first AC switch and the second AC switch are turned on together to form a local closed loop.

[0071] Within the 30 nanosecond time interval for forming a local closed loop, the controller superimposes the base feedforward duty cycle and the compensation duty cycle of the 0.2 microhenry target inductance, and writes the generated transient target duty cycle sequence into the internal shadow register. When the system clock count reaches the absolute valley point, the hardware trigger source turns off the first AC switch and synchronously loads the transient target duty cycle sequence in the shadow register into the active register. The main switch turns on in the subsequent switching cycle according to the duty cycle parameter in the active register, and the controller resumes the execution of the closed-loop control program.

[0072] Experimental verification and effect comparison: The experiment included two control groups: Traditional control group: adopts conventional hard switching logic, that is, directly turns off the first AC switch and immediately turns on the second AC switch, and does not have a duty cycle feedforward compensation algorithm.

[0073] The control group of this invention adopts the control method proposed in this invention, which includes overlapping conduction microstate injection and volt-second balance feedforward compensation.

[0074] Running the two comparison groups above, the software's built-in data checker extracts the maximum overshoot of the inductor current, the maximum drop in output voltage, the peak value of the drain-source voltage of the first AC switch, and the system's recovery time to the nominal voltage at the moment when the inductance switches from 1.0 microhenries to 0.2 microhenries. The extreme values ​​of the electrical parameters of the four test items are quantified and registered, and a comparison data table of transient electrical parameters of inductor switching is generated.

[0075] Comparison table of transient electrical parameters for inductor switching: ; From the table above and appendix Figure 8 To be continued Figure 10 We can obtain: Please see the appendix Figure 8 , Figure 8 This is a schematic diagram of the transient comparison waveform of inductor current according to the present invention. The horizontal axis represents time, and the vertical axis represents inductor current. Combined with the maximum overshoot data of inductor current recorded in the transient electrical parameter comparison data table for inductor switching, the maximum overshoot value of the inductor current waveform curve of the traditional control group at the moment of inductor switching is recorded as 18.5 Amperes, while the maximum overshoot value of the inductor current waveform curve of the control group of the present invention at the moment of inductor switching is recorded as 0.4 Amperes. The maximum overshoot of the inductor current of the control group of the present invention is lower than that of the traditional control group.

[0076] Please see the appendix Figure 9 , Figure 9This is a schematic diagram of the transient output voltage comparison waveform of the present invention. The horizontal axis represents time, and the vertical axis represents output voltage. Combining the data on the maximum output voltage drop and the system recovery nominal voltage adjustment time recorded in the inductor switching transient electrical parameter comparison data table, the maximum output voltage drop of the traditional control group at the inductor switching moment is recorded as 52.0 mV, while the maximum output voltage drop of the present invention control group at the inductor switching moment is recorded as 8.0 mV. The adjustment time from the start of the drop to the recovery of the nominal voltage of the traditional control group is recorded as 15.0 microseconds, while the adjustment time from the start of the drop to the recovery of the nominal voltage of the present invention control group is recorded as 2.5 microseconds. The maximum output voltage drop of the present invention control group is lower than that of the traditional control group, and the system recovery nominal voltage adjustment time of the present invention control group is shorter than that of the traditional control group.

[0077] Please see the appendix Figure 10 , Figure 10 This is a schematic diagram of the waveform structure of the drain-source voltage comparison of the first AC switch of the present invention. The horizontal axis of the figure represents time, and the vertical axis represents the drain-source voltage. Combined with the peak value data of the drain-source voltage of the first AC switch recorded in the inductor switching transient electrical parameter comparison data table, the peak value of the drain-source voltage waveform curve of the first AC switch of the conventional control group at the moment of the first AC switch being turned off is recorded as 28.4 volts, while the peak value of the drain-source voltage waveform curve of the first AC switch of the control group of the present invention at the moment of the first AC switch being turned off is recorded as 12.5 volts. The peak value of the drain-source voltage of the first AC switch of the control group of the present invention is lower than that of the conventional control group.

[0078] Although embodiments of the invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the appended claims and their equivalents.

Claims

1. A control method of a switchable winding parameter inductor, characterized by, The method is applied to a control system comprising a controller, a conversion circuit, and a variable parameter inductor, wherein the control method includes: The electrical parameters of the conversion circuit are obtained. When the electrical parameters meet the preset inductance switching conditions, a control command is generated to switch from the initial inductance to the target inductance, and the action trigger window is locked at the absolute valley point of the inductance current. During the freewheeling phase of the conversion circuit and before the absolute valley point is reached, the first AC switch and the second AC switch in the variable parameter inductor are simultaneously turned on to form an overlapping conduction state. During the control cycle that maintains the overlapping conduction state, the basic feedforward duty cycle is calculated according to the equivalent current increment constraint equation, and the compensation duty cycle is calculated according to the equivalent nonlinear inductor decay trajectory. The basic feedforward duty cycle and the compensation duty cycle are superimposed to generate a transient target duty cycle sequence. When the time reaches the absolute valley point, the first AC switch is turned off, and the conversion circuit is controlled according to the transient target duty cycle sequence.

2. The control method of a switchable winding parameter inductor according to claim 1, characterized in that, The process of acquiring the electrical parameters of the conversion circuit, generating a control command to switch from the initial inductance to the target inductance when the electrical parameters meet the preset inductance switching conditions, and locking the action trigger window at the absolute valley point of the inductance current includes: The average load current flowing through the variable parameter inductor is collected. When the average load current continuously exceeds a preset static threshold and the duration exceeds a preset anti-disturbance period, it is determined that the inductor switching condition is met, and the control command is generated. The control command is written into the latch, the closed-loop adjustment algorithm in the digital control loop is paused, and the current duty cycle control variable of the main switch in the conversion circuit is locked to the initial duty cycle of the previous switching cycle before the control command is received.

3. The control method of a switchable winding parameter inductor of claim 1, wherein, The step of locking the action trigger window at the absolute valley point of the inductor current includes: Read the data from the comparison register in the pulse width modulation module, extract the time interval during which the conversion circuit is in the off state, and define it as the freewheeling phase; The input and output voltages of the conversion circuit are collected, and the instantaneous value of the inductor current is calculated along with the demagnetizing voltage and the initial inductance during the freewheeling phase. The absolute valley point time is locked based on the switching cycle in the continuous conduction mode or the instantaneous value of the inductor current in the intermittent conduction mode of the conversion circuit.

4. The control method of a switchable winding parameter inductor of claim 1, wherein, The control of simultaneously turning on the first AC switch and the second AC switch in the variable parameter inductor to form an overlapping conduction state includes: The output maintains the first AC switch on, and extracts a preset advance time, and outputs a drive level to turn on the second AC switch before the absolute valley point arrives; During the overlapping conduction state, the two winding nodes of the variable parameter inductor with different physical lead-out nodes are simultaneously connected to a common bus. The coil segment between the two winding nodes forms a local closed loop with the first AC switch and the second AC switch to dissipate the attenuated circulating current generated by the uncoupled leakage inductance in the variable parameter inductor.

5. The control method of a switchable winding parameter inductor of claim 1, wherein, The calculation of the basic feedforward duty cycle based on the equivalent current increment constraint equation includes: The equivalent current increment constraint equation is constructed based on the principle that the current increment during the conduction period of the main switch of the conversion circuit is equal before and after the inductance switching. The basic feedforward duty cycle is obtained by multiplying the initial duty cycle of the previous steady-state cycle before receiving the control command by the target inductance and dividing by the initial inductance.

6. The control method of a switchable winding parameter inductor of claim 1, wherein, The calculation of the compensation duty cycle based on the equivalent nonlinear inductor attenuation trajectory includes: Using the duration of the overlapping conduction state as an input variable, the equivalent nonlinear inductance attenuation trajectory is calculated based on the mutual inductance and damping coefficient of the variable parameter inductor. A definite integral operation is performed on the equivalent nonlinear inductor attenuation trajectory during the duration of the overlapping conduction state to convert the volt-second area loss caused by the nonlinear decrease of the inductor into the corresponding pulse width compensation amount, thereby obtaining the compensation duty cycle.

7. The control method of a switchable winding parameter inductor of claim 1, wherein, When the time reaches the absolute valley point, the first AC switch is turned off, and the conversion circuit is controlled according to the transient target duty cycle sequence, including: The generated transient target duty cycle sequence is written into the shadow register of the pulse width modulation module; The trigger source of the hardware counter is set. When the system clock count matches the absolute valley point, a trigger update action is generated to turn off the first AC switch. Simultaneously, the transient target duty cycle sequence in the shadow register is loaded into the active register to drive the main switch in the conversion circuit, and the control instruction state in the latch is cleared to restore the closed-loop regulation algorithm.

8. A control system for a switchable winding parameter inductor, characterised in that, The control method for a switchable winding parameter inductor according to any one of claims 1-7, wherein the system includes a controller, a conversion circuit, and a variable parameter inductor; The conversion circuit includes a main switch transistor, and the control terminal of the main switch transistor is connected to the controller. The variable parameter inductor is connected in the conversion circuit. The variable parameter inductor includes a first winding node, a second winding node, a first AC switch, and a second AC switch. The first AC switch is connected to the first winding node, and the second AC switch is connected to the second winding node. The control terminals of the first AC switch and the second AC switch are respectively connected to the controller. The controller is configured to perform the control method for a switchable winding parameter inductor as described in any one of claims 1 to 7.

9. A control system for a switchable winding parameter inductor as claimed in claim 8, characterised in that, Both the first AC switch and the second AC switch are constructed using anti-series metal-oxide-semiconductor field-effect transistors; the power supply terminal of the first AC switch is used to connect the initial inductance value, and the power supply terminal of the second AC switch is used to connect the target inductance value.

10. An electronic device, comprising: A control method for a switchable winding parameter inductor according to any one of claims 1-7, comprising: a processor, a memory, and a control system; The memory is used to store computer program instructions, and the processor is used to read and execute the computer program instructions in the memory to drive the control system.

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