A method for driving control strategy of an alternating current converter

Abstract

This invention relates to the field of power electronic conversion control technology, and discloses a drive control strategy method for an AC converter, including: triggering a transitional synchronization stage when the mains power bus voltage reaches the start-up reference; issuing a damping pre-modulation signal to control the inverter bridge arm to construct a current throughput loop across the bus capacitor, generating dynamic active damping to deform the bus voltage trajectory with the time-varying envelope of the AC rectified voltage; at the moment when the voltage coordination condition is met and the transient current of the high-power current-limiting resistor crosses zero, controlling the drive circuit of the power relay to conduct and engage to cut off the rectified limiting path, directly connecting the bus to the main power supply path. This invention eliminates contact arc damage and voltage overshoot caused by potential mismatch during start-up switching, achieves low-stress arc-free switching, and extends the mechanical and electrical life of the switching devices.

Description

Technical Field

[0001] This invention relates to a drive control strategy method for an AC converter, belonging to the field of power electronic conversion control technology. Background Technology

[0002] Currently, in the construction of AC / DC power conversion topologies, the use of a rectifier network in conjunction with an energy storage bus capacitor to form a DC voltage carrying center is a common circuit structure. Typically, a passive current-limiting impedance is introduced during the initial stage of high-power AC power conversion to suppress abnormal current surges generated by the transient charging of the bus capacitor, ensuring the initial electrical stress safety of the power devices in the front-end topology and the rear-end inverter bridge arm. The soft-start circuit mainly relies on the transient potential difference between the rectifier output potential and the bus capacitor voltage to drive the unidirectional flow of charge. Since the AC power input exhibits a periodic time-varying sinusoidal potential characteristic, while the DC bus voltage exhibits a nonlinear, progressively rising trajectory under the passive current-limiting charging path, the two potentials manifest as dynamically changing geometric envelopes in the time domain. This misalignment of waveform phase and deformation trajectory results in a dynamic potential difference at the moment of closing and switching of the power relay contact group responsible for disconnecting the soft-start impedance. Under the condition of independent AC power start-up, the high-frequency arc released by the instantaneous contact exacerbates the physical erosion of the contact surface and accumulates potential insulation degradation.

[0003] The mainstream linear improvement approach adopted in the industry often attempts to resist arc erosion by increasing the size of the relay physical contacts or increasing the contact current rating. However, this increases hardware space occupation and material costs, and also increases the design burden of the auxiliary power supply main circuit due to the increased power consumption required for the drive coil. Some solutions attempt to lengthen the pre-charge cycle in the hope that the bus capacitor voltage will reach saturation, but the AC sinusoidal voltage still maintains periodic dynamic fluctuations. Although hardware specification adjustments alleviate contact erosion, there are limitations in structural improvement and shortcomings in the control method. For example, Chinese invention patent CN217159264U discloses a fast power-off and power-on protection circuit. The method obtains a square wave signal reflecting the AC power supply status through a voltage divider network and comparator, and controls the relay operation by combining the static target value of the DC bus voltage. However, this strategy implicitly relies on the premise that the power grid conditions are ideal and the bus voltage changes monotonically. It can eliminate the dynamic deformation misalignment between the time-varying sinusoidal potential at the mains input and the asymptotic trajectory of the DC bus voltage. When facing complex power grid harmonic distortion or deep transient voltage drops in non-ideal environments, its fixed static threshold judgment logic is very easy to be fundamentally misaligned with the actual phase fluctuations of the power grid. Not only can it fail to achieve smooth switching of near-zero potential difference between the two ends of the contacts, but it may also cause the relay to reciprocate and chatter during power grid disturbances, leading to accelerated contact deterioration.

[0004] Therefore, how to dynamically track the AC rectified voltage envelope by controlling the DC bus voltage trajectory through high-frequency micro-amplitude damping modulation of the inverter bridge arm to achieve near-zero stress switching has become the technical problem to be solved by this invention. Summary of the Invention

[0005] To address the problems in the background art, the technical solution of the present invention is as follows: A drive control strategy method for an AC converter, comprising the following steps: Step S1: When the AC converter is in a state of no battery and single AC input, and the external input voltage meets the start-up threshold, the external input voltage is introduced by using the dual diode rectification path of the input terminal and the high-power current limiting resistor to charge the DC bus, and the photovoltaic auxiliary power supply path is connected simultaneously to complete the start-up of the control unit. Step S2: When the DC bus voltage rises to the transition synchronization level reference, a damping pre-modulation signal is sent to control the inverter bridge arm to enter the damping pre-modulation state. By adjusting the drive modulation duty cycle of the inverter bridge arm in real time, the transient voltage difference generated across the normally open contact group of the power relay on the main power supply path is adjusted to within the low-stress arc-free switching voltage window. Step S3: When the transient voltage difference remains within the low-stress arc-free switching voltage window and the external grid voltage variation rate gradient meets the safety slope threshold, the drive circuit of the control power relay is turned on to make the normally open contact group close, cut off the rectifier limiting path composed of the high-power current limiting resistor, and directly connect the DC bus to the main power supply path. Step S4: After the normally open contact group completes the engagement action and the contact mechanical bounce stabilizes and delays, the damping pre-modulation signal is stopped, and the drive control logic smoothly switches to the normal power inverter pulse width modulation state to complete the safe start-up.

[0006] Preferably, before the normally open contact group is closed in step S3, the external grid voltage variation rate gradient is continuously calculated using the digital filter inside the control unit. When the external AC grid experiences a voltage drop or high-frequency harmonic distortion, causing the external grid voltage variation rate gradient to exceed the safety slope threshold, the control unit activates the lockout logic, cuts off the level flip enable of the control signal, keeps the drive circuit in a non-operational resting state, and the inverter bridge arm continues to maintain the damped pre-modulation state.

[0007] Preferably, in step S2, when adjusting the transient voltage difference generated across the normally open contact group to within the low-stress arc-free switching voltage window, the line operating temperature is collected in real time using a physical thermistor located around the DC bus; the control unit calculates the thermistor impedance drift of the high-power current-limiting resistor during the power-on charging process based on the line operating temperature, and dynamically tightens the upper limit of the switching threshold of the low-stress arc-free switching voltage window as the line operating temperature increases. The upper limit of the switching threshold satisfies the formula: ,in, This is the upper limit of the tightened switching threshold. To switch the voltage reference constant, This is the temperature correction factor. This represents the thermistor drift.

[0008] Preferably, when the line operating temperature increases and the upper limit of the switching threshold of the low-stress arc-free switching voltage window is tightened, the control unit synchronously extends the upper limit of the transition pre-modulation time for maintaining the damping pre-modulation state in step S2 to correct the overcurrent deviation caused by the thermistor impedance drift of the high-power current-limiting resistor.

[0009] Preferably, in step S2, controlling the inverter bridge arm to enter the damping pre-modulation state includes the following sub-steps: Step S21, obtaining the real-time voltage phase and current frequency of the external AC power grid; Step S22, increasing the switching frequency of the inverter bridge arm to twice the operating frequency of the conventional inverter, and sending a damping pre-modulation signal to control the inverter bridge arm to output a controlled AC voltage with the same frequency, phase and amplitude as the external AC power grid, so that the two ends of the normally open contact group of the power relay are at the zero potential difference switching point.

[0010] Preferably, in step S3, the transient voltage difference being within the low-stress, arc-free switching voltage window refers to the moment when the control unit detects the intersection of the transient voltage across the normally open contact group and the main power supply path loop current simultaneously falling into the set dual voltage and current window.

[0011] Preferably, in step S4, after the normally open contact group completes the closing action and the mechanical bounce of the contact is stabilized and delayed, the control unit maintains the damping pre-modulation signal continuously for 5ms after the normally open contact group completes the closing action. After the mechanical bounce of the normally open contact group is eliminated and the contact state is stabilized, the drive control logic is smoothly switched to the conventional power inverter pulse width modulation state.

[0012] Preferably, after the control unit activates the latching logic and keeps the drive circuit in a dormant state, the digital filter continuously monitors the voltage fluctuations of the external AC power grid; when the voltage fluctuations of the external AC power grid are eliminated and the calculated external power grid voltage variation rate gradient recovers to below the safety slope threshold, the control unit automatically closes the latching logic, restores the level flip enable of the control signal, and allows the normally open contact group to close again.

[0013] Preferably, in step S1, an external input voltage is introduced through the dual diode rectifier circuit of the input terminal and the high-power current-limiting resistor, eliminating the need to set up a separate AC auxiliary power supply for the main circuit. The dual diode rectifier circuit of the input terminal includes a first rectifier diode and a second rectifier diode, and the high-power current-limiting resistor is connected between the output terminal of the dual diode rectifier circuit and the DC bus.

[0014] Preferably, in step S3, controlling the power relay drive circuit to conduct so that the normally open contact group is energized and energized by the DC 12V coil to drive the power relay contacts to operate, switching the rectifier limiting path to the normally open power supply state, and cutting off the rectifier limiting path.

[0015] Compared with the prior art, the beneficial effects of the present invention are: 1. In the drive control strategy of the AC converter, the control unit collects the DC bus voltage and AC rectified voltage and calculates the difference. During the transition synchronization stage, the transition modulation signal is output to regulate the power switching tubes of the inverter bridge arm. A controllable current throughput loop is constructed on both sides of the bus capacitor to generate dynamic active damping, correct the DC bus voltage change trajectory and track the periodic time-varying envelope of the AC rectified voltage in real time. This makes the potential difference across the power relay contacts continuously converge to the low-stress switching threshold voltage. Combined with the zero-point capture of the transient current waveform flowing through the high-power current-limiting resistor, the soft-start control signal is flipped at the moment when the voltage and current windows intersect to make the contacts close without arcing, reducing the current impact on the downstream power devices at the moment of switching.

[0016] 2. During the transition and synchronization phase, the digital filter inside the control unit continuously calculates the rate of change gradient of the transient voltage difference. When the external AC power grid experiences a momentary voltage drop or a large-scale distortion of high-frequency harmonics in the power grid, causing the rate of change gradient to exceed the set safety slope threshold, the control unit adaptively shuts down the level flipping channel of the soft-start control signal, keeping the power relay in an inactive state and the inverter bridge arm in a micro-amplitude damping modulation state until the external AC power grid fluctuations are eliminated and the rate of change gradient returns to below the safety threshold. This avoids high-frequency reciprocating vibration of the power relay caused by irregular power grid disturbances and protects the mechanical configuration stability of the coil and contacts.

[0017] 3. By real-time acquisition of the line operating temperature through physical thermistors installed around the DC bus, the control unit calculates the thermistor impedance drift of the high-power current-limiting resistor during continuous start-up pre-charging based on the operating temperature. When the line is in a high-temperature state, the low-stress switching threshold voltage at both ends of the contact is dynamically tightened and the upper limit of the transition modulation time is extended to correct the overcurrent deviation caused by changes in physical impedance. This enables the AC converter to accurately lock the zero potential switching point within the temperature rise boundary caused by different ambient temperatures and frequent starts, enhancing the transient safety boundary and long-term thermal stability of the converter power circuit in a wide-temperature complex operating environment. Attached Figure Description

[0018] Figure 1 This is a flowchart of the soft-start control process for the AC converter of the present invention. Figure 2 This is a waveform diagram of DC bus voltage envelope tracking according to the present invention.

[0019] The objectives, features, and advantages of this invention will be further explained in conjunction with the embodiments and with reference to the accompanying drawings. Detailed Implementation

[0020] The technical solutions of the embodiments of this application will be clearly described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. All other embodiments obtained by those skilled in the art based on the embodiments of this application are within the scope of protection of this application.

[0021] A drive control strategy method for an AC converter includes the following steps: Step S1: When the AC converter is in a state of no battery and single AC input, and the external input voltage meets the start-up threshold, the external input voltage is introduced by using the dual diode rectification path of the input terminal and the high-power current limiting resistor to charge the DC bus, and the photovoltaic auxiliary power supply path is connected simultaneously to complete the start-up of the control unit. Step S2: When the DC bus voltage rises to the transition synchronization level reference, a damping pre-modulation signal is sent to control the inverter bridge arm to enter the damping pre-modulation state. By adjusting the drive modulation duty cycle of the inverter bridge arm in real time, the transient voltage difference generated across the normally open contact group of the power relay on the main power supply path is adjusted to within the low-stress arc-free switching voltage window. Step S3: When the transient voltage difference remains within the low-stress arc-free switching voltage window and the external grid voltage variation rate gradient meets the safety slope threshold, the drive circuit of the control power relay is turned on to make the normally open contact group close, cut off the rectifier limiting path composed of the high-power current limiting resistor, and directly connect the DC bus to the main power supply path. Step S4: After the normally open contact group completes the engagement action and the contact mechanical bounce stabilizes and delays, the damping pre-modulation signal is stopped, and the drive control logic smoothly switches to the normal power inverter pulse width modulation state to complete the safe start-up.

[0022] Preferably, before the normally open contact group is closed in step S3, the external grid voltage variation rate gradient is continuously calculated using the digital filter inside the control unit. When the external AC grid experiences a voltage drop or high-frequency harmonic distortion, causing the external grid voltage variation rate gradient to exceed the safety slope threshold, the control unit activates the lockout logic, cuts off the level flip enable of the control signal, keeps the drive circuit in a non-operational resting state, and the inverter bridge arm continues to maintain the damped pre-modulation state.

[0023] Preferably, in step S2, when adjusting the transient voltage difference generated across the normally open contact group to within the low-stress arc-free switching voltage window, the line operating temperature is collected in real time using a physical thermistor located around the DC bus; the control unit calculates the thermistor impedance drift of the high-power current-limiting resistor during the power-on charging process based on the line operating temperature, and dynamically tightens the upper limit of the switching threshold of the low-stress arc-free switching voltage window as the line operating temperature increases. The upper limit of the switching threshold satisfies the formula: ,in, This is the upper limit of the tightened switching threshold. To switch the voltage reference constant, This is the temperature correction factor. This represents the thermistor drift.

[0024] Preferably, when the line operating temperature increases and the upper limit of the switching threshold of the low-stress arc-free switching voltage window is tightened, the control unit synchronously extends the upper limit of the transition pre-modulation time for maintaining the damping pre-modulation state in step S2 to correct the overcurrent deviation caused by the thermistor impedance drift of the high-power current-limiting resistor.

[0025] Preferably, in step S2, controlling the inverter bridge arm to enter the damping pre-modulation state includes the following sub-steps: Step S21, obtaining the real-time voltage phase and current frequency of the external AC power grid; Step S22, increasing the switching frequency of the inverter bridge arm to twice the operating frequency of the conventional inverter, and sending a damping pre-modulation signal to control the inverter bridge arm to output a controlled AC voltage with the same frequency, phase and amplitude as the external AC power grid, so that the two ends of the normally open contact group of the power relay are at the zero potential difference switching point.

[0026] Preferably, in step S3, the transient voltage difference being within the low-stress, arc-free switching voltage window refers to the moment when the control unit detects the intersection of the transient voltage across the normally open contact group and the main power supply path loop current simultaneously falling into the set dual voltage and current window.

[0027] Preferably, in step S4, after the normally open contact group completes the closing action and the mechanical bounce of the contact is stabilized and delayed, the control unit maintains the damping pre-modulation signal continuously for 5ms after the normally open contact group completes the closing action. After the mechanical bounce of the normally open contact group is eliminated and the contact state is stabilized, the drive control logic is smoothly switched to the conventional power inverter pulse width modulation state.

[0028] Preferably, after the control unit activates the latching logic and keeps the drive circuit in a dormant state, the digital filter continuously monitors the voltage fluctuations of the external AC power grid; when the voltage fluctuations of the external AC power grid are eliminated and the calculated external power grid voltage variation rate gradient recovers to below the safety slope threshold, the control unit automatically closes the latching logic, restores the level flip enable of the control signal, and allows the normally open contact group to close again.

[0029] Preferably, in step S1, an external input voltage is introduced through the dual diode rectifier circuit of the input terminal and the high-power current-limiting resistor, eliminating the need to set up a separate AC auxiliary power supply for the main circuit. The dual diode rectifier circuit of the input terminal includes a first rectifier diode and a second rectifier diode, and the high-power current-limiting resistor is connected between the output terminal of the dual diode rectifier circuit and the DC bus.

[0030] Preferably, in step S3, controlling the power relay drive circuit to conduct so that the normally open contact group is energized and energized by the DC 12V coil to drive the power relay contacts to operate, switching the rectifier limiting path to the normally open power supply state, and cutting off the rectifier limiting path.

[0031] Example 1: When the system faces the operating condition of an AC converter with independent mains power input and no battery connected, the external AC grid experiences voltage drops and high-frequency harmonic distortion, causing dynamic fluctuations in the phase of the sinusoidal voltage at the input terminal. Simultaneously, changes in ambient temperature and frequent starts cause localized heat accumulation, leading to thermistor drift in the soft-start circuit. In the line's default trigger logic, the soft-start charging process and the power inverter drive are in a time-domain isolated state. At the moment of closing, the power relay's contact group experiences a transient potential difference due to the distortion and misalignment between the phase of the AC rectified voltage waveform and the progressive trajectory of the DC bus voltage. At the switching point, the parasitic inductance of the control circuit generates a transient induced electromotive force, inducing contact arcing, contact surface ablation, and surface welding. This results in a voltage spike and transient surge current at the DC bus terminal. When the external AC grid voltage passes through the input terminal... Introduced by CN20, when the initial charging voltage is established to the downstream DC bus via a dual-diode rectification path consisting of the first rectifier diode D43 and the second rectifier diode D44, and a high-power current-limiting resistor R245, the control unit completes startup through the photovoltaic auxiliary power supply path. Specifically, an isolated flyback switching power supply circuit connected between the positive and negative terminals of the DC bus serves as the physical carrier. When the external single-phase AC grid voltage charges the DC bus voltage to the hardware start-up threshold of 80V, the core of the flyback switching power supply circuit starts working. After high-frequency rectification and filtering on the secondary side, it stably outputs 12V and 5V DC levels to the power supply pins of the digital signal processor, forcing the digital signal processor to execute a reset release and load the register initialization configuration, outputting a stable high-level power supply enable state, enabling the control unit to complete the ready configuration, and monitoring the DC bus voltage. Real-time tracking voltage at the rectified voltage potential The electrical connection architecture of the AC converter main circuit of this invention is designed as follows: the input terminal is connected to an external single-phase AC power grid, and its output terminal is connected to the anode of the first rectifier diode and the anode of the second rectifier diode. The cathodes of the two diodes are connected in parallel to form the positive rectifier output. A high-power current-limiting resistor is connected in series between the positive rectifier output and the positive DC bus. A bus capacitor is connected in parallel between the positive and negative DC bus. The power switches of each phase of the inverter bridge arm are connected in series and then connected in parallel between the positive and negative DC bus, so that the inverter bridge arm and the bus capacitor form a direct parallel topology. The normally open contact group of the power relay on the main power supply path is connected in parallel between the positive rectifier output and the positive DC bus, that is, its normally open contact group and the high-power current-limiting resistor form a hardware parallel dual-path architecture.

[0032] When the power relay is in the normally open state, the rectifier limiting path is active, and the grid voltage can only passively charge the bus capacitor through the high-power current-limiting resistor. However, since the inverter arm power devices are directly connected in parallel across the bus capacitor, the control unit can activate a high-frequency switching signal to directly excite controlled charge throughput across the bus capacitor, completing the construction of the active damping circuit. As the charging process progresses, when the digital filter measures the DC bus voltage... When the voltage rises to the transition synchronization level reference, that is, when the digital value of the voltage across the DC bus capacitor captured by the control unit through the analog-to-digital converter chip reaches the discrete and fixed 280V potential boundary, this value is within the fluctuation range of the time-varying voltage trajectory after rectification by dual rectifier diodes of the external 220V single-phase AC power grid. This triggers the write value toggling of the state control register inside the control unit and starts the clock counter. The control unit maintains a constant level at the control signal output port, forcibly triggering a transition synchronization phase with a time span of 20ms to 50ms, and sends a damping pre-modulation signal to control the inverter bridge arm to enter the damping pre-modulation state. During the transition synchronization phase, the control unit adjusts the voltage based on the DC bus voltage. With real-time tracking voltage The real-time difference dynamically adjusts the drive modulation duty cycle of the inverter bridge arm, wherein the transient control duty cycle of the transition modulation signal is used. Satisfying the formula: ,in, The transient control duty cycle for the transition modulation signal. The voltage feedback gain coefficient is pre-stored in the control unit register, and its value ranges from 0.15 to 0.35; This is the DC bus voltage. The voltage feedback gain coefficient is a real-time tracking voltage at the AC rectified voltage potential. The range of values ​​for the aforementioned voltage feedback gain coefficient is locked by physical boundary derivation using the hardware loop parameters of the AC converter.

[0033] In engineering design, the gain coefficient has a deterministic correlation with the capacitance of the aforementioned bus capacitor, the rated frequency of the external AC power grid, and the nominal resistance of the high-power current-limiting resistor. To ensure that the active damping current generated by the inverter arm can effectively correct the bus voltage trajectory without causing closed-loop divergent oscillations in the control loop, the theoretical lower bound of the gain coefficient is determined by the minimum charge throughput required by the bus capacitor in a single switching cycle. Calculations show that when the corresponding capacitance is 820 microfarads and the current-limiting resistor is 47 ohms, the minimum gain boundary for stable tracking is 0.15. The theoretical upper bound of the gain coefficient is limited by the floating-point operation overhead of the control unit at a switching frequency of 100 kHz, which is twice the operating frequency of the conventional inverter, and the capacitor voltage ripple suppression limit. To prevent the bus voltage from generating high-frequency pulsations exceeding 5%, the upper bound of its gain feedback is constrained to 0.35.

[0034] Through the engineering derivation of the above hardware constraints, an optimal value range balancing tracking accuracy and control loop stability was finally established. During the transition synchronization phase before the power relay in the main power supply path engages, the AC side of the inverter bridge arm maintains a hardware electrical connection with the external single-phase AC power grid using a filter network composed of filter inductors and filter capacitors. The control unit uses an internally integrated second-order generalized integrator phase-locked loop algorithm to capture the current phase and frequency of the external single-phase AC power grid in real time. The internal digital signal processor periodically reads the real-time tracking voltage at the DC bus voltage and AC rectified voltage potential at a fixed interrupt frequency of 100 kHz. The voltage feedback gain coefficient is then used to calculate the... The transient control duty cycle of the modulation signal directly acts on the complementary pulse width modulation drive register of the inverter arm to change the on-time ratio of the upper and lower arm power switches. This controls the exchange of high-frequency pulsating reactive and active currents between the bus capacitor and the energy storage inductor in the filter network. The dynamic charging and discharging effect across the bus capacitor causes a nonlinear dip in the transient trajectory of the DC bus voltage towards the trough of the AC rectified voltage envelope. This high-frequency micro-amplitude damping modulation controls the operation of the inverter arm switches, constructing a controlled current throughput loop across the bus capacitor C28 to generate dynamic active damping, correcting the DC bus voltage change trajectory and ensuring the DC bus voltage... The transient trajectory generates nonlinear deformation to track the AC rectified voltage. The periodic time-varying envelope of the power relay RY4 on the main power supply path adjusts the transient voltage difference generated across the normally open contact group to within the low-stress arc-free switching voltage window. The two ends of the aforementioned normally open contact group are respectively connected between the positive terminal of the rectifier output and the positive terminal of the DC bus. The transient voltage difference across the contacts is essentially equal to the difference between the AC rectified voltage and the DC bus voltage. Although the main power supply path is in a hardware disconnected state before the normally open contact group is engaged, the high-frequency modulation action of its switching transistor can directly change the direction and amplitude of the current flowing through the bus capacitor, thereby adjusting the waveform of the transient trajectory of the DC bus voltage, as the downstream inverter bridge arm is connected in parallel across the bus capacitor as an active reactive and active control unit.

[0035] The control unit drives the inverter arm power switch to feed specific reactive or active charges to the bus capacitor. It uses the dynamic charging and discharging effect across the bus capacitor to raise or lower the DC bus voltage in real time, forcibly fitting the sinusoidal envelope waveform generated by the external AC grid after rectification. When the DC bus voltage is adjusted in the time domain to be exactly equal to the transient amplitude and synchronized with the phase of the AC rectified voltage, the voltage across the high-power current-limiting resistor connected in series between the two returns to zero. This also makes the normally open contact group connected in parallel across the current-limiting resistor reach a zero potential difference state, eliminating the discontinuity problem of cross-topology control.

[0036] While the inverter bridge arm maintains the damping pre-modulation state, the control unit uses the physical thermistor NTC2 located around the DC bus to collect the line operating temperature in real time, and calculates the thermistor impedance drift of the high-power current-limiting resistor R245 during the start-up charging process based on the line operating temperature. Furthermore, the upper limit of the switching threshold for the low-stress, arc-free switching voltage window is dynamically tightened as the line operating temperature increases. Switch threshold upper limit Satisfying the formula: ,in, This is the upper limit of the tightened switching threshold. The switching voltage reference constant ranges from 3V to 8V; This is the temperature correction factor. For the thermistor impedance drift of the high-power current-limiting resistor R245, the aforementioned control unit has a pre-stored impedance temperature mapping lookup table or a pre-set linear conversion coefficient based on the temperature coefficient of the resistor material. The quantization conversion process of the aforementioned thermistor impedance drift is as follows: the control unit reads the voltage signal of the aforementioned physical thermistor in real time through the analog-to-digital conversion channel and converts it into the absolute value of the current line operating temperature.

[0037] The control unit calculates the temperature difference between the absolute value of the circuit's operating temperature and the standard ambient temperature of 25°C. The control unit multiplies this temperature difference by a preset temperature coefficient for the high-power current-limiting resistor, which is set to 0.004 per°C in this embodiment. The result is the thermistor impedance drift of the high-power current-limiting resistor during the power-on charging process. If the operating temperature increases, the impedance drift is positive, indicating an increase in actual resistance. Based on this, the control unit dynamically tightens and adjusts the switching window to ensure high-precision locking of the switching point. The control unit internally stores the inherent mechanical delay time of the power relay from coil excitation to the complete closure of the normally open contact group. During the damping pre-modulation state, the digital signal processor utilizes a phase-locked loop algorithm. The provided time axis uses the delay time as a feedforward shift to calculate the voltage difference and loop current trajectory at the future contact closing moment. When the absolute value of the transient voltage difference at the current moment is detected to be less than or equal to the upper limit of the switching threshold for 3 consecutive milliseconds, and the absolute value of the loop current flowing through the high-power current-limiting resistor at the intersection after the delay time is predicted to fall into the zero-current bias window of 0.05 amps, the control unit sends a high-level soft-start control signal in advance to inject into the base of the preceding logic switch. This offsets the mechanical lag of the power relay, so that the closing moment when the normally open contact group makes physical contact is completely aligned with the zero potential difference switching point and zero current window of the transient trajectory of the DC bus voltage and the time-varying envelope of the AC rectified voltage in the time domain.

[0038] When the control unit detects that the transient voltage difference across the normally open contact group remains at a condition that is met... When the low-stress, arc-free switching voltage window reaches 3ms and the external AC grid voltage fluctuation rate gradient meets the safety slope threshold, the control unit captures the zero-crossing point of the transient current flowing through the high-power current-limiting resistor R245 by monitoring the transient current waveform. At the moment when the transient current zero-crossing point intersects with the dual voltage and current window, the soft-start control signal BUSSOFT.ON-D is switched from low to high to inject into the base of the preceding logic switch transistor Q43, driving... Transistor Q43 turns on to pull down the base potential of transistor Q41, causing transistor Q41, whose emitter is connected to the +12V_RELAY drive power supply, to turn on and pull up the gate potential of MOSFET power switch Q38. This controls the DC 12V coil to be excited and energized, driving the contacts of power relay RY4 to produce a mechanical action, causing the normally open contact group to turn on and close, disconnecting the rectification limiting path formed by the high-power current-limiting resistor R245 and the anti-reverse-feedback diode D42, and directly connecting the DC bus to the main power supply path.

[0039] After the normally open contact group is turned on to connect the main power supply path, the control unit continues to send the damping pre-modulation signal for 5ms. After the physical contacts of the normally open contact group complete the closing action and pass the mechanical bounce stabilization delay, the control unit stops sending the damping pre-modulation signal, and the drive control logic directly switches to the conventional power inverter pulse width modulation state. At the transient intersection point of the normally open contact group action, the DC bus voltage trajectory generates dynamic offset with the AC rectified voltage envelope through active damping deformation. The potential difference across the contact converges to a lower voltage difference value. At the moment of contact closure, there is no physical medium to maintain arc discharge. The system eliminates contact arc erosion damage, bus voltage overshoot, and surge current caused by potential mismatch during switching transients, achieving low electrical stress arc-free switching and increasing the mechanical and electrical life of switching devices. The cross-time-domain coupling structure between the AC / DC energy conversion channel switching logic and the active current throughput capability of the subsequent inverter network transforms the inherent fixed hardware boundary of the passive topology into an envelope tracking process dominated by a dynamic feedback control loop. This maintains the system's transient electrical protection characteristics and long-term thermal stability under dynamic changes in multiple energy inputs while reducing the hardware overhead of dedicated auxiliary power supplies.

[0040] Example 2: When the system faces the start-up impact and switching stress verification of the AC converter under complex distorted power grid conditions, the physical experimental platform used for this verification utilizes a programmable AC power supply capable of simulating grid-side voltage dips and harmonic injection. The power supply is connected to the input terminal CN20, and a charging path is established to the DC bus and bus capacitor C28 via a dual-diode rectification path composed of the first rectifier diode D43 and the second rectifier diode D44, and a high-power current-limiting resistor R245. An external dynamic temperature control box is arranged around the high-power current-limiting resistor R245 to simulate the high-temperature heat accumulation environment caused by continuous charging and discharging. The electrical measuring instrument used to capture the dynamic characteristics of the system has a voltage measurement range of 0V to 1000V, a quantization resolution of 0.05V, and a real-time sampling rate of 100kHz. At the same time, a current sensor with a core operating bandwidth of 150MHz is used to dynamically capture the contact current waveform of the power relay RY4 in the main power supply path. The sampling window length used for digital filtering and envelope tracking within the control unit is determined. At this time, it is controlled by the highest order of the grid-side harmonics and the floating-point operation overhead of the control chip. The technical considerations for its setting are to achieve a balance between the real-time response performance of signal acquisition and the data processing load of the system. When the proportion of high-frequency distortion components in the external input grid increases and its highest harmonic frequency shifts upward, in order to avoid signal aliasing under the Nyquist sampling theorem and accurately lock the time-varying sinusoidal potential, the sampling window length is... By narrowing the range to its lower limit, and applying this deterministic rule to a specific operating condition where the external power grid contains typical group harmonics of the fifth and seventh orders, the sampling window length is calculated as an unrestricted instance. The number of data points is set to 128.

[0041] To verify the applicability and anti-interference specificity of the control strategy in a noisy industrial environment, Gaussian white noise with a signal-to-noise ratio of 25dB was actively superimposed onto the input voltage waveform as an environmental disturbance source. The startup process was initiated under gradient temperature rise conditions with initial temperatures of 25.3℃, 60.5℃, 85.1℃, and 120.4℃ for the high-power current-limiting resistor R245. In the control group where the inverter bridge arm power switch driver modulation was not enabled, due to the DC bus voltage... It only produces a monotonically smooth passive exponential increase, the trajectory of which cannot fit the AC rectified voltage. The sinusoidal envelope of the DC bus voltage When the transition synchronization level reference is reached and the power relay RY4 is triggered, the transient voltage difference between the two ends before the normally open contact group closes is measured to be at a high level of 45.2V, 62.7V, 78.4V, and 91.3V respectively under the aforementioned temperature gradient. At the instant the physical contacts close, a transient induced voltage is released by the parasitic inductance. The measured arc duration between the physical contacts reaches 4.15ms, 5.82ms, 7.66ms, and 9.34ms respectively, and transient surge currents with peak values ​​of 38.6A, 45.2A, 51.7A, and 59.1A are generated in the main power supply path. Correspondingly, in the test group running the complete power control strategy, when the DC bus voltage... After the signal rises to the transition synchronization level reference and enters the transition synchronization phase, the control unit activates high-frequency micro-amplitude damping modulation and calculates the transient control duty cycle of the transition modulation signal according to the formula. Under the same noise crosstalk and temperature gradient conditions, a controllable active current is generated across the bus capacitor C28 to correct the DC bus voltage. The rising trajectory was forced to generate an AC rectified voltage just before the contacts closed at operating temperatures of 25.3℃, 60.5℃, 85.1℃, and 120.4℃. The nonlinear dip in the direction of the envelope trough causes the transient voltage difference between the two ends measured before the normally open contact group closes to be lowered and stabilized at 1.15V, 1.68V, 2.24V, and 2.95V respectively in the above temperature rise sequence, thereby controlling the physical voltage difference within the low-stress, arc-free switching voltage window. The transient control duty cycle of the transition modulation signal... Satisfying the formula: ,in, The transient control duty cycle for the transition modulation signal. This is the voltage feedback gain coefficient, with a value ranging from 0.15 to 0.35; This is the DC bus voltage. To ensure the real-time tracking voltage at the AC rectified voltage potential, in the actual software control logic, to ensure that the transient control duty cycle calculated by the aforementioned formula always conforms to the physical boundaries of the power electronic devices, the control unit enforces a closed-loop bottom-line limiting constraint mechanism after calculation. When in the early stage of the transition synchronization phase, if the algebraic subtraction result is negative due to the low DC bus voltage and the AC rectified voltage being at the peak of the sine wave, the limiter inside the control unit will automatically take the absolute value of the calculation result, or directly force the negative result to be cleared to zero, so that the calculation reference of the duty cycle is locked at the physical lower bound of 0. Similarly, if the calculation result exceeds the maximum physical boundary, the limiter will forcibly truncate it and lock it at the soft limit upper bound of 0.95, thereby avoiding the logic deadlock caused by illegal negative duty cycles to the control system and ensuring the smooth and self-consistent switching action of the inverter arm pulse width modulation.

[0042] To clarify the engineering basis for the control boundaries of key variables and determine the performance inflection point, the voltage feedback gain coefficient in the transient control duty cycle calculation formula of the transition modulation signal was changed by wide-range, wide-gradient variations during testing. The value of is used to examine the control boundary, when the voltage feedback gain coefficient is ... When the bias is lowered to 0.10, the amplitude of the dynamic active damping current generated by the inverter arm switching transistors is insufficient, and the DC bus voltage... The transient trajectory could not generate sufficient nonlinear deformation, causing the transient voltage difference between the two ends of the normally open contact group to rebound and increase to 14.25V just before the contact group turned on. This fluctuation failed to fall within the capture range of the low-stress, arc-free switching voltage window. Furthermore, when the voltage feedback gain coefficient was increased... When the bias is increased to the upper limit of 0.45, the gain overload of the closed-loop feedback loop causes the DC bus voltage to rise. The deformation trajectory generates high-frequency divergent oscillations, causing the voltage fluctuation amplitude on both sides of the bus to surge to 28.6V, which not only hinders the tracking of the AC rectified voltage. The periodic time-varying envelope of the voltage feedback circuit, instead of preventing overvoltage protection from tripping, actually induced a secondary overvoltage protection trip in the control circuit, thus confirming the voltage feedback gain coefficient. Choosing a working range of 0.15 to 0.35 is an optimal working window that balances tracking accuracy and loop stability.

[0043] The NTC2 physical thermistor installed around the DC bus collects the line operating temperature in real time. The control unit calculates the thermistor impedance drift of the high-power current-limiting resistor R245 during the power-on charging process based on the line operating temperature. Furthermore, the upper limit of the switching threshold for the low-stress, arc-free switching voltage window is dynamically tightened as the line operating temperature increases. When the control unit detects that the transient voltage difference across the normally open contact group remains within the low-stress, arc-free switching voltage window for 3ms, and the external AC grid voltage variation rate gradient meets the safety slope threshold, the control unit captures the zero-crossing point of the transient current flowing through the high-power current-limiting resistor R245 by monitoring the transient current waveform. At the moment when the transient current zero-crossing point intersects with the dual voltage and current window, the control unit controls the soft-start control signal BUSSOFT.ON-D to flip from low to high to inject into the preceding logic switching transistor. The base of Q43 drives transistor Q43 to conduct, pulling down the base potential of transistor Q41. This causes transistor Q41, whose emitter is connected to the +12V_RELAY power supply, to conduct and pull up the gate potential of the MOSFET power switch Q38. This, in turn, excites the 12V DC coil, driving the contacts of power relay RY4 to mechanically actuate. This closes the normally open contact group, disconnecting the rectifier limiting path formed by the high-power current-limiting resistor R245 and the anti-reverse-feedback diode D42, directly connecting the DC bus to the main power supply path. The upper limit of the switching threshold is [not specified in the original text]. Satisfying the formula: ,in, This is the upper limit of the tightened switching threshold. The switching voltage reference constant has a value range of 3V to 8V; This is the temperature correction factor. The thermistor impedance drift of the high-power current-limiting resistor R245 is determined by the control unit maintaining the damping pre-modulation signal for 5ms after the normally open contact group is turned on to connect the main power supply path. After the physical contacts of the normally open contact group complete the closing action and pass the mechanical bounce stabilization delay, the control unit stops sending the damping pre-modulation signal, and the drive control logic directly switches to the conventional power inverter pulse width modulation state.

[0044] At the transient junction of the normally open contact group, the DC bus voltage trajectory dynamically offsets the AC rectified voltage envelope through active damping deformation. The potential difference across the contacts converges to a lower voltage difference. At the moment the contacts close, the physical medium conditions necessary to sustain arc discharge are absent, eliminating contact arc erosion damage, bus voltage overshoot, and surge current caused by potential mismatch during switching. Furthermore, at the moment the main power supply path is connected, high-precision current transformer monitoring across the entire range shows that the leakage current increment between physical contacts is less than 0.02A and no high-frequency electromagnetic radiation pulses are generated. Simultaneously, the peak voltage overshoot at the DC bus terminal drops from 415.2V in the control group to 5.3V, confirming that the inverter bridge... The control mechanism of dynamic tracking of the DC bus voltage trajectory by high-frequency micro-amplitude damping modulation can suppress the electrical and thermal stress of the AC / DC switching node below the physical damage threshold of the device. It solves the contact welding problem under the single mains power start-up without auxiliary switch hardware shunt conditions. The cross-time domain coupling structure of the AC / DC energy conversion channel switching logic and the active current throughput of the subsequent inverter network transforms the inherent fixed hardware boundary of the passive topology into an envelope tracking process dominated by the dynamic feedback control loop. It maintains the transient electrical defense characteristics and long-term thermal stability of the system under the rheological boundary of multiple energy inputs while reducing the hardware overhead of the dedicated auxiliary power supply.

[0045] Example 3: This example combines Figures 1 to 2 This describes a drive control strategy method for an AC converter, such as... Figure 1 As shown, in step S1, when the AC converter is in a battery-free and single AC input power-on state, and the external input voltage meets the power-on threshold, the external input voltage is introduced through the dual diode rectification path of the input terminal and the high-power current-limiting resistor to charge the DC bus, and the photovoltaic auxiliary power supply path is simultaneously connected to complete the control unit startup; in step S2, when the DC bus voltage rises to the transition synchronization level reference, a damping pre-modulation signal is sent to control the inverter bridge arm to enter the damping pre-modulation state. By adjusting the drive modulation duty cycle of the inverter bridge arm in real time, the transient signals generated at both ends of the normally open contact group of the power relay on the main power supply path are controlled. The voltage difference is adjusted to be within the low-stress, arc-free switching voltage window. In step S3, when the transient voltage difference remains within the low-stress, arc-free switching voltage window and the external grid voltage variation rate gradient meets the safety slope threshold, the drive circuit of the control power relay is turned on to make the normally open contact group close, cutting off the rectifier limiting path formed by the high-power current-limiting resistor, and directly connecting the DC bus to the main power supply path. In step S4, after the normally open contact group completes the closing action and after the contact mechanical bounce stabilization delay, the damping pre-modulation signal is stopped, and the drive control logic smoothly switches to the normal power inverter pulse width modulation state to complete the safe start-up.

[0046] like Figure 2As shown, the graph displays time (t / ms) on the horizontal axis, with calibration scales including 0, 20, 40, 60, 80, and 100, and voltage (V / V) on the vertical axis, with calibration scales including 0, 100, 200, 300, and 400. It includes the AC rectified voltage envelope. Modulation bus voltage not turned on Control group and active damping modulated bus voltage The test group did not activate the modulation bus voltage. The control group reflects the voltage state under passive current-limited charging conditions, while the active damping modulation bus voltage is activated. The test group reported that during the transition synchronization phase, active damping drive control enabled the DC bus voltage to track the AC rectified voltage envelope. The time-varying envelope state causes the transient voltage difference generated across the normally open contact group of the power relay to converge to the low-stress, arc-free switching voltage window.

[0047] Example 4: When the system faces extreme conditions such as a sudden change in the harmonic distortion rate of the external AC power grid and an increase in ambient operating temperature to 65°C, the high-power current-limiting resistor R245 experiences passive power loss and local heat accumulation under continuous full-load operation, causing its resistance value to deviate from its initial state at room temperature and resulting in thermistor drift. Simultaneously, the rapid phase drift caused by high-frequency harmonic components in the sinusoidal waveform of the grid-side voltage causes the DC bus voltage to... With AC rectified voltage The transient voltage difference between them generates high-frequency, violent oscillations, which in turn causes misjudgment and misalignment of the normally triggered logic in the control circuit and mechanical jitter of the contacts of the power relay RY4. During the transition synchronization phase, the digital filter inside the control unit initiates a sliding time window containing discrete sampling periods, at the current time-domain sampling point... The DC bus voltage introduced via input terminal CN20 and high-power current-limiting resistor R245 is read. Sequence value and transient AC rectified voltage Sequence value; the transient voltage difference at the current moment is calculated using a differential processor. The gradient of the external grid voltage variation rate, which characterizes the intensity of grid harmonic distortion variation, is obtained through first-order discrete difference operations. external power grid voltage variation rate gradient Satisfying the formula: ,in, The gradient of the external power grid voltage variation rate. This represents the transient voltage difference at the current sampling time. This represents the transient voltage difference at the previous sampling time. For the discrete sampling period of the digital filter, the control unit will use the gradient of the external grid voltage variation rate. The voltage slope threshold is compared with a safety slope threshold preset in non-volatile memory. In this process, the aforementioned safety slope threshold preset in non-volatile memory is set to a specific quantized value, specifically calibrated to 1500 volts per second in this embodiment. The engineering basis for this technical indicator is that when the external power grid is in a state of normal fluctuation or contains low-order harmonics within a standard range, the calculated absolute value of the external power grid voltage variation rate gradient will not exceed this threshold. However, once the external power grid experiences a momentary voltage drop or a large-scale high-frequency harmonic distortion, the transient slope of the sine wave will change drastically, causing the partial derivative to surge rapidly and exceed 1500 volts per second. The control unit identifies the abnormal power grid condition based on this quantized indicator and then adaptively activates the subsequent blocking logic. When the external power grid voltage variation rate gradient is detected... When the voltage is within the safety slope threshold, the control unit outputs a transition modulation signal by calculating the differential proportional adjustment amount, controlling the inverter arm to generate symmetrical active current throughput during the transition synchronization phase to correct the DC bus voltage change trajectory; conversely, if the external grid voltage fluctuation rate gradient... If the voltage slope threshold is crossed, it indicates that the grid-side harmonic distortion has caused a sudden change in the voltage trajectory. The control unit forcibly blocks the level flip-over enable of the soft-start control signal BUSSOFT.ON-D through the enable latch, and controls the inverter bridge arm to continue to maintain the high-frequency micro-amplitude damped modulation state.

[0048] During the inverter bridge arm's damped pre-modulation state, the high-power current-limiting resistor R245 experiences resistance changes due to increased operating temperature. The control unit periodically acquires the line operating temperature sequence using the physical thermistor NTC2 located around the DC bus. Within specific time windows when the system is under no-load or in a quiescent state, a preset low-amplitude micro-frequency pulse excitation is applied to the high-power current-limiting resistor R245, and the transient micro-amplitude current of the return flow is read. This process corrects the initial impedance reference and aging attenuation factor, and calculates the thermistor impedance drift under the current temperature rise condition. The control register will be used to lock the upper limit of the switching threshold for the switching window. The convergence threshold is dynamically adjusted to fit the current thermal stress boundary, wherein the upper limit of the switching threshold is... Satisfying the formula: ,in, This is the upper limit of the tightened switching threshold. To switch the voltage reference constant, This is the temperature correction factor. To account for the thermistor impedance drift of the high-power current-limiting resistor R245, in actual operation, the aforementioned operation of applying low-amplitude micro-frequency pulse excitation to correct the initial impedance reference and aging attenuation factor is limited to the resting disconnection phase before the AC converter is first powered on, thereby completing the reference calibration for the aging of passive components; once the system enters the dynamic non-resting condition of power-on charging, in order to avoid the superposition interference between the pulse excitation and the high-current charging waveform, the control unit automatically cuts off the pulse excitation transmission channel and fully switches to the real-time traceability mode based on temperature acquisition.

[0049] During dynamic charging, the control unit relies solely on the circuit operating temperature sequence periodically collected by the aforementioned physical thermistor. Through pure software deduction using the temperature difference and dynamic impedance thermal coefficient, it achieves disturbance-free, high-real-time online dynamic updating of thermistor drift within the high-current interference blind zone of continuous charging and discharging. This ensures the temporal continuity of state switching and physical safety boundaries. When the transient voltage difference measured by the digital filter remains within the specified conditions... The low-stress, arc-free switching voltage window reaches the 3ms threshold, and the external grid voltage variation rate gradient... When the current returns to below the safe slope threshold, the microcontroller capture register inside the control unit monitors the transient current waveform flowing through the high-power current-limiting resistor R245 in real time via a high-speed comparator and captures its physical current zero-crossing point. At the moment when the physical current zero-crossing point intersects with the dual voltage-current window, the soft-start control signal BUSSOFT.ON-D flips from low to high to inject into the base of the preceding logic switch transistor Q43, driving transistor Q43 to conduct and pull down the base potential of transistor Q41, connecting its emitter to the +12V drive power supply. The transistor Q41 of V_RELAY turns on and pulls up the gate potential of the MOSFET power switch Q38, thereby controlling the DC 12V coil to be energized and driven to mechanically actuate the contacts of the power relay RY4. This causes the normally open contact group to close, disconnecting the rectifier limiting path formed by the high-power current-limiting resistor R245 and the anti-reverse-feedback diode D42, and directly connecting the DC bus to the main power supply path. In actual operation, the specific determination architecture of the low-stress arc-free switching voltage window in the physical control circuit is as follows: the control unit determines the voltage window through its internal independent analog-to-digital converter. The system switches channels and synchronously reads the digital values ​​of the DC bus voltage and AC rectified voltage within each 10μs discrete sampling period. These are subtracted by an internal digital subtractor, and the absolute value is taken to obtain the absolute value of the actual voltage difference across the current contact. Simultaneously, a Hall current sensor deployed on the main power supply path acquires the absolute value of the actual current flowing through the high-power current-limiting resistor in real time. The control unit continuously compares the calculated absolute value of the actual voltage difference with the currently calculated tightened switching threshold upper limit. If the absolute value of the actual voltage difference is less than or equal to the upper limit of the switching threshold, the internal voltage flag register is written to 1; otherwise, it is written to 0. Similarly, if the absolute value of the actual current is less than or equal to 0.05A, the internal current flag register is written to 1; otherwise, it is written to 0. The digital signal processor performs a bitwise logical AND operation on the states of the voltage and current flag registers. When the operation result remains continuously at 1 for 3ms and the external power grid voltage fluctuation rate gradient is below the safety slope threshold, the electrical environment window for physical contact closure is deemed fully established, thus triggering the subsequent level flip signal.

[0050] After the normally open contact group is turned on to connect the main power supply path, the control unit continues to send the damping pre-modulation signal for 5ms. After the physical contacts of the normally open contact group complete the closing action and pass the mechanical bounce stabilization delay, the control unit stops sending the damping pre-modulation signal, and the drive control logic directly switches to the conventional power inverter pulse width modulation state. At the transient junction of the normally open contact group, the DC bus voltage trajectory generates dynamic offset with the AC rectified voltage envelope through active damping deformation. The potential difference across the contact converges to a lower voltage difference. At the moment of contact closure, there is no physical medium condition to maintain arc discharge, eliminating contact arc erosion damage, bus voltage overshoot, and surge current caused by potential mismatch during switching. Moreover, at the moment the main power supply path is connected, the high-precision current transformer passes through the full range of the circuit. Monitoring revealed that the leakage current increment between physical contacts was less than 0.02A and no high-frequency electromagnetic radiation pulses were generated. At the same time, the peak voltage overshoot at the DC bus terminal decreased from 415.2V in the control group to 5.3V. This controlled the electrical and thermal stress of the AC / DC switching node to be below the physical damage threshold of the device, solving the contact welding technology problem under single mains power start-up without auxiliary switch hardware shunt conditions. The cross-time domain coupling structure of the AC / DC energy conversion channel switching logic and the active current throughput capability of the subsequent inverter network transforms the inherent fixed hardware boundary of the passive topology into an envelope tracking process dominated by the dynamic feedback control loop. While simplifying the hardware overhead of the dedicated auxiliary power supply, it maintains the transient electrical defense characteristics and long-term thermal stability of the system under the rheological boundary of multiple energy inputs.

[0051] Example 5: When the system faces the initial environmental calibration operation during field deployment, the control unit runs a standardized pre-test calibration program before the main power supply path is powered on to eliminate baseline offset caused by differences in the acquisition components; when the line is in a resting disconnected state with zero input current, the control unit reads the DC bus voltage through the internal analog-to-digital converter port. With real-time tracking voltage The zero-point drift difference is defined as the baseline calibration bias. Then, the baseline calibration bias is used. Correct the switching voltage reference constant in the control register. The corrected switching voltage reference constant Satisfying the linear compensation formula: ,in, This is the corrected switching voltage reference constant. Preset voltage constant This is the baseline calibration bias.

[0052] During the long-term continuous operation of the converter, the control unit periodically calls the pre-test calibration program repeatedly in the no-load start-up window, and updates the baseline data by introducing a sliding time window, thereby switching the voltage reference constant. It can adaptively track the time-varying trend of parameters caused by device aging.

[0053] When the switching voltage reference constant is detected Within the specified safety boundaries, and the baseline calibration bias... When the rate of change converges to a preset threshold, the control unit releases the enable latch from its locked state, and the control system enters the damping pre-modulation control program, so that the transient voltage difference at the moment of subsequent normally open contact group closure can converge to the upper limit of the tightened switching threshold. Within the specified range, the closed-loop control instability problem caused by unknown initial values ​​of passive component parameters is eliminated, enabling the converter output node to maintain a safe switching steady state with long-term electrical adaptability.

[0054] Example 6: When the system faces initial zero-point drift caused by hardware batch differences and component aging, in the resting disconnected state before the AC converter is connected to the main power supply circuit, the control unit collects the DC bus voltage through the analog-to-digital conversion port. With AC rectified voltage The initial sampled value at the potential is used by the control unit to calculate the arithmetic mean of the DC bus voltage sampling signal and the AC rectified voltage sampling signal over 100 sampling points, and the difference between the two is calculated to obtain the baseline calibration bias. The control unit uses baseline calibration bias. Correct the preset voltage constant in the internal register. Determine the corrected switching voltage reference constant. Switching voltage reference constant Satisfy the calculation formula: ;in, This is the corrected switching voltage reference constant. As a preset voltage constant, This is the baseline calibration bias.

[0055] During the first test condition of the system's pre-calibration, the control unit sets the switching voltage reference constant. Set the voltage feedback gain coefficient to 3V. The time span of the transition synchronization phase is set to 20ms, with a value of 0.15. The control unit controls the inverter bridge arm to operate at twice the normal inverter operating frequency, causing the bus capacitor to generate high-frequency micro-amplitude damping modulation and construct a current throughput loop. Before the normally open contact group of the high-power current-limiting resistor on the main power supply path closes, the transient voltage difference generated across the contact group converges to 2.95V, and the transient current flowing through the high-power current-limiting resistor is 0.15A. Under the second test condition of the system pre-calibration, the control unit sets the switching voltage reference constant. Set the voltage feedback gain coefficient to 5.5V. The value is 0.25. The time span of the transition and synchronization phase is set to 35ms. The control unit controls the inverter bridge arm to send a damping pre-modulation signal and adjusts the drive modulation duty cycle of the inverter bridge arm so that the DC bus voltage trajectory changes with the time-varying envelope of the AC rectified voltage. It is measured that before the normally open contact group of the high-power current limiting resistor closes, the transient voltage difference generated across the contact group converges to 1.68V, and the transient current flowing through the high-power current limiting resistor is 0.08A.

[0056] In the third test condition for system pre-calibration, the control unit sets the switching voltage reference constant. Set the voltage feedback gain coefficient to 8V. The value was set to 0.35, and the time span of the transition synchronization phase was set to 50ms. The control unit controlled the inverter bridge arm to increase the dynamic potential adjustment amplitude at the bus end. It was measured that the transient voltage difference generated across the contact group of the high-power current-limiting resistor converged to 1.15V just before the normally open contact group of the high-power current-limiting resistor closed, and the transient current flowing through the high-power current-limiting resistor was 0.02A. Since the high-power current-limiting resistor experienced thermistor drift due to thermal effects during continuous operation, the control unit introduced a temperature correction coefficient. When the circuit operating temperature collected by the temperature sensor changes, the control unit determines the temperature based on the thermistor impedance drift of the high-power current-limiting resistor at the current circuit operating temperature. Dynamically tighten the upper limit of the switching threshold for low-stress, arc-free switching voltage windows. Switch threshold upper limit Satisfy the calculation formula: ;in, This is the upper limit of the tightened switching threshold. This is the corrected switching voltage reference constant. This is the temperature correction factor. To measure the thermistor impedance drift of the high-power current-limiting resistor at the current line operating temperature, during the transition synchronization phase, the control unit collects the transient voltage digital signal sequence across the normally open contact group of the high-power current-limiting resistor and the main power supply path loop current digital signal sequence in real time. The control unit then converts the DC bus voltage... With AC rectified voltage The absolute difference of transient voltage between the two and the upper limit of the switching threshold In comparison, the absolute value of the loop current is simultaneously compared with the preset zero-current bias window of 0.05A. When the transient voltage meets the voltage coordination condition... And this continues for 3ms. At the same time, when the loop current crosses zero and enters the zero current bias window, the control signal level of the control logic output port of the control unit changes from low to high. The high-level signal of the logic output port is injected into the base of the front-stage logic switch to drive the front-stage logic switch to conduct, so that the DC 12V coil is excited and attracted. The normally open contact group of the high-power current limiting resistor operates and cuts off the rectification limiting path, directly connecting the DC bus and the main power supply path. After the main power supply path is connected, the control unit continues to send the damping pre-modulation signal for 5ms. After the normally open contact group of the high-power current limiting resistor crosses the mechanical bounce delay and the contact state stabilizes, the control unit turns off the damping pre-modulation signal and drives the control logic to switch to the conventional power inverter pulse width modulation state. The converter enters the inverter power supply steady state.

[0057] It will be apparent to those skilled in the art that the present invention is not limited to the details of the exemplary embodiments described above, and that the present invention can be implemented in other specific forms without departing from the spirit or essential characteristics of the present invention.

[0058] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit it. Although the present invention has been described in detail with reference to preferred embodiments, those skilled in the art should understand that modifications or equivalent substitutions can be made to the technical solutions of the present invention without departing from the spirit and scope of the technical solutions of the present invention.

Claims

1. A drive control strategy method for an AC converter, characterized in that, Includes the following steps: Step S1: When the AC converter is in a state of no battery and single AC input, and the external input voltage meets the start-up threshold, the external input voltage is introduced by using the dual diode rectification path of the input terminal and the high-power current limiting resistor to charge the DC bus, and the photovoltaic auxiliary power supply path is connected simultaneously to complete the start-up of the control unit. Step S2: When the DC bus voltage rises to the transition synchronization level reference, a damping pre-modulation signal is sent to control the inverter bridge arm to enter the damping pre-modulation state. By adjusting the drive modulation duty cycle of the inverter bridge arm in real time, the transient voltage difference generated across the normally open contact group of the power relay on the main power supply path is adjusted to within the low-stress arc-free switching voltage window. Step S3: When the transient voltage difference remains within the low-stress arc-free switching voltage window and the external grid voltage variation rate gradient meets the safety slope threshold, the drive circuit of the control power relay is turned on to make the normally open contact group close, cut off the rectifier limiting path composed of the high-power current limiting resistor, and directly connect the DC bus to the main power supply path. Step S4: After the normally open contact group completes the engagement action and the contact mechanical bounce stabilizes and delays, the damping pre-modulation signal is stopped, and the drive control logic smoothly switches to the normal power inverter pulse width modulation state to complete the safe start-up.

2. The drive control strategy method for an AC converter according to claim 1, characterized in that, Before the normally open contact group is closed in step S3, the digital filter inside the control unit continuously calculates the voltage variation rate gradient of the external power grid. When the voltage drop or high-frequency harmonic distortion of the external AC power grid causes the voltage variation rate gradient of the external power grid to exceed the safety slope threshold, the control unit activates the blocking logic, cuts off the level flip enable of the control signal, keeps the drive circuit in a static state without operation, and the inverter bridge arm continues to maintain the damped pre-modulation state.

3. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S2, when the transient voltage difference generated across the normally open contact group is adjusted to within the low-stress arc-free switching voltage window, the line operating temperature is collected in real time using a physical thermistor located around the DC bus. The control unit calculates the thermistor impedance drift of the high-power current-limiting resistor during the power-on charging process based on the line operating temperature, and dynamically tightens the upper limit of the switching threshold of the low-stress arc-free switching voltage window as the line operating temperature increases. The upper limit of the switching threshold satisfies the formula: ,in, This is the upper limit of the tightened switching threshold. To switch the voltage reference constant, This is the temperature correction factor. This represents the thermistor drift.

4. The drive control strategy method for an AC converter according to claim 3, characterized in that, When the line operating temperature increases and the upper limit of the switching threshold of the low-stress arc-free switching voltage window is tightened, the control unit synchronously extends the upper limit of the transition pre-modulation time in step S2 to maintain the damping pre-modulation state, and corrects the overcurrent deviation caused by the thermistor impedance drift of the high-power current-limiting resistor.

5. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S2, controlling the inverter bridge arm to enter the damping pre-modulation state includes the following sub-steps: Step S21, obtain the real-time voltage phase and current frequency of the external AC power grid; Step S22, increase the switching frequency of the inverter bridge arm to twice the operating frequency of the conventional inverter, and send a damping pre-modulation signal to control the inverter bridge arm to output a controlled AC voltage with the same frequency, phase and amplitude as the external AC power grid, so that the two ends of the normally open contact group of the power relay are at the zero potential difference switching point.

6. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S3, the transient voltage difference being within the low-stress, arc-free switching voltage window refers to the moment when the control unit detects the intersection of the transient voltage across the normally open contact group and the main power supply path loop current simultaneously falling into the set dual voltage and current window.

7. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S4, after the normally open contact group completes the closing action and the mechanical bounce of the contact is stabilized and delayed, the control unit maintains the damping pre-modulation signal for 5ms after the normally open contact group completes the closing action. After the mechanical bounce of the normally open contact group is eliminated and the contact state is stabilized, the drive control logic is smoothly switched to the conventional power inverter pulse width modulation state.

8. The drive control strategy method for an AC converter according to claim 2, characterized in that, After the control unit activates the latching logic and keeps the drive circuit in a dormant state, the digital filter continuously monitors the voltage fluctuations of the external AC power grid. When the voltage fluctuations of the external AC power grid are eliminated and the calculated external power grid voltage variation rate gradient recovers to below the safety slope threshold, the control unit automatically closes the latching logic, restores the level flip enable of the control signal, and allows the normally open contact group to close again.

9. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S1, an external input voltage is introduced through the dual diode rectifier circuit of the input terminal and the high-power current-limiting resistor, eliminating the need to set up a separate AC auxiliary power supply for the main circuit. The dual diode rectifier circuit of the input terminal includes a first rectifier diode and a second rectifier diode, and the high-power current-limiting resistor is connected between the output terminal of the dual diode rectifier circuit and the DC bus.

10. The drive control strategy method for an AC converter according to claim 1, characterized in that, In step S3, the drive circuit of the control power relay is turned on to make the normally open contact group close. This is achieved by energizing the DC 12V coil to drive the contacts of the power relay to operate, switching the rectifier limiting path to the normally open power supply state, and cutting off the rectifier limiting path.

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