Control method of an inverter

Abstract

This invention discloses a control method for an inverter. The method includes: obtaining a cost function; the cost function includes a first influence factor and a second influence factor, the first influence factor being determined based on the deviation between the predicted voltage value of each flying capacitor at the next time step and the reference voltage value at the next time step; the predicted voltage value of each flying capacitor at the next time step being related to the current switching state combination; the switching state combination including the switching states of each first switch; the second influence factor being determined based on the switching states of each first switch at the previous time step and the current switching state; at the current time step, calculating the cost function values ​​of multiple switching state combinations, and taking the switching state combination corresponding to the minimum cost function value as the target switching state combination at the current time step. As the inverter operates, under the control of the target switching state combination at each time step, the voltage of the flying capacitor remains near the reference voltage value, thereby achieving voltage balance of the flying capacitor.

Description

Technical Field

[0001] This invention relates to the field of power electronic converter control technology, and more particularly to a control method for inverters. Background Technology

[0002] The flying capacitor five-level inverter has the advantages of uniform voltage stress distribution of switching devices and high output waveform quality, and has broad application prospects in medium and high voltage and high power applications such as industrial transmission and renewable energy power generation.

[0003] Under high-power operation conditions, the inverter's switching frequency typically needs to be controlled within a low range to effectively reduce system switching losses. Specific harmonic cancellation modulation techniques can achieve high-quality output voltage waveforms at low switching frequencies. Applying these techniques to five-level inverters with flying capacitors significantly enhances the competitiveness of this topology in high-power applications. However, this advantage is highly dependent on the balance of the flying capacitor voltage. If the flying capacitor voltage becomes unbalanced, it will not only cause output voltage waveform distortion but also lead to uneven voltage stress distribution among power switching devices, severely impacting system performance and long-term reliability.

[0004] Currently, there is a lack of sufficiently effective control strategies to address the capacitor voltage balance problem in five-level inverters with flying capacitors under specific harmonic cancellation modulation. Summary of the Invention

[0005] This invention provides a control method for an inverter to solve the problem of voltage imbalance in a flyback capacitor.

[0006] According to one aspect of the present invention, a control method for an inverter is provided, applied to a five-level inverter with a flying capacitor. The inverter includes a plurality of first switching transistors connected in series between the positive terminal of a DC power supply and an intermediate node, and a plurality of second switching transistors connected in series between the negative terminal of a DC power supply and the intermediate node. The second switching transistors correspond one-to-one with the first switching transistors. A flying capacitor is connected between the connection node of each pair of adjacent first switching transistors and the connection node of each pair of adjacent second switching transistors. The first switching transistors and the corresponding second switching transistors are complementary in conduction. The method includes: Obtain a cost function; wherein the cost function includes a first influence factor and a second influence factor, the first influence factor being determined based on the deviation between the predicted voltage value of each flying capacitor at the next time step and the reference voltage value at the next time step; the predicted voltage value of each flying capacitor at the next time step is related to the current switching state combination; the switching state combination includes the switching state of each of the first switching transistors; the second influence factor being determined based on the switching state of each of the first switching transistors at the previous time step and the switching state at the current time step; At the current moment, the cost function value of multiple switch state combinations is calculated, and the switch state combination corresponding to the minimum cost function value is taken as the target switch state combination at the current moment. The operation of each first switch and each second switch is controlled according to the target switch state combination.

[0007] Optionally, before obtaining the cost function, the method further includes: Based on the backward Euler method, the voltage values ​​of each flying capacitor at the next time step are obtained according to the dynamic equation of each flying capacitor and the voltage values ​​of each flying capacitor sampled at the current time step.

[0008] Optionally, before obtaining the cost function, the method further includes: The total harmonic distortion (THD) function of the inverter's output voltage is determined based on a function of a specific harmonic cancellation modulation waveform; the THD function of the inverter's output voltage is related to each level switching angle; the specific harmonic cancellation modulation waveform is the target waveform of the output voltage; An optimization function is constructed based at least on the total harmonic distortion rate function; The optimal set of level switching angles is solved with the goal of minimizing the optimization function, and the set of level switching angles includes each of the level switching angles. Based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment, determine the level signal of the specific harmonic cancellation modulation waveform at the current moment; Based on the level signal of the specific harmonic cancellation modulation waveform at the current moment, a preset switch state combination group is determined; wherein, the preset switch state combination group includes each switch state combination that makes the output voltage equal to the level signal of the specific harmonic cancellation modulation waveform at the current moment. At the current moment, calculating the cost function value for multiple combinations of switching states includes: At the current moment, calculate the cost function value for each switch state combination in the preset switch state combination group.

[0009] Optionally, the function for determining the total harmonic distortion rate of the inverter's output voltage based on a function of a specific harmonic cancellation modulation waveform includes: Obtain the fundamental amplitude function and the amplitude functions of each odd harmonic of the specific harmonic cancellation modulation waveform, wherein the fundamental amplitude function is related to each level switching angle, and the amplitude functions of each odd harmonic are related to each level switching angle; The total harmonic distortion rate function is constructed based on the fundamental amplitude function and the amplitude functions of each odd-order harmonic.

[0010] Optionally, the function of the specific harmonic cancellation modulation waveform is: ; n is an odd number. For the first level switching angle, This is the second level switching angle.

[0011] Optionally, before determining the output voltage at the current moment based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment, the method further includes: Based on droop control, a phase reference value for the specific harmonic cancellation modulation waveform at the current moment is generated according to the output voltage and output current of the inverter at the current moment.

[0012] Optionally, the step of generating the phase reference value of the specific harmonic cancellation modulation waveform at the current moment based on droop control and the output voltage and output current of the inverter includes: Calculate the active power of the inverter at the current moment based on the inverter's output voltage and output current. Based on the droop coefficient of the active power and the deviation between the active power of the inverter at the current moment and the active power reference value at the current moment, a phase reference value for the specific harmonic cancellation modulation waveform at the current moment is generated.

[0013] Optionally, before constructing the optimization function based at least on the total harmonic distortion rate function, the method further includes: Based on droop control, an amplitude reference value for the specific harmonic cancellation modulation waveform at the current moment is generated according to the output voltage and output current of the inverter at the current moment. A voltage deviation function is established based on the deviation between the fundamental amplitude function of the specific harmonic cancellation modulation waveform and the amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment; the fundamental amplitude function is related to each of the level switching angles; At least the optimization function constructed based on the total harmonic distortion rate function includes: The optimization function is constructed based on the voltage deviation function and the total harmonic distortion rate function.

[0014] Optionally, the step of generating the amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment based on droop control and the output voltage and output current of the inverter at the current moment includes: Calculate the reactive power of the inverter at the current moment based on the inverter's output voltage and output current. Based on the droop factor of reactive power and the deviation between the reactive power of the inverter at the current moment and the reactive power reference value at the current moment, an amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment is generated.

[0015] Optionally, the reactive power of the inverter at the current moment is calculated based on the inverter's output voltage and output current, including: The output voltage of the inverter at the current moment is input into the first second-order generalized integrator to obtain the quadrature component of the output voltage of the inverter at the current moment; the output current of the inverter at the current moment is input into the second second-order generalized integrator to obtain the quadrature component of the output current of the inverter at the current moment. The reactive power of the inverter at the current moment is generated based on the inverter's output voltage, output current, quadrature component of output voltage, and quadrature component of output current.

[0016] In this embodiment, a cost function is established based on the deviation between the predicted voltage value and the reference voltage value of each flying capacitor at the next moment. The switching state combination that minimizes the cost function value is then identified as the target switching state combination. This ensures that the voltage deviation between the flying capacitor and the reference voltage value is small, guaranteeing that as the inverter operates, the voltage of the flying capacitor remains near the reference voltage value under the control of the target switching state combination at each moment, thus achieving voltage balance of the flying capacitor. Simultaneously, the cost function also considers the number of switching actions, incorporating both the voltage error of the flying capacitor and the number of switching operations. This effectively suppresses unnecessary switching actions while achieving real-time capacitor voltage balance.

[0017] It should be understood that the description in this section is not intended to identify key or essential features of the embodiments of the present invention, nor is it intended to limit the scope of the invention. Other features of the invention will become readily apparent from the following description. Attached Figure Description

[0018] To more clearly illustrate the technical solutions in the embodiments of the present invention, the accompanying drawings used in the description of the embodiments will be briefly introduced below. Obviously, the accompanying drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0019] Figure 1 This is a schematic diagram of a five-level inverter with a flying capacitor provided in an embodiment of the present invention; Figure 2A flowchart of an inverter control method provided in an embodiment of the present invention; Figure 3 A flowchart of another inverter control method provided in an embodiment of the present invention; Figure 4 A flowchart of another inverter control method provided in an embodiment of the present invention; Figure 5 A steady-state simulation result diagram of an inverter provided in an embodiment of the present invention; Figure 6 A switching transistor driving waveform diagram provided in an embodiment of the present invention; Figure 7 This is a dynamic simulation diagram of the power level step of an inverter provided in an embodiment of the present invention. Detailed Implementation

[0020] To enable those skilled in the art to better understand the present invention, the technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings of the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort should fall within the scope of protection of the present invention.

[0021] It should be noted that the terms "first," "second," etc., in the specification, claims, and accompanying drawings of this invention are used to distinguish similar objects and are not necessarily used to describe a specific order or sequence. It should be understood that such data can be interchanged where appropriate so that the embodiments of the invention described herein can be implemented in orders other than those illustrated or described herein. Furthermore, the terms "comprising" and "having," and any variations thereof, are intended to cover non-exclusive inclusion; for example, a process, method, system, product, or apparatus that comprises a series of steps or units is not necessarily limited to those steps or units explicitly listed, but may include other steps or units not explicitly listed or inherent to such processes, methods, products, or apparatus.

[0022] This invention provides a control method for an inverter to ensure voltage balance among the flying capacitors in the inverter, especially when using specific harmonic cancellation modulation techniques to achieve high-quality output voltage waveforms at low switching frequencies. The inverter control method in this embodiment is applied to a five-level inverter with flying capacitors. Figure 1 This is a schematic diagram of a five-level inverter with a flying capacitor provided in an embodiment of the present invention, with reference to... Figure 1This flying capacitor five-level inverter includes multiple first switching transistors connected in series between the positive terminal IN+ of the DC power supply and intermediate node A, and multiple second switching transistors connected in series between the negative terminal IN- of the DC power supply and intermediate node A. Each second switching transistor corresponds one-to-one with a first switching transistor. A flying capacitor is connected between the connection node of each pair of adjacent first switching transistors and the connection node of each pair of adjacent second switching transistors. The first switching transistors and their corresponding second switching transistors are complementary in conduction. Complementary conduction can be understood as follows: when a first switching transistor is on, its corresponding second switching transistor is off; when a first switching transistor is off, its corresponding second switching transistor is on. In this embodiment, four first switching transistors are included, denoted as the first first switching transistor T1, the second first switching transistor T2, the third first switching transistor T3, and the fourth first switching transistor T4. These four transistors are connected in series between intermediate node A and the positive terminal IN+ of the DC power supply. The inverter includes four second switching transistors, denoted as T10, T20, T30, and T40. These transistors are connected in series between the intermediate node A and the negative terminal IN- of the DC power supply. The i-th first switching transistor corresponds to the i-th second switching transistor, where i = 1, 2, 3, or 4. The inverter also includes three flying capacitors, denoted as C1, C2, and C3. One end of the first flying capacitor C1 is connected to the connection node of the first switching transistor T1 and the second switching transistor T2, and the other end is connected to the connection node of the first second switching transistor T10 and the second second switching transistor T20. One end of the second flying capacitor C2 is connected to the connection node of the second first switch T2 and the third first switch T3, and the other end is connected to the connection node of the second second switch T20 and the third second switch T30. One end of the third flying capacitor C3 is connected to the connection node of the third first switch T3 and the fourth first switch T4, and the other end is connected to the connection node of the third second switch T30 and the fourth second switch T40. The inverter also includes a filter inductor Lf, a filter capacitor Cf, and a load. One end of the filter inductor Lf is connected to the intermediate node A, and the other end is connected to one end of the filter capacitor Cf and one end of the load, respectively. The other end of the filter capacitor Cf and the other end of the load are both grounded to GND. In this embodiment, the load is exemplarily shown as a resistor R. In this embodiment, a first bus capacitor C10 and a second bus capacitor C20 are connected in series between the positive terminal IN+ and the negative terminal IN- of the DC power supply, and the connection node of the first bus capacitor C10 and the second bus capacitor C20 is grounded to GND.

[0023] Figure 2 A flowchart of an inverter control method provided in an embodiment of the present invention is shown below. Figure 1 and Figure 2 The method includes: S110: Obtain the cost function. The cost function includes a first influence factor and a second influence factor. The first influence factor is determined based on the deviation between the predicted voltage value of each flying capacitor at the next time step and the reference voltage value at the next time step. The predicted voltage value of each flying capacitor at the next time step is related to the current switching state combination. The switching state combination includes the switching state of each first switch. The second influence factor is determined based on the switching state of each first switch at the previous time step and the switching state at the current time step.

[0024] The first impact factor f1 satisfies: d represents the total number of flying capacitors included in the inverter. This is the reference value for the voltage of the h-th flying capacitor at the next moment. Let be the predicted voltage value of the flying capacitor at the next moment in the h-th moment. Let be the balance weighting coefficient corresponding to the voltage of the h-th flying capacitor. The reference value of the voltage of the h-th flying capacitor can be considered to correspond to the same reference value at all times, which is equal to . Where E is the voltage of the DC power supply, h is an integer greater than or equal to 1 and less than or equal to d, and d is a positive integer. Time m represents the current time, and time m+1 represents the next time.

[0025] The second influence factor f2 satisfies: ; g represents the number of first-stage switching transistors included in the inverter. This represents the switching state of the i-th first switch at the previous moment. This represents the switching state of the i-th first switch at the current moment. represents the weighting coefficient for the switching frequency constraint. When the first switch is turned on, the corresponding switching state is "1", and when the first switch is turned off, the corresponding switching state is "0".

[0026] by Figure 1 Taking the inverter mentioned above as an example, specifically... Figure 1 In the inverter shown, the cost function satisfies: ; in, The voltage of the first flying capacitor C1 is respectively The balance weighting coefficient, the voltage of the second flying capacitor C2 The balance weighting coefficient, the voltage of the third flying capacitor C3 The balance weighting coefficients. Wherein, , , .

[0027] The first influencing factor considers the deviation between the predicted voltage value and the reference voltage value of each flying capacitor at the next moment, while the second influencing factor considers the number of operations of the first switching transistor. The smaller the deviation between the predicted voltage value and the reference voltage value of the flying capacitor at the next moment, and the fewer the number of switching transistor operations, the smaller the cost function value. Therefore, the cost function balances the capacitor voltage balance and the limitation on the number of switching operations.

[0028] S120: At the current moment, calculate the cost function value of multiple switch state combinations, and take the switch state combination with the minimum cost function value as the target switch state combination at the current moment, and control the operation of each first switch and each second switch according to the target switch state combination.

[0029] by Figure 1 Taking a medium-sized inverter with four first switching transistors as an example, there are a total of 16 possible switching state combinations, as shown in Table 1. S1 is the switching state of the first first switching transistor T1, S2 is the switching state of the second first switching transistor T2, S3 is the switching state of the third first switching transistor T3, and S4 is the state of the fourth first switching transistor T4.

[0030] Table 1 Inverter Switching State Combinations Calculate the cost function values ​​corresponding to at least some of the 16 switch state combinations mentioned above. If the cost function value corresponding to the second set of switch state combinations is the smallest, then "1110" is taken as the target switch state combination at the current moment. Then, at the current moment, control the first first switch T1 to turn on, the second first switch T2 to turn on, the third first switch T3 to turn on, the fourth first switch T4 to turn off, the first second switch T10 to turn off, the second second switch T20 to turn off, the third second switch T30 to turn off, and the fourth second switch T40 to turn on. In an optional implementation, all switch state combinations can be traversed, the cost function value under each switch state combination can be calculated, and then the switch state combination corresponding to the smallest cost function value can be taken as the target switch state combination.

[0031] In this embodiment, a cost function is established based on the deviation between the predicted voltage value and the reference voltage value of each flying capacitor at the next moment. The switching state combination that minimizes the cost function value is then identified as the target switching state combination. This ensures that the voltage deviation between the flying capacitor and the reference voltage value is small, guaranteeing that as the inverter operates, the voltage of the flying capacitor remains near the reference voltage value under the control of the target switching state combination at each moment, thus achieving voltage balance of the flying capacitor. Simultaneously, the cost function also considers the number of switching actions, incorporating both the voltage error of the flying capacitor and the number of switching operations. This effectively suppresses unnecessary switching actions while achieving real-time capacitor voltage balance.

[0032] Optionally, based on the principle of charge conservation, the dynamic equation of the flying capacitor in a five-level flying capacitor inverter can be expressed as: ; In the above formula The capacitance value of the first flying capacitor C1, The capacitance value of the second flying capacitor C2, This is the capacitance value of the third flying capacitor C3.

[0033] Prior to S110, it also included: Based on the backward Euler method, the voltage values ​​of each flying capacitor at the next moment are obtained according to the dynamic equation of the flying capacitor and the voltage values ​​of each flying capacitor sampled at the current moment.

[0034] The predicted value of the flying capacitor voltage can be calculated using the backward Euler method: ; Among them, T s For power frequency cycle, The voltage value of the first flying capacitor C1 sampled at time m (the current time), The voltage value of the second flying capacitor C2 sampled at time m, Let be the voltage value of the third flying capacitor C3 sampled at time m. Let m be the inverter current value sampled at time m.

[0035] Figure 3 A flowchart of another inverter control method provided in an embodiment of the present invention is shown below. Figure 3 The method includes: S111: Determine the total harmonic distortion rate (THD) function of the inverter's output voltage based on the function of the specific harmonic cancellation modulation waveform; the THD function of the inverter's output voltage is related to each level switching angle; the specific harmonic cancellation modulation waveform is the target waveform of the output voltage.

[0036] Specific Harmonic Elimination Modulation (SHEPWM) is an optimized pulse width modulation technique that focuses on determining the level switching angles to remove specific low-order harmonics. Unlike traditional SPWM (Sinusoidal Pulse Width Modulation) or SVPWM (Space Vector Pulse Width Modulation), SHEPWM does not rely on a high-frequency carrier wave and triangular wave comparison. Instead, it starts from a desired output voltage waveform, calculates a set of precise level switching angles, and then controls the switching transistors to operate at these angles, thereby ensuring that certain specified harmonic components in the Fourier series of the output voltage are zero.

[0037] Once the specific harmonic cancellation modulation waveform required is determined, the corresponding waveform function can be determined. Furthermore, based on the function of the specific harmonic cancellation modulation waveform, the total harmonic distortion (THD) function can be determined. The THD is a core power quality indicator that measures the degree of distortion in voltage or current waveforms; it quantifies the deviation of the actual waveform from the ideal sine wave.

[0038] Furthermore, S111 includes: Step a1: Obtain the fundamental amplitude function and the amplitude functions of each odd harmonic of the specific harmonic cancellation modulation waveform, wherein the fundamental amplitude function is related to each level switching angle, and the amplitude functions of each odd harmonic are related to each level switching angle.

[0039] To reduce computational complexity, a 1 / 4-cycle symmetrical waveform is typically used, with each output voltage level switching only once within the 1 / 4-cycle. Therefore, the function of the specific harmonic cancellation modulation waveform... It can be represented as: ; n is an odd number. For the first level switching angle, This is the second level switching angle. In this embodiment, the fundamental amplitude function is related to the first and second level switching angles, and the amplitude functions of each odd-order harmonic are also related to the first and second level switching angles.

[0040] Once the function for harmonic cancellation modulation waveform is determined, the fundamental amplitude function can be obtained. and the amplitude function of each odd harmonic , , wait.

[0041] Step b1: Construct the total harmonic distortion rate function based on the fundamental amplitude function and the amplitude functions of each odd-order harmonic.

[0042] Total Harmonic Distortion Rate Function satisfy: For example, the sum of squares of the amplitudes of odd-order harmonics between the 3rd and 39th harmonics can be calculated, that is, n is at most 39.

[0043] S121: Construct an optimization function based at least on the total harmonic distortion rate function.

[0044] In this embodiment, the total harmonic distortion rate function can be directly used as the optimization function.

[0045] S131: Solve for the optimal set of level switching angles with the goal of minimizing the optimization function. The set of level switching angles includes each level switching angle.

[0046] Related algorithms from relevant technologies, such as particle swarm optimization, can be used to solve the problem. The optimal level switching angle group is the one that minimizes the search function.

[0047] S141: Determine the level signal of the specific harmonic cancellation modulation waveform at the current moment based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment.

[0048] Once the optimal level switching angle set is determined, the optimal first level switching angle can be determined. and the optimal second level switching angle Within one fundamental frequency period, the interval [θ, θ+] ]、[θ+π- ,θ+π]、[θ+π,θ+π+ ] and [θ+2π- Within the range [θ+2π], the output level is zero; within the range [θ+2π], the output level is zero. ,θ+ ] and [θ+π- ,θ+π- Within [θ+], output E / 4 level; within the interval [θ+] ,θ+π- Within the range [θ+π+], it is denoted as E / 2 level; the interval [θ+π+] ,θ+π+ ] and [θ+2π- ,θ+2π- Within the range [θ+π+], the output level is -E / 4; within the range [θ+π+], the output level is -E / 4. ,θ+2π- Within [the specified range], the output level is -E / 2.

[0049] Based on the current time interval, the level signal corresponding to the specific harmonic cancellation modulation waveform at the current time can be determined.

[0050] S151: Determine a preset switch state combination group based on the level signal corresponding to the specific harmonic cancellation modulation waveform at the current moment; wherein, the preset switch state combination group includes various switch state combinations that make the output voltage equal to the level signal corresponding to the specific harmonic cancellation modulation waveform.

[0051] The relationship between each switch state combination and the corresponding output voltage is shown in Table 2.

[0052] Table 2: Correspondence between Switch State Combinations and Output Voltage For example, if the current time is in [θ, θ+] If the current output voltage is 0 within the specified range, then the preset switch state combination group can be determined to include 6 switch state combinations, namely (1,1,0,0), (1,0,1,0), (1,0,0,1), (0,1,1,0), (0,1,0,1), (0,0,1,1).

[0053] S161: Obtain the cost function.

[0054] S171: At the current moment, calculate the cost function value of each switch state combination in the preset switch state combination group, and take the switch state combination corresponding to the minimum cost function value as the target switch state combination at the current moment, and control the operation of each first switch and each second switch according to the target switch state combination.

[0055] Iterate through each switch state combination in the preset switch state combination group, and select the switch state combination with the minimum cost function value as the target switch state combination. It is not necessary to traverse all switch state combinations.

[0056] In this embodiment, the level switching angle is solved with the goal of minimizing the total harmonic distortion of the output voltage, which ensures that the output voltage waveform has high power quality. At the same time, after receiving the level signal of the specific harmonic cancellation modulation waveform, that is, after receiving the output voltage at the current moment, the optimization is performed in the set of switching states under the corresponding level, reducing the number of optimizations.

[0057] Figure 4 A flowchart of another inverter control method provided in an embodiment of the present invention is shown below. Figure 4 The method includes: S112: Based on droop control, generate the phase reference value and amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment according to the inverter's output voltage and output current at the current moment.

[0058] At the current moment, the inverter's output voltage is sampled to obtain the current output voltage, and the output current is sampled to obtain the current output current.

[0059] First, based on the inverter's output voltage and output current at the current moment, calculate the inverter's active power and reactive power at the current moment.

[0060] The inverter's output voltage u0(t) at the current moment is input into the first-order generalized integrator to obtain the quadrature components of the inverter's output voltage at the current moment. (This can be understood as a component of the output voltage lagging 90° at the current moment.) The inverter's output current i0(t) at the current moment is input into the second-order generalized integrator to obtain the orthogonal component of the inverter's output current at the current moment. (This can be understood as a component of the output current that lags behind the current moment by 90°).

[0061] , ; in, The output voltage angular frequency, The damping coefficient of the second-order generalized integrator (assuming that the damping coefficients of both the first and second-order generalized integrators are equal to...) And corresponding to the same output voltage angular frequency.

[0062] Based on the inverter's output voltage, output current, quadrature components of the output voltage, and output current at the current moment, the reactive power and active power of the inverter at the current moment are generated. Then, a notch filter is used to remove the secondary fluctuations in the active power calculation. Finally, the active power P(t) and reactive power Q(t) at the current moment satisfy: , ,in, ; The damping coefficient of the notch filter is generally... and All set to .

[0063] Secondly, based on the active power droop factor and the deviation between the inverter's active power at the current moment and the active power reference value at the current moment, a phase reference value for the specific harmonic cancellation modulation waveform at the current moment is generated. Based on the reactive power droop factor and the deviation between the inverter's reactive power at the current moment and the reactive power reference value at the current moment, an amplitude reference value for the specific harmonic cancellation modulation waveform at the current moment is generated.

[0064] Based on the active-frequency and reactive-voltage droop characteristics of traditional synchronous generators, it is known that the phase reference value of the modulation voltage can be obtained using active power control, and its amplitude reference value can be determined based on reactive power control. Therefore, the phase reference value θ and amplitude reference value Vm of the specific harmonic cancellation modulation waveform at the current moment satisfy: ; ; in, k m and k n These are the droop coefficients for active and reactive power, respectively. P * is the active power reference value. Q * represents the reactive power reference value, which can be set based on experience or requirements. V is the output voltage amplitude feedforward, which can be obtained based on the reactive power reference value and the load impedance. The amplitude reference value Vm may not include V.

[0065] S122: Establish a voltage deviation function based on the deviation between the fundamental amplitude function of the specific harmonic cancellation modulation waveform and the amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment; the fundamental amplitude function is related to each level switching angle.

[0066] The absolute value of the difference between the fundamental amplitude function and the amplitude reference value is taken as the voltage deviation function; that is, the voltage deviation function is: , .

[0067] S132: Determine the total harmonic distortion rate function of the inverter's output voltage based on a function of a specific harmonic cancellation modulation waveform.

[0068] S142: Construct the optimization function based on the voltage deviation function and the total harmonic distortion rate function.

[0069] The sum of the voltage deviation function and the total harmonic distortion rate function is used as the optimization function.

[0070] The optimization function satisfies: .

[0071] S152: Solve for the optimal set of level switching angles with the goal of minimizing the optimization function. The set of level switching angles includes each level switching angle.

[0072] S162: Determine the level signal of the specific harmonic cancellation modulation waveform at the current moment based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment.

[0073] S172: Determine the preset switch state combination group based on the level signal corresponding to the specific harmonic cancellation modulation waveform at the current moment.

[0074] S182: Obtain the cost function.

[0075] S192: At the current moment, calculate the cost function value of each switch state combination in the preset switch state combination group, and take the switch state combination corresponding to the minimum cost function value as the target switch state combination at the current moment, and control the operation of each first switch and each second switch according to the target switch state combination.

[0076] In this embodiment, droop control is used to acquire the amplitude and phase reference values ​​of a specific harmonic cancellation modulation waveform in real time, achieving closed-loop regulation of the output power. The level switching angle is solved with the goal of minimizing the total harmonic distortion of the output voltage, which improves the output waveform quality. Based on this, a model predictive control algorithm is introduced. After receiving the level signal of the specific harmonic cancellation modulation waveform, the model predictive control algorithm optimizes the switching state combinations at the corresponding level, greatly reducing the computational burden. The designed cost function includes constraints on both capacitor voltage error and the number of switching operations, effectively suppressing unnecessary switching actions while achieving real-time capacitor voltage balance.

[0077] Figure 5 This is a steady-state simulation result diagram of an inverter provided in an embodiment of the present invention. Figure 5 To adopt the above Figure 4 The simulation results obtained by the method shown are Figure 5 The graph includes four sub-graphs: flying capacitor voltage, output current, output voltage, and switching node voltage, with the horizontal axis representing time. As can be seen from the graph, the three flying capacitor voltages are all balanced at their respective voltage reference values, and the output voltage and output current remain stable with high waveform quality. The switching node voltage exhibits a five-level stepped waveform, consistent with the theoretical waveform. This demonstrates that the proposed control method has good steady-state performance.

[0078] Figure 6 This invention provides a waveform diagram of a switching transistor drive, where the horizontal axis represents time and the vertical axis represents the switching state or driving voltage of the first switching transistor. Figure 6 The diagram illustrates the drive waveforms over two power frequency cycles. As can be seen from the figure, within one power frequency cycle, each first switch transistor switches 4 to 6 times. This demonstrates that the proposed method effectively suppresses the number of switching operations and significantly reduces switching losses.

[0079] Figure 7 This invention provides a dynamic simulation diagram of the step power level of an inverter, from which... Figure 7As can be seen, when the power changes, the modulation ratio of the output voltage changes, and the switching node voltage can be dynamically adjusted in real time to ensure that the output power tracks the reference value. During dynamic switching, the flying capacitor voltage remains balanced and is almost unaffected. This demonstrates that the proposed method has good dynamic performance under dynamic power reference switching.

[0080] It should be understood that the various forms of processes shown above can be used, with steps reordered, added, or deleted. For example, the steps described in this invention can be executed in parallel, sequentially, or in different orders, as long as the desired result of the technical solution of this invention can be achieved, and this is not limited herein.

[0081] The specific embodiments described above do not constitute a limitation on the scope of protection of this invention. Those skilled in the art should understand that various modifications, combinations, sub-combinations, and substitutions can be made according to design requirements and other factors. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of this invention should be included within the scope of protection of this invention.

Claims

1. A control method for an inverter, characterized in that, This invention relates to a five-level inverter with a flying capacitor, wherein the inverter includes a plurality of first switching transistors connected in series between the positive terminal of the DC power supply and an intermediate node, and a plurality of second switching transistors connected in series between the negative terminal of the DC power supply and the intermediate node, wherein each second switching transistor corresponds one-to-one with a first switching transistor; a flying capacitor is connected between the connection node of each pair of adjacent first switching transistors and the connection node of each pair of adjacent second switching transistors; the first switching transistors and the corresponding second switching transistors are complementary in conduction. The method includes: Obtain a cost function; wherein the cost function includes a first influence factor and a second influence factor, the first influence factor being determined based on the deviation between the predicted voltage value of each flying capacitor at the next time step and the reference voltage value at the next time step; the predicted voltage value of each flying capacitor at the next time step is related to the current switching state combination; the switching state combination includes the switching state of each of the first switching transistors; the second influence factor being determined based on the switching state of each of the first switching transistors at the previous time step and the switching state at the current time step; At the current moment, the cost function value of multiple switch state combinations is calculated, and the switch state combination corresponding to the minimum cost function value is taken as the target switch state combination at the current moment. The operation of each first switch and each second switch is controlled according to the target switch state combination.

2. The control method of the inverter according to claim 1, characterized by, Before obtaining the cost function, the following is also included: Based on the backward Euler method, the voltage values ​​of each flying capacitor at the next time step are obtained according to the dynamic equation of each flying capacitor and the voltage values ​​of each flying capacitor sampled at the current time step.

3. The control method of the inverter according to claim 1, characterized by, Before obtaining the cost function, the following is also included: The total harmonic distortion (THD) function of the inverter's output voltage is determined based on a function of a specific harmonic cancellation modulation waveform; the THD function of the inverter's output voltage is related to each level switching angle; the specific harmonic cancellation modulation waveform is the target waveform of the output voltage; An optimization function is constructed based at least on the total harmonic distortion rate function; The optimal set of level switching angles is solved with the goal of minimizing the optimization function, and the set of level switching angles includes each of the level switching angles. Based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment, determine the level signal of the specific harmonic cancellation modulation waveform at the current moment; Based on the level signal of the specific harmonic cancellation modulation waveform at the current moment, a preset switch state combination group is determined; wherein, the preset switch state combination group includes each switch state combination that makes the output voltage equal to the level signal of the specific harmonic cancellation modulation waveform at the current moment. At the current moment, calculating the cost function value for multiple combinations of switching states includes: At the current moment, calculate the cost function value for each switch state combination in the preset switch state combination group.

4. The control method of the inverter according to claim 3, characterized by, The function for determining the total harmonic distortion rate of the inverter's output voltage based on a specific harmonic cancellation modulation waveform includes: Obtain the fundamental amplitude function and the amplitude functions of each odd harmonic of the specific harmonic cancellation modulation waveform, wherein the fundamental amplitude function is related to each level switching angle, and the amplitude functions of each odd harmonic are related to each level switching angle; The total harmonic distortion rate function is constructed based on the fundamental amplitude function and the amplitude functions of each odd-order harmonic.

5. The control method of the inverter according to claim 4, characterized by, The function of the specific harmonic cancellation modulation waveform is: ； n is an odd number, is a first level switching angle, is a second level switching angle.

6. The control method of the inverter according to claim 3, characterized by, Before determining the output voltage at the current moment based on the optimal level switching angle group and the phase reference value of the specific harmonic cancellation modulation waveform at the current moment, the method further includes: Based on droop control, a phase reference value for the specific harmonic cancellation modulation waveform at the current moment is generated according to the output voltage and output current of the inverter at the current moment.

7. The control method for the inverter according to claim 6, characterized in that, The step of generating the phase reference value of the specific harmonic cancellation modulation waveform at the current moment based on droop control, according to the output voltage and output current of the inverter at the current moment, includes: Calculate the active power of the inverter at the current moment based on the inverter's output voltage and output current. Based on the droop coefficient of the active power and the deviation between the active power of the inverter at the current moment and the active power reference value at the current moment, a phase reference value for the specific harmonic cancellation modulation waveform at the current moment is generated.

8. The control method for the inverter according to claim 3, characterized in that, Before constructing the optimization function based at least on the total harmonic distortion rate function, the method further includes: Based on droop control, an amplitude reference value for the specific harmonic cancellation modulation waveform at the current moment is generated according to the output voltage and output current of the inverter at the current moment. A voltage deviation function is established based on the deviation between the fundamental amplitude function of the specific harmonic cancellation modulation waveform and the amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment; the fundamental amplitude function is related to each of the level switching angles; At least the optimization function constructed based on the total harmonic distortion rate function includes: The optimization function is constructed based on the voltage deviation function and the total harmonic distortion rate function.

9. The control method of the inverter according to claim 8, characterized by, The method of generating the amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment based on droop control and the output voltage and output current of the inverter includes: Calculate the reactive power of the inverter at the current moment based on the inverter's output voltage and output current. Based on the droop factor of reactive power and the deviation between the reactive power of the inverter at the current moment and the reactive power reference value at the current moment, an amplitude reference value of the specific harmonic cancellation modulation waveform at the current moment is generated.

10. The control method of the inverter according to claim 9, characterized by, Based on the inverter's output voltage and output current at the current moment, the reactive power of the inverter at the current moment is calculated as follows: The output voltage of the inverter at the current moment is input into the first second-order generalized integrator to obtain the quadrature component of the output voltage of the inverter at the current moment; the output current of the inverter at the current moment is input into the second second-order generalized integrator to obtain the quadrature component of the output current of the inverter at the current moment. The reactive power of the inverter at the current moment is generated according to the output voltage of the inverter at the current moment, the output current of the inverter at the current moment, the orthogonal component of the output voltage of the inverter at the current moment, and the orthogonal component of the output current of the inverter at the current moment.

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