Dynamic avoidance and synchronization control method of piezoelectric actuator under high-pressure microfluidic environment

Abstract

The application relates to the technical field of microfluidic chip micro-actuator control, and discloses a dynamic avoidance and synchronous control method of a piezoelectric actuator under a high-voltage microfluidic environment.The method extracts transient characteristic parameters and hardware link parameters of a high-voltage electric field generating unit, calculates a transient interference attenuation time domain, and demarcates a transient forbidden area; the initial control sequence is subjected to overlap determination with the forbidden area, a dynamic phase shift operation is performed to avoid interference when there is an intersection, and a time shift amount is recorded; real-time equivalent admittance data are acquired through an output pulse width limited detection signal to calculate an admittance offset coefficient; and the time shift amount and the admittance offset coefficient are taken as inputs, the equivalent pulse width is reconstructed under the condition that the driving voltage peak value is kept constant, and a control instruction is issued.The application actively avoids transient interference, converts multi-dimensional deviation into a pulse width adjustment amount without adjusting the power supply voltage, and completes actuating energy closed loop synchronous compensation.

Description

Technical Field

[0001] This invention relates to the field of microfluidic chip microactuator control technology, specifically a dynamic avoidance and synchronization control method for piezoelectric actuators in a high-voltage microfluidic environment. Background Technology

[0002] In complex microfluidic analysis systems, high-voltage electric fields and piezoelectric actuators are often used in tandem to perform operations such as electrophoretic separation and fluid pumping. When the high-voltage electric field generating unit switches its operating voltage state, the rapid voltage jumps due to parasitic capacitances on both sides of the hardware isolation barrier can induce transient interference currents. This common-mode transient interference can affect the cross-isolation communication link between the main controller and the underlying drive module, causing digital control commands to flip levels or become out of sequence, resulting in deviations from the preset operating cycle of the piezoelectric actuator's basic control timing.

[0003] Simultaneously, the transient electric field generated by the high-voltage state switching couples to the piezoelectric actuator in the microfluidic chip's flow channel region, causing a deflection in the crystal polarization state within the piezoelectric material. This results in transient deviations in the mechanical stiffness and electromechanical conversion efficiency of the piezoelectric element. Conventional systems struggle to accurately assess this physical distortion. Directly injecting conventional test levels for parameter acquisition can easily trigger macroscopic mechanical displacement in the piezoelectric actuator, thereby disrupting the original continuity of fluid movement within the microfluidic channel.

[0004] Furthermore, miniaturized microfluidic control platforms often use fixed-amplitude power supplies to power piezoelectric actuators. With a hardware architecture where the peak drive voltage is limited and kept constant, existing control logic lacks a multi-dimensional joint adjustment mechanism to address the time-domain deviations caused by electromagnetic interference and the electromechanical state deviations induced by high-pressure stress. The system cannot effectively integrate time loss and work efficiency decay at the digital control level, resulting in uneven effective energy injection into the piezoelectric actuator during each drive cycle. This makes it difficult to achieve closed-loop energy compensation in the digital domain, ultimately causing the piezoelectric actuator's mechanical output action to lose consistency and stability. Summary of the Invention

[0005] To address the shortcomings of existing technologies, this invention provides a dynamic avoidance and synchronization control method for piezoelectric actuators in a high-voltage microfluidic environment. This method solves the problems of control timing deviation caused by common-mode transient interference during high-voltage state switching, difficulty in obtaining non-destructive electromechanical characteristic distortions of piezoelectric materials, and lack of closed-loop energy compensation for time-domain and physical state deviations under constant driving voltage.

[0006] To achieve the above objectives, the present invention provides a method for dynamic avoidance and synchronization control of piezoelectric actuators under high-voltage microfluidic conditions, comprising the following steps: Extract transient characteristic parameters and hardware link parameters of the high-voltage electric field generating unit, calculate the attenuation time domain of common-mode transient interference, and delineate transient no-entry zones on the global time axis; The initial control sequence and the transient restricted area are time-domain overlap determined. When there is a time-domain intersection, a dynamic phase shift operation is performed to avoid the transient restricted area and the resulting time shift is recorded. The isolation unit sends a probe detection command to the local controller, which then outputs a probe electrical signal to the piezoelectric actuator and obtains the real-time equivalent admittance data to calculate the admittance offset coefficient. Using time shift and admittance offset coefficient as input variables, state calculations are performed to reconstruct the equivalent pulse width while keeping the upper limit of the peak value of the driving voltage constant, and to generate reconstruction control commands. The reconfiguration control command is sent to the local controller, which controls the drive circuit to output electrical energy to complete the mechanical drive of the piezoelectric actuator.

[0007] In the above technical solution, a transient current attenuation model is established by extracting the physical parameters of the system hardware, and a closed interval of electromagnetic interference influence is pre-defined on the time axis, so that the underlying control command can actively avoid transient interference at the time domain level; the actual electromechanical characteristic deviation of the piezoelectric material under high pressure stress is obtained by using parameter-limited digital detection signals; the mechanical momentum deviation introduced by phase shift avoidance and the actuation efficiency deviation caused by high voltage electric field are converted into pulse width adjustment variables, and the equivalent integral reconstruction of electrical energy to mechanical energy is completed without adjusting the external power supply hardware voltage, so as to maintain the continuous mechanical drive of the piezoelectric actuator.

[0008] Furthermore, transient characteristic parameters and hardware link parameters of the high-voltage electric field generating unit are extracted, the attenuation time domain of common-mode transient interference is calculated, and a transient no-entry zone is delineated on the global time axis, specifically including: When a pre-trigger signal indicating that the voltage state of the high-voltage electric field generating unit is about to switch is received, or before the high-voltage electric field generating unit is about to switch voltage state according to the preset schedule, the slew rate parameter is extracted as a transient characteristic parameter, and the capacitor time constant is extracted as a hardware link parameter. Establish a common-mode transient current decay model equation describing the process across isolation units; Substitute the system's preset safe communication threshold current parameter into the common-mode transient current decay model equation, and solve in reverse the time span required for the transient common-mode current prediction value to dissipate to within the safe limit. Define the solution result as the electromagnetic interference dissipation time. By combining the propagation delay of the digital signal of the isolation unit with the preset timing protection margin, a time closed interval consisting of the start time boundary and the end time boundary of the restricted area is determined on the global time axis as the transient restricted area.

[0009] Furthermore, the initial control sequence and the transient restricted area are subjected to time-domain overlap determination. When a time-domain intersection exists, a dynamic phase-shifting operation is performed, specifically including: Extract the initial control sequence containing the initial trigger phase and the initial reference pulse width, establish a basic working interval representing the original execution cycle, and determine whether the start node of the basic working interval is earlier than or equal to the end time boundary of the restricted area, and whether the end node of the basic working interval is later than or equal to the start time boundary of the restricted area. If the determination result is true, then it is determined that there is a temporal intersection, and the forward advance translation node and the backward delay translation node are calculated respectively; Select the node with the smaller absolute value of the time deviation from the initial trigger phase as the target safe time domain for translation, calculate the difference between the target safe time domain and the initial trigger phase as the time translation amount and record it, and translate the initial trigger phase to the target safe time domain as the new trigger phase.

[0010] Furthermore, the local controller outputs a probe electrical signal to the piezoelectric actuator, specifically including: The test pulse width of the probed electrical signal is limited to the lower limit of the basic mechanical resonance period ratio of the piezoelectric actuator; The amount of charge injected in a single probe signal is limited to below the minimum charge threshold required for the piezoelectric actuator to generate measurable macroscopic displacement, and a charge release interval is set between adjacent probe detections.

[0011] Furthermore, the admittance offset coefficient is calculated from the real-time equivalent admittance data obtained from the return, specifically including: Real-time equivalent admittance data is derived by performing definite integral operations in the time dimension on discrete transient response current data and combining it with fixed supply voltage amplitude parameters. Calculate the ratio of the preset reference steady-state admittance parameter to the real-time equivalent admittance data to obtain the admittance offset coefficient; When calculating the ratio, the system's preset lower limit threshold for dead zone admittance is compared with the real-time equivalent admittance data, and the maximum value is taken as the denominator for the ratio calculation.

[0012] Furthermore, state calculations are performed using the time shift and admittance offset coefficient as input variables to reconstruct the equivalent pulse width while keeping the upper limit of the driving voltage peak constant. Specifically, this includes: Based on the second-order equivalent electromechanical inertia, the time compensation factor for correcting the time displacement deviation is calculated using the time translation and the damping coefficient of the equivalent electromechanical system. The state compensation factor for correcting the attenuation of work efficiency is calculated using the admittance offset coefficient and the piezoelectric strain correction gain. Multiply the time compensation factor and the state compensation factor, and then multiply them by the initial reference pulse width to obtain the reconstructed equivalent pulse width; The information containing the reconstructed equivalent pulse width and the new trigger phase is encapsulated to generate a reconstructed control instruction.

[0013] Furthermore, after obtaining the reconstructed equivalent pulse width, and before generating the reconstructed control command, the process also includes execution cycle timing out-of-bounds verification and compensation margin accumulation. The calculated reconstructed equivalent pulse width is compared with the upper limit tolerance of the currently set PWM running cycle. If the reconstructed equivalent pulse width exceeds the maximum available time slot of the period, it will be forcibly truncated to the specific value of the maximum available time slot. Calculate the difference between the reconstructed equivalent pulse width before truncation and the maximum available time slot. Store the difference as an uncompensated time margin in the accumulator register. In the reconstruction calculation of the next PWM running cycle, add the time margin to the newly calculated reconstructed equivalent pulse width.

[0014] Furthermore, after the execution cycle timing out-of-bounds check and before the generation of the reconfiguration control instruction, a secondary check is also performed: The reconstruction working range is re-established based on the currently calculated equivalent pulse width. A secondary check is performed on the reconstructed working interval and the transient restricted area. If the reconstructed working interval and the transient restricted area overlap, the reconstructed equivalent pulse width is truncated according to the maximum safe available time slot. If the truncated reconstructed equivalent pulse width is less than the minimum effective driving pulse width, then the reconstructing control instruction for this cycle will be set to the zero pulse width safety instruction.

[0015] Furthermore, before issuing the reconfiguration control command to the local controller, the following steps are also included: Periodically send synchronization data frames containing timestamps to establish time synchronization relationships across isolation units, and convert new trigger phases into local counter trigger values ​​that can be recognized by the local controller; Determine if the sum of the current time, data frame transmission time, isolation unit propagation delay, and local controller register loading time is earlier than the new trigger phase; if the condition is not met, discard the current cycle drive instruction or generate a zero-pulse-width safety control instruction.

[0016] Furthermore, after the reconfiguration control command is issued to the local controller, it also includes: The local controller extracts the cyclic redundancy check code appended to the end of the data frame and performs cyclic redundancy check verification; If the verification result contains an error, the abnormal data frame is discarded, the preset safe pulse width output is maintained, and the internal continuous error counter is incremented. When the accumulated value of the continuous error counter exceeds the preset safe shutdown threshold, a low-level hardware interrupt is triggered, forcibly cutting off the power output of the drive circuit. If the verification passes, the reconstructed equivalent pulse width is written into the shadow compare register of the timer, and automatically loaded into the active compare register by the hardware at the moment the update event is triggered.

[0017] This invention provides a dynamic avoidance and synchronization control method for piezoelectric actuators in a high-voltage microfluidic environment. It has the following beneficial effects: 1. This invention extracts transient characteristic parameters and hardware link parameters from the high-voltage electric field generating unit, calculates the attenuation time domain of common-mode transient interference, and delineates a transient restricted zone on the global time axis. Dynamic phase-shifting is then performed on control sequences where the basic operating interval and the transient restricted zone intersect. This ensures that the underlying control commands avoid the common-mode transient interference interval caused by high-voltage electric field state switching in timing, preventing pulse execution errors due to electromagnetic interference and improving the stability of the control system in complex electromagnetic environments.

[0018] 2. This invention uses a limited probe signal for detection, restricting the pulse width and injected charge of the probe signal to below the minimum threshold for macroscopic displacement generated by the piezoelectric actuator. Real-time equivalent admittance data is calculated based on the returned transient response current. This allows for the acquisition of accurate electromechanical characteristic shift data of the piezoelectric material under high-voltage electric field stress without triggering actual mechanical action of the piezoelectric actuator, providing accurate data input for subsequent energy reconfiguration of the system.

[0019] 3. This invention improves upon the method of using the time shift generated by dynamic phase shift and the admittance offset coefficient obtained by probe detection as input variables to reconstruct the equivalent pulse width through state calculation while maintaining a constant peak driving voltage. Therefore, no additional hardware voltage regulation circuit is needed. The deviation in the time dimension and the deviation in electromechanical conversion efficiency are directly converted into the adjustment amount of the driving pulse width, completing closed-loop compensation of the actuation energy and ensuring the continuity and consistency of the piezoelectric actuator's mechanical output. Attached Figure Description

[0020] Figure 1 This is a schematic diagram of the system architecture of the present invention; Figure 2 This is a schematic diagram of the method flow of the present invention; Figure 3 This is a timing distribution diagram of the transient restricted area in this invention; Figure 4 This is the control sequence timing prediction and dynamic phase shifting logic diagram of the present invention; Figure 5 This is the logic diagram of the ultra-narrow pulse width detection timing and admittance operation of the present invention; Figure 6This is the logic diagram of the multidimensional variable equivalent energy reconstruction algorithm of the present invention; Figure 7 This is a diagram illustrating the cross-isolation driver execution logic of the reconfiguration instructions in this invention. Figure 8 This is a schematic diagram of the control sequence timing prediction and dynamic phase shifting logic of the present invention; Figure 9 This is a schematic diagram illustrating the effective mechanical displacement stability comparison of the present invention; Figure 10 This is a schematic diagram of the piezoelectric actuator admittance offset feature tracking of the present invention.

[0021] Among them, 10 is the main controller; 20 is the isolation unit; 30 is the high-voltage floating platform; 31 is the local controller; 32 is the drive circuit; 33 is the current sampling unit; 40 is the piezoelectric actuator; and 50 is the high-voltage electric field generating unit. Detailed Implementation

[0022] The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0023] See attached document Figure 1 This invention provides a dynamic avoidance and synchronization control system for piezoelectric actuators under high-voltage microfluidic conditions. The system includes: The system includes a main controller 10, an isolation unit 20, a high-voltage floating platform 30, a piezoelectric actuator 40, and a high-voltage electric field generating unit 50. The high-voltage electric field generating unit 50 includes a high-voltage power supply, high-voltage electrodes, and a high-voltage drive interface for applying a high-voltage electric field to the flow channel region of the microfluidic chip.

[0024] The main controller 10 establishes a digital communication link with the high-voltage floating platform 30 without DC conductive connection through the isolation unit 20. The main controller 10 acquires the status of the high-voltage electric field generating unit 50 through the low-voltage side feedback terminal of the high-voltage power supply, the isolation sampling module, or the high-resistance voltage divider sampling module. This sampling link does not constitute a direct conductive communication link between the main controller 10 and the high-voltage floating platform 30.

[0025] The internal reference potential of the high-voltage floating platform 30 is synchronized with the potential of the high-voltage electric field applied by the high-voltage electric field generating unit 50. Specifically, the high-voltage floating platform 30 is powered by an isolated power supply module, the output reference terminal of which is connected to the follow-up reference node of the high-voltage electric field generating unit 50, so that the high-voltage floating platform 30, the local controller 31, and the drive circuit 32 are in a common-mode floating state that changes with the high-voltage potential. The high-voltage floating platform 30 internally houses the local controller 31, the drive circuit 32, and the current sampling unit 33. The current sampling unit 33 is connected between the drive circuit 32 and the piezoelectric actuator 40, and is used to collect the transient response current of the drive port of the piezoelectric actuator 40 within the probe detection window, and input the sampled signal to the analog-to-digital conversion port of the local controller 31.

[0026] The local controller 31 receives digital instructions from the main controller 10 and outputs a pulse width modulation signal to control the on / off state of the drive circuit 32. The drive circuit 32 applies corresponding drive energy to the piezoelectric actuator 40 based on an internally fixed isolated power supply voltage.

[0027] The piezoelectric actuator 40 is located in the flow channel structure of the microfluidic chip and changes the fluid volume parameters through mechanical deformation.

[0028] The high-voltage floating platform 30 adopts a fixed-amplitude power supply mode, and the upper limit of the peak value of the drive voltage across the piezoelectric actuator 40 remains constant. The drive circuit 32 is configured with a preset output impedance, a current-limiting branch, or a controlled discharge branch, so that the piezoelectric actuator 40 is in a determinable initial charge state before the start of each drive cycle, and a calibrable correspondence is formed between the injected charge amount and the pulse width of a single drive pulse within a preset pulse width range.

[0029] See attached document Figure 2 This invention provides a method for dynamic avoidance and synchronization control of piezoelectric actuators under high-voltage microfluidic conditions, the method comprising the following steps: S10: When the main controller 10 receives a pre-trigger signal indicating that the voltage state of the high-voltage electric field generating unit 50 is about to switch, or before the main controller 10 is about to trigger the voltage state of the high-voltage electric field generating unit 50 to switch according to the preset schedule, the slew rate parameter of the high-voltage electric field generating unit 50 and the capacitance time constant of the hardware link are extracted, the attenuation time domain of the common-mode transient interference is calculated, and the transient no-pass zone is defined on the global time axis. S20, generate an initial control sequence containing the initial trigger phase and the initial reference pulse width, determine the time domain overlap between the initial control sequence and the transient restricted area, perform dynamic phase shifting operation when there is time domain intersection, shift the initial trigger phase to a safe time domain outside the transient restricted area and record the time shift amount; S30, if no zero-pulse-width safety control command is generated in the current cycle, and after completing the time-domain overlap determination and before the drive command is officially issued, the isolation unit 20 sends a probe detection command to the local controller 31, causing the local controller 31 to output a detection electrical signal with a limited test pulse width to the piezoelectric actuator 40, and to obtain the real-time equivalent admittance data returned by the local controller 31, and calculate the admittance offset coefficient; wherein, when there is no time-domain intersection, the main controller 10 sets the time shift to zero, and enters the probe detection process after using the initial trigger phase as the new trigger phase; S40 uses the time shift as the input variable for fluid momentum timing compensation and the admittance offset coefficient as the input variable for electromechanical conversion efficiency compensation. Under the condition of keeping the upper limit of the peak value of the driving voltage constant, it performs multi-dimensional variable state calculation, reconstructs the equivalent pulse width according to the input variable parameters, and generates reconstruction control commands. S50 sends a reconfiguration control command, which includes the reconfiguration equivalent pulse width and the new trigger phase, to the local controller 31. The control drive circuit 32 outputs electrical energy according to the updated timing and duty cycle to complete the mechanical drive of the piezoelectric actuator 40.

[0030] To further clarify the implementation of each technical aspect of the present invention, the following will provide a detailed description of the implementation of each functional module involved above and its internal processing flow.

[0031] See attached document Figure 3 In this embodiment, step S10 is used to extract transient characteristic parameters of the high-voltage microfluidic control system and construct the time domain boundary, which may specifically include the following steps: S101, the main controller 10 extracts the slew rate parameters of the high voltage electric field generating unit 50 and the capacitance time constant of the hardware link.

[0032] Because parasitic capacitance exists between the physical isolation barriers on the high and low voltage sides of the microfluidic system, the extremely high voltage change rate can induce transient interference current. Therefore, extracting the slew rate parameter and the equivalent capacitive reactance of the system is a physical prerequisite for evaluating the system's disturbance strength.

[0033] The main controller 10 obtains the theoretical slew rate parameters of the high-voltage electric field generation unit 50 by reading the slope configuration register in the low-voltage side control interface of the high-voltage power management chip in the high-voltage electric field generation unit 50. For dynamic fluctuation conditions, the main controller 10 uses the built-in analog-to-digital converter module to collect the real-time voltage feedback signal of the high-voltage electric field generation unit 50 through the low-voltage side feedback terminal of the high-voltage power supply, the isolated sampling module, or the high-resistivity voltage divider sampling module, and performs discrete differential operation on the continuously sampled voltage data to obtain the dynamic slew rate parameters.

[0034] The capacitance time constant of the hardware link is jointly determined by the parasitic capacitance of the primary and secondary sides of the isolation unit 20 and the distributed impedance of the high-voltage floating platform 30 to ground. The non-volatile memory inside the main controller 10 records the parasitic parameter matrix calibrated at the factory. Based on the operating voltage threshold set by the current high-voltage electric field generating unit 50, the main controller 10 searches for and retrieves the matching parasitic capacitance value and equivalent impedance value from this parasitic parameter matrix, multiplies the two, and obtains the capacitance time constant under the current operating condition.

[0035] S102, the main controller 10 establishes a common-mode transient current decay model across the isolation unit 20 based on the extracted parameters.

[0036] When the voltage state switching action of the high-voltage electric field generating unit 50 causes a potential jump, the abrupt electric field will trigger a charging and discharging effect through the physical barrier of the isolation device. Based on the slew rate parameters and hardware physical parameters obtained above, the main controller 10 constructs a transient common-mode current decay model equation describing the decay state of this charging and discharging effect over time: ; In the formula, For time Decaying transient common-mode current prediction; The equivalent parasitic capacitance value of isolation unit 20 obtained by looking up the table is generally in the picofarad range. The slew rate parameter of the extracted high-voltage electric field generating unit 50 is used to characterize the voltage fluctuation per unit time. The capacitance time constant is determined in step S101; The relative time parameter is set to zero at the trigger point of the voltage switching action of the high-voltage electric field generating unit 50. It is the base of natural numbers.

[0037] For the measurement of parasitic parameters of the isolation barrier and the evaluation of high-voltage impedance matching, those skilled in the art can use conventional impedance analysis instruments or circuit parameter identification algorithms to extract them. The parameter acquisition and calculation methods are well-known technologies in this field and will not be elaborated here.

[0038] S103, the main controller 10 calculates the interference boundary and delineates the transient no-entry zone based on the common-mode transient current attenuation model.

[0039] The main controller 10 has a preset safe communication threshold current parameter in its program logic. This safe communication threshold current parameter is calculated based on the common-mode transient immunity index provided in the datasheet of the isolation unit 20, combined with the tolerance margin of the reference communication level. This safe communication threshold current parameter corresponds to the physical tolerance limit of the isolation unit 20 when it is expected that no common-mode transient level flipping errors will occur during data transmission.

[0040] The main controller 10 substitutes the secure communication threshold current parameter into the aforementioned common-mode transient current attenuation model. The location is used to inversely calculate the time span required for the predicted transient common-mode current to dissipate to within the safety limit. This solution is defined as the electromagnetic interference dissipation time.

[0041] To avoid a program dead zone caused by the infinite extension of the calculated electromagnetic interference dissipation time due to sampling noise or abnormal values, the main controller 10 presets a maximum waiting period constraint. When the calculated electromagnetic interference dissipation time exceeds this maximum waiting period constraint, the main controller 10 truncates it to the specific value of the maximum waiting period constraint.

[0042] Taking into account the inherent propagation delay of the digital signal in isolation unit 20, the main controller 10 calibrates the absolute time boundary of system scheduling. Let the target trigger time for the main controller 10 to plan and instruct the high-voltage electric field generation unit 50 to perform a state switch be... The transient no-entry zone defined by the main controller 10 on the global control time axis Defined as a continuous closed time interval, its mathematical relationship is expressed as: ; In the formula, The digital signal instruction propagation delay compensation time for isolation unit 20 covers the entire hardware transmission time of the main controller 10 sending a level signal through the digital pin to the local controller 31 to complete level recognition; To calculate and verify the electromagnetic interference dissipation time, which has been verified by the above-mentioned dead zone cutoff, This is a preset timing protection margin used to cover clock jitter, sampling errors, propagation delay dispersion of isolation devices, and uncertainties at switching transient boundaries. The value is determined based on the propagation delay jitter of the isolation unit 20, the clock synchronization error between the main controller 10 and the local controller 31, the analog-to-digital sampling period, and the switching transient response time of the drive circuit 32, and ranges from 0.5μs to 200μs.

[0043] The main controller 10 writes the defined time-closed interval into the control register, marking it as a reserved area where no piezoelectric control signal level transition bands are allowed to be sent to the local controller 31. Within this transient restricted area, the main controller 10 does not send piezoelectric drive commands, probe detection commands, time synchronization frames, or other cross-isolation digital communication data frames; the local controller 31 also does not send back admittance detection data within this transient restricted area. By establishing the upper and lower limits of the time-closed interval, the main controller 10 achieves quantization mapping and boundary isolation of physical electromagnetic transient interference at the system timing level.

[0044] See attached document Figure 4In this embodiment, step S20 is used to schedule the underlying pulse command in combination with specific timing parameters to avoid the electromagnetic interference range, to complete the safety isolation on the time axis while prioritizing the continuity of microfluidic actions, and to avoid abnormal driving by using a zero-pulse-width safety control command when the safety time slot is insufficient. Specifically, it may include the following steps: S201, the main controller 10 extracts the control sequence parameters to be issued and completes the time domain mapping.

[0045] The timer scheduling module inside the main controller 10 continuously generates an initial digital control pulse sequence for controlling the piezoelectric actuator 40 based on preset piezoelectric actuation frequency parameters. Before issuing a command, the main controller 10 extracts the initial trigger phase of the specific pulse to be executed. Compared with the initial reference pulse width The main controller 10 maps the control sequence to be executed onto the system's global time axis, establishing a basic working interval representing the original execution cycle. The basic operating range characterizes the time span during which the system is expected to output driving power under normal operating conditions without external high-voltage interference.

[0046] S202, the main controller 10 uses Boolean intersection logic to determine sequence conflicts between the basic working area and the transient restricted area.

[0047] The main controller 10 reads the established transient restricted area. ,in This is the starting time boundary of the transient restricted area. This is the end time boundary of the transient restricted area. The main controller 10 extracts the basic working interval. The time boundary parameters and the transient restricted area Perform logical intersection operations in the time domain. For determining the collision characteristics of the control sequences, the main controller 10 uses the following Boolean conditional expression: ; In the formula, A Boolean variable used to characterize the conflict state. This represents a logical AND operation. When the start node of the basic working interval is earlier than or equal to the end node of the restricted area, and the end node of the basic working interval is later than or equal to the start node of the restricted area, it indicates that the time domain covered by the original pulse intersects with the transient restricted area that already includes timing protection margin.

[0048] The main controller 10 will use Boolean variables Set to true. If the Boolean variable... The evaluation being a false value indicates that there is no risk of physical interference with the current sequence arrangement, and the main controller 10 proceeds according to the original initial trigger phase. Continue with the subsequent control process and shift the time. Set to zero, and change the original initial trigger phase. Record as a new trigger phase.

[0049] For conventional comparator circuits or logic gate array designs that determine timing crossover, those skilled in the art can program and configure them according to the register structure of the microprocessor. The underlying comparison logic is a well-known technology in the field and will not be described in detail here.

[0050] S203, when the main controller 10 determines that there is a sequence conflict, it performs a dynamic phase shift decision and quantifies the time shift amount.

[0051] When Boolean variable When the result is true, the main controller 10 invokes the phase-shifting algorithm logic to calculate candidate safe time nodes for escaping the interference boundary. The main controller 10 calculates the forward advance shift nodes respectively. With backward delay translation node .

[0052] Forward advance translation node according to The calculation indicates that the entire pulse command will be moved forward and executed before the transient forbidden zone, which already includes timing protection margins, is reached. The backward delay shift node is... The calculation indicates that the pulse command is delayed until the transient restricted area, which already includes timing protection margins, ends.

[0053] The main controller 10 calculates the absolute value of the time deviation corresponding to the two translation schemes, follows the principle of minimizing timing deviation, and selects the node with the smaller absolute value of the deviation as the initial translation target in the safe time domain. To avoid dead-time issues in the algorithm logic, the main controller 10 incorporates the current system clock reading and the next cycle pulse time node as boundary constraints. The current forward shift node... When the calculation result is earlier than the hardware clock reading of the current system at this moment, i.e., when a causal contradiction is caused, the main controller 10 determines that the forward translation has failed and forcibly changes the target safe time domain. Assigned as backward delay translation node .

[0054] Furthermore, when the system assigns the target safe time domain as backward delay translation nodes... At that time, the main controller 10 extracts the start time node of the next predetermined execution pulse. If the calculation and verification result This indicates that the current delayed operation will cause time-domain congestion between the pulse of this cycle and the pulse of the next cycle. The main controller 10 will discard the single pulse instruction to be executed and clear the time shift to prevent the PWM generator from generating uncontrollable duty cycle superposition.

[0055] In this situation, the main controller 10 generates a zero-pulse-width safety control command and ends the admittance detection and pulse width reconstruction process of the current cycle; the transient no-entry zone determination, dynamic phase shift and compensation control process are re-executed in the next PWM running cycle.

[0056] The main controller 10 completes the decision and determines the final target safety time domain. Then, calculate the actual accurate time shift. : ; In the formula, the time shift amount The value is calculated by the main controller 10 as a signed constant. A positive value indicates that the system has taken a delayed avoidance action, while a negative value indicates that the system has taken an early avoidance action. The main controller 10 records this time shift in its internal buffer. This value quantifies the accumulated time variable of electromechanical load lost or introduced ahead of time due to proactive avoidance in the time domain.

[0057] See attached document Figure 5 In this embodiment, step S30 is used to detect the transient influence of the high-voltage environment on the electromechanical properties of the piezoelectric material through a limited digital electrical signal without changing the amplitude of the hardware driving voltage, while ensuring that the detection process does not mistakenly trigger the macroscopic mechanical displacement of the piezoelectric actuator 40. Specifically, it may include the following steps: S301, the transient electric field in space caused by the voltage state switching of the high voltage electric field generating unit 50 will act on the piezoelectric actuator 40 through the high voltage electrode region of the microfluidic chip, the parasitic capacitance path, or the package coupling path.

[0058] Under the stress of an external electric field, the piezoelectric ceramic material inside the piezoelectric actuator 40 undergoes transient polarization deflection of its internal crystal domain walls. This physical change causes the equivalent mechanical stiffness and dielectric constant of the piezoelectric actuator 40 to deviate from their conventional set values, resulting in distortions in charge absorption and electromechanical conversion efficiency.

[0059] S302, after performing the dynamic phase shift operation and before the drive command is officially issued, the main controller 10 sends a probe detection command to the local controller 31 through the isolation unit 20.

[0060] The timing of sending the probe detection command, outputting the probe electrical signal, and transmitting the admittance data are all arranged outside the transient restricted area. When there is no minimum safe time slot required for probe detection and data transmission before the new trigger phase, the main controller 10 skips the probe detection in this cycle and uses the effective admittance offset coefficient of the previous cycle or the preset default admittance offset coefficient to participate in the subsequent reconstruction calculation.

[0061] After parsing the probe detection command, the local controller 31 controls the drive circuit 32 to output a probe electrical signal with a limited test pulse width to the piezoelectric actuator 40. For the pulse width of the probe electrical signal, the control system sets a mechanical constraint equation based on the inertial characteristics of the piezoelectric oscillator. The piezoelectric actuator 40 has a fixed mechanical resonance period parameter, which characterizes the shortest physical response time required for its mechanical structure to undergo effective deformation.

[0062] The local controller 31 strictly limits the test pulse width of the probe electrical signal within the lower limit of the mechanical resonant period ratio of the piezoelectric actuator 40. Simultaneously, the local controller 31 limits the amount of charge injected into the probe electrical signal in a single pulse to below the minimum charge threshold required for the piezoelectric actuator 40 to generate measurable macroscopic displacement. Furthermore, a charge release interval is set between adjacent probe detections to prevent the accumulation of multiple probe pulses to form effective mechanical drive. The time constraint relationship is expressed as: ; In the formula, To test the pulse width of a pulse-width-limited probe electrical signal; This is the fundamental mechanical resonance period of the piezoelectric actuator 40; To prevent displacement constraint proportionality coefficients from being used, their value range is set to 0.01 to 0.05.

[0063] The fundamental mechanical resonant period parameter of the piezoelectric actuator 40 can be obtained by measuring the resonant frequency of the piezoelectric actuator 40 with an impedance analyzer and taking its reciprocal. Due to the inherent inertial damping of the mechanical mass and the low-pass filtering characteristics of the mechanical system, when the duration of the driving electrical signal is much lower than the lower limit of the mechanical response, the piezoelectric actuator 40 cannot produce macroscopic mechanical deformation, but only exhibits purely capacitive charging and discharging physical characteristics. The above parameter configuration allows the system to directly evaluate the static clamping capacitance state of the piezoelectric ceramic through a digital square wave.

[0064] S303, the local controller 31 collects the transient response current of the piezoelectric actuator 40 drive port through the current sampling unit 33 within the synchronous time window of the output detection electrical signal.

[0065] Under extremely narrow pulse excitation, the electromechanical coupling motion branch of the piezoelectric material is approximately open-circuited, and its external port characteristics are equivalent to pure capacitive reactance characteristics. The high-voltage floating platform 30 adopts a fixed amplitude power supply mode, and the upper limit of the peak value of the drive voltage output by the drive circuit 32 to the piezoelectric actuator 40 remains constant.

[0066] The local controller 31 uses a built-in high-speed analog-to-digital converter module to sample the transient response current at high frequency, and performs a definite integral operation in the time dimension on the sampled discrete current data to derive the real-time equivalent admittance data. The sampling frequency of the high-speed analog-to-digital converter module is set to a preset multiple not less than the reciprocal of the detection pulse width so that no less than a preset number of current sampling points are obtained within the duration of the detection pulse. The current sampling unit 33 can be implemented using a sampling resistor, a transimpedance amplifier, a passive RC conditioning network, or an isolated current sensing device.

[0067] The calculation model for real-time equivalent admittance data is as follows: ; In the formula, The calculated real-time equivalent admittance data; The transient response current data that varies with time and is collected by the local controller 31; The power supply voltage amplitude parameters are fixed for the high-voltage floating platform 30; To detect the pulse width of an electrical signal; This is the time variable for the integration operation. After the local controller 31 completes the operation, it sends a digital frame containing real-time equivalent admittance data back to the main controller 10 through the isolation unit 20.

[0068] For the peripheral conditioning circuit and signal low-pass filtering noise reduction processing of high-speed transient current sampling, those skilled in the art can use conventional transimpedance amplifiers or passive RC networks for design. The circuit hardware configuration is a well-known technology in this field and will not be described in detail here.

[0069] S304, the main controller 10 receives the real-time equivalent admittance data returned and establishes a deviation calibration calculation model.

[0070] The internal memory of the main controller 10 contains pre-determined reference steady-state admittance parameters. The reference steady-state admittance parameters, admittance lower limit threshold, probe pulse width constraint parameters, time compensation parameters, and state compensation parameters are all stored as factory calibration parameters in the non-volatile memory of the main controller 10 or the local controller 31 and are read when the system is powered on and initialized.

[0071] The main controller 10 calculates the admittance offset coefficient by comparing the ratio of the reference steady-state admittance parameter to the real-time equivalent admittance data. The formula for calculating the admittance offset coefficient is defined as follows: ; In the formula, The calculated admittance offset coefficient; The preset reference steady-state admittance parameter characterizes the nominal charge absorption rate of the piezoelectric actuator under ideal electromechanical coupling conditions, free from external transient electric field interference. The specific value of this parameter is determined by the static equivalent capacitance of the selected piezoelectric material and the pulse width of the probe signal. In embodiments of the present invention, the preset reference steady-state admittance parameter is set to a range typically within 1 × 10⁻⁶. -3 Between 10.0S (Siemens); The real-time equivalent admittance data obtained in step S303; This is a function to find the maximum value. The lower limit threshold of the dead zone admittance is preset for the system, and its value is one percent of the reference steady-state admittance parameter.

[0072] When hardware disconnection or sampling anomaly causes real-time equivalent admittance data to approach zero, a fallback threshold for the admittance to prevent dead zone is introduced for a fallback judgment, thereby avoiding the underlying algorithm dead zone that causes the system program to crash when divided by zero.

[0073] See attached document Figure 6 In this embodiment, step S40 is used to convert the deviation in the time dimension and the deviation in the physical state dimension into an adjustment amount for the PWM duty cycle through mathematical operations in the pure digital domain while keeping the driving voltage constant, so as to realize the reconstruction of the equivalent energy of multi-dimensional variables. Specifically, it may include the following steps: S401, the main controller 10 establishes an energy integral equivalent calculation mechanism under a fixed drive amplitude.

[0074] In embodiments of the present invention, the amplitude of the supply voltage output by the drive circuit 32 to the piezoelectric actuator 40 is set to a fixed constant. The drive circuit 32 keeps the charging start state of the piezoelectric actuator 40 controllable in each drive cycle by presetting the output impedance, current limiting branch, or controlled discharge branch, so that the effective injected charge under the fixed voltage amplitude condition can vary with the duration of the drive pulse.

[0075] In the physical process of converting electrical energy into mechanical energy, under the current-limited charging or quasi-charge control conditions of the drive circuit 32, the effective injected charge obtained by the piezoelectric actuator 40 in a single drive cycle is positively correlated with the time area covered by the single pulse in a calibrable manner.

[0076] The main controller 10 adopts the principle of energy integral equivalence and changes the total amount of charge injected into the piezoelectric actuator 40 in a single work cycle by adjusting the duration of the digital drive pulse.

[0077] S402, the main controller 10 extracts the time shift, establishes an electromechanical timing compensation model, and derives the time compensation function.

[0078] The translation of the drive command on the time axis alters the kinetic energy accumulation state of the unidirectional actuation of the electromechanical load system. When the drive command is delayed, unexpected inertial loss occurs in the electromechanical response; when the drive command is advanced, the electromechanical quantities are superimposed in advance. The main controller 10 establishes a time compensation function based on second-order equivalent electromechanical inertia to correct timing displacement deviations. ; In the formula, This is the calculated dimensionless time compensation factor; The recorded signed time shift; The equivalent electromechanical system damping coefficient is preset for the system, and its dimension is the reciprocal of time.

[0079] The main controller 10 obtains this coefficient based on the initial calibration file of the piezoelectric load. To prevent the compensation multiplier from diverging due to excessive time shift, the main controller 10 sets upper and lower limit saturation constraints on the time compensation factor, limiting its value range to between 0.8 and 1.2.

[0080] S403, the main controller 10 extracts the admittance offset coefficient, establishes an electromechanical conversion efficiency compensation model, and derives the state compensation function.

[0081] The admittance offset coefficient characterizes the degree of piezoelectric stiffness distortion of the piezoelectric actuator 40 after being subjected to high-voltage transient interference. When the equivalent stiffness of the piezoelectric ceramic increases, the proportion of effective mechanical work transmitted to it by the drive circuit 32 decreases. The main controller 10 constructs a state compensation function to correct the decrease in the work efficiency of the piezoelectric actuator 40. ; In the formula, This is the calculated dimensionless state compensation factor; The input admittance offset coefficient; Gain is corrected for piezoelectric strain.

[0082] The piezoelectric strain correction gain is a fixed constant, which can be calibrated by measuring the displacement-voltage hysteresis curve of the piezoelectric actuator 40 and extracting the nonlinear attenuation rate. The main controller 10 applies boundary constraints to the state compensation factor, limiting it to a closed interval of 1.0 to 1.5.

[0083] S404, the main controller 10 executes the joint state equation solution, reconstructs the equivalent pulse width using the comprehensive compensation factor, and generates instructions.

[0084] The main controller 10 incorporates the calculation results of the two compensation functions mentioned above into the joint state equation. The mathematical equation for reconstructing the equivalent pulse width by the main controller 10 is as follows: ; In the formula, To reconstruct the equivalent pulse width; This is the original initial reference pulse width; and These are the aforementioned independent compensation factors.

[0085] The main controller 10 performs cycle timing out-of-bounds checks and accumulates compensation margins. The main controller 10 compares the reconstructed equivalent pulse width with the upper limit tolerance of the currently set PWM operating cycle. If the reconstructed equivalent pulse width exceeds the maximum available time slot of the cycle, the main controller 10 forcibly truncates it to the specific value of the maximum available time slot.

[0086] Simultaneously, the main controller 10 calculates the difference between the reconstructed equivalent pulse width before truncation and the maximum available time slot, and stores this difference as an uncompensated time margin in the accumulator register. In the reconstruction calculation of the next PWM running cycle, the main controller 10 adds this time margin to the newly calculated reconstructed equivalent pulse width.

[0087] Before encapsulating and generating reconfiguration control instructions, the main controller 10 bases the reconfiguration equivalent pulse width on... Re-establish and reconstruct the workspace and to Transient restricted areas The starting node of the next cycle pulse And perform a secondary verification at the communication cutoff time.

[0088] like and Overlap, or If the current cycle goes out of bounds, the main controller 10 will truncate the time slot according to the maximum safe available time slot. If truncated If the pulse width is less than the minimum effective drive pulse width, the main controller 10 sets the current cycle reconfiguration control instruction to a zero-pulse-width safety instruction and writes the uncompensated time margin into the accumulator register. This completes the secondary verification and determines the final value. Then, the main controller 10 will bind the final reconstructed equivalent pulse width with the target safe time domain and encapsulate it to generate a reconstructed control command.

[0089] See attached document Figure 7 In this embodiment, step S50 is used to execute cross-isolation drive logic, enabling the reconfiguration control command to cross the isolation physical barrier and be converted into piezoelectric actuator mechanical displacement. Specifically, it may include the following steps: S501, during the system power-on initialization phase, the main controller 10 and the local controller 31 establish a time synchronization relationship across the isolation unit 20, and periodically update the time synchronization relationship during system operation.

[0090] The main controller 10 periodically sends synchronization data frames containing the main controller's timestamp to the local controller 31. The local controller 31 calibrates its local counter based on the synchronization data frames and compensates for the local execution time by incorporating the calibration propagation delay of the isolation unit 20. The transmission time of the synchronization data frames is scheduled outside the transient restricted area. When the scheduled transmission time of the synchronization data frame overlaps with the transient restricted area, the main controller 10 postpones the transmission of the synchronization data frame to the safe communication time slot after the transient restricted area ends.

[0091] When generating a reconfiguration control command, the main controller 10 converts the new trigger phase into a local counter trigger value that can be recognized by the local controller 31, so as to ensure that the reconfiguration control command can be executed within a predetermined safe time domain.

[0092] The communication peripheral module of the main controller 10 encapsulates the timing and duty cycle parameters into a standard asynchronous communication data frame. Before encapsulation, the main controller 10 determines whether the sum of the current time, the data frame transmission time, the propagation delay of the isolation unit 20, and the register loading time of the local controller 31 is earlier than the new trigger phase; if the condition is not met, the main controller 10 abandons the drive instruction of the current cycle or generates a zero-pulse-width safety control instruction.

[0093] The main controller 10 appends a cyclic redundancy check (CRC) code to the end of the data frame. The encapsulated data frame is then sent to the local controller 31 through the isolation channel inside the isolation unit 20.

[0094] S502, the local controller 31 extracts the tail check code and performs cyclic redundancy check verification on the data payload.

[0095] If the verification result is incorrect, the local controller 31 will discard the abnormal data frame and maintain the preset safe pulse width output; the preset safe pulse width is zero pulse width, or the minimum sustaining pulse width below the effective mechanical actuation threshold of the piezoelectric actuator 40.

[0096] Meanwhile, the continuous error counter inside the local controller 31 performs an accumulation count. When the accumulated value of the continuous error counter exceeds the preset safety shutdown threshold, it indicates that the communication link is in a severely disrupted state. The local controller 31 triggers a low-level hardware interrupt, forcibly clears the duty cycle of the internal timer to zero, and cuts off the power output of the drive circuit 32.

[0097] If the verification passes, the local controller 31 clears the continuous error counter and parses the reconstructed equivalent pulse width and the new trigger phase parameters. The PWM timer module inside the local controller 31 is configured with a shadow register mechanism. The local controller 31 writes the parsed reconstructed equivalent pulse width into the shadow comparison register of the timer.

[0098] At the instant the update event triggers at the end of the current PWM cycle, the value in the shadow compare register is automatically loaded into the active compare register by the underlying hardware. This hardware-level synchronous loading mechanism ensures that no drive level glitches caused by timing interruptions occur during the adjustment of the duty cycle parameter.

[0099] The local controller 31 generates a PWM control level based on the updated timer comparison value and outputs it to the driver circuit 32. For the dead time setting and gate drive logic of the half-bridge or full-bridge switching transistor topology, those skilled in the art can perform conventional configurations based on the junction capacitance characteristics of the selected power switching transistor. The underlying hardware driver circuit design is well-known in the field and will not be elaborated upon here.

[0100] S503, the drive circuit 32 periodically turns on and off under the control of the PWM control level, applying the constant power supply voltage amplitude provided by the high-voltage floating platform 30 to the drive port of the piezoelectric actuator 40.

[0101] When the piezoelectric ceramic material inside the piezoelectric actuator 40 receives an electrical energy injection with a duration equal to the reconstruction equivalent pulse width, it undergoes a corresponding inverse piezoelectric deformation.

[0102] Since the reconstructed equivalent pulse width incorporates dual compensation components for time shift loss and electromechanical efficiency distortion caused by high voltage transient interference, the single effective mechanical displacement achieved by the piezoelectric actuator 40 under the condition of a constant upper limit of the driving voltage peak follows the following integral relationship. This integral relationship is based on the current-limited charging, controlled discharging, or quasi-charge control working mode of the drive circuit 32 and is used to characterize the mechanical displacement response caused by the change of the effective injected charge amount with the driving duration under the condition of a constant upper limit of the driving voltage peak.

[0103] The formula is as follows: ; In the formula, The effective mechanical displacement generated by the piezoelectric actuator within a single actuation cycle of 40; To characterize the electromechanical coupling displacement constant representing the conversion from charge injection to physical displacement in the inverse piezoelectric effect, this constant is obtained by comprehensively calibrating the piezoelectric strain coefficient of the piezoelectric material used and the transmission ratio of the mechanical structure, with its value range set at 1.0 × 10⁻⁶. -6 m / C to 1.0×10 -2 m / C (meters per coulomb).

[0104] This range of values ​​covers typical electromechanical conversion efficiencies from micro / nano-scale high-frequency oscillators to large-displacement piezoelectric stack piezoelectric actuators 40. Those skilled in the art can measure the actual mechanical displacement under standard charge injection using a laser interferometer, perform factory calibration on this constant, and write it into the read-only memory of the local controller 31.

[0105] The reconstructed equivalent pulse width of the actual output of the underlying timer; The constant power supply voltage amplitude parameter provided to the high-voltage floating platform 30; This is the real-time equivalent admittance function affected by transient high voltage; This is the time variable for integration.

[0106] Through the aforementioned physical closed-loop operation, the system achieves precise and stable control of the electromechanical actuation system. The slew rate parameters, capacitance time constant, mechanical resonance period, reference steady-state admittance parameters, time compensation parameters, state compensation parameters, protection margin, and maximum waiting period in the above embodiments can all be obtained through factory calibration, reading parameters from the device manual, online sampling calculation, or table lookup, and stored in the non-volatile memory of the main controller 10 or the local controller 31. Based on the aforementioned structural connection relationships, timing avoidance logic, probe detection process, and pulse width reconstruction method, those skilled in the art can achieve dynamic avoidance and synchronization control of the piezoelectric actuator 40 in a high-voltage microfluidic environment without changing the upper limit of the driving voltage peak value.

[0107] To enable those skilled in the art to more clearly understand the purpose, technical solution, and advantages of this invention, the present invention will be further described in detail below with reference to specific application embodiments, real experimental test data, and corresponding drawings. It should be noted that the embodiments described herein are only for explaining the present invention and are not intended to limit the scope of protection of the present invention.

[0108] I. Specific Numerical Application Examples In this embodiment, the high-voltage microfluidic system uses a constant supply voltage amplitude. V. The fundamental mechanical resonant frequency of the piezoelectric actuator 40 is 50kHz, corresponding to the fundamental mechanical resonant period. μs. The main controller 10 is planned to be in At a time of μs, the high-voltage electric field generating unit 50 is instructed to perform a step switching action from 0V to 2000V.

[0109] (1) Transient restricted area delineation calculation The main controller 10 extracts the slew rate parameters of the high-voltage electric field. V / μs, and look up the system equivalent parasitic capacitance in the table. pF and capacitance time constant μs. Substituting into the common-mode transient current decay model, the electromagnetic interference dissipation time is obtained. μs. Sets the digital signal command propagation delay compensation time. μs, timing protection margin μs.

[0110] Based on the formula for the boundary of the restricted area, the initial boundary is calculated. μs; End boundary μs. This defines a transient no-entry zone on the global timeline. .

[0111] (2) Sequence conflict determination and dynamic phase shift The main controller 10 extracts the initial trigger phase of the single drive pulse to be executed. μs, initial reference pulse width μs. Establish the basic working interval. .

[0112] Based on the Boolean intersection condition, the starting node 50.5μs ≤ 51.7μs and the ending node 55.5μs ≥ 49.3μs meet the conflict condition. (Boolean variable...) It is true.

[0113] The main controller 10 calculates the forward advance translation node. μs. If the current clock reading at this moment is 45.0 μs, Earlier than the current time, a causal conflict occurs, causing the system to forcibly allocate the target's safe time domain as a backward delay translation node. μs. The time shift can be calculated from this. μs (i.e., a backward delay of 1.2 μs).

[0114] (3) Probe detection and admittance calculation Take the displacement constraint proportional coefficient The system outputs the detection pulse width. The probe signal is measured in μs (much smaller than the 20μs resonant period, so it does not induce macroscopic displacement). The local controller 31 acquires transient current and derives real-time equivalent admittance data through integration. S. Reference steady-state admittance parameters read from non-volatile memory. S. The admittance offset coefficient is calculated. This indicates that the high voltage electric field increases the equivalent stiffness of the piezoelectric ceramic and decreases its charge absorption capacity.

[0115] (4) Reconstruction of equivalent energy of multidimensional variables Extract the damping coefficient of the system calibration μs, calculate the time compensation factor Extracting piezoelectric strain-corrected gain Calculate the state compensation factor .

[0116] Substitute into the reconstruction equation: μs.

[0117] The main controller 10 ultimately packages and sends out the new trigger phase of 51.7 μs and the reconstructed equivalent pulse width of 6.15 μs. This reconstructing instruction not only avoids the scrambling period of high-voltage discharge, but also precisely compensates for the time loss caused by delay and the increase in stiffness caused by high voltage by extending the power-on time by 1.15 μs, thus ensuring the constant mechanical displacement of a single output.

[0118] II. Experimental Verification and Effect Comparison To verify the effectiveness and reliability of the multidimensional variable dynamic avoidance and synchronization control method of the present invention in a real high-voltage microfluidic environment, a hardware test bench including a high-voltage electric field sorting electrode, an isolated digital communication link and a piezoelectric micropump actuator was built for comparative verification.

[0119] Comparison group settings: Experimental group: The dynamic avoidance and synchronization control system and method of piezoelectric actuator in high-voltage microfluidic environment provided by the present invention are adopted, namely, global activation transient restricted area delineation, time domain overlap determination, dynamic phase shift avoidance, and closed-loop control strategy based on real-time admittance detection and time shift for multi-dimensional variable equivalent energy reconstruction.

[0120] Control group: The traditional high-voltage microfluidic drive scheme is adopted, namely the open-loop constant voltage drive strategy with fixed PWM duty cycle, no timing active avoidance mechanism, and no real-time impedance / admittance detection.

[0121] The test results are as follows: according to Figure 8 It is known that when faced with common-mode transient interference caused by the switching of high-voltage electric field states, the control system of this invention can accurately calculate the interference boundary and delineate the transient no-entry zone (e.g., Figure 8 (As shown in the medium gray shaded area). When the originally planned basic work area ( Figure 8 When the dashed box in the middle collides with the restricted area in the time domain, the experimental group successfully triggered the dynamic phase-shifting logic, shifting the trigger phase backward and generating a safe operating range completely free from the electromagnetic pulse peak. Figure 8 (Solid line frame). Combined with long-term steady-state test data of 100,000 consecutive pulse transmissions at the bottom layer, it is confirmed that: due to the failure to avoid high-voltage jump spikes, the isolation channel communication bit error rate of the control group was as high as 18.4%, and it repeatedly caused the bottom layer drive circuit to get stuck in the old duty cycle state, resulting in fluid overshoot; while the experimental group... Figure 8 Under the protection of the physical timing avoidance mechanism shown, the effective execution rate of instructions reaches 100%, completely eliminating the program dead zone caused by transient electromagnetic interference.

[0122] according to Figure 10It can be seen that as the bias high voltage applied by the external microfluidic chip gradually increases from 0V to 2000V, the real-time equivalent admittance data of the piezoelectric actuator exhibits a significant nonlinear decay trend. This scatter plot characteristic conclusively proves that the high-voltage electric field in space will cause distortion (increased stiffness) of the equivalent mechanical stiffness inside the piezoelectric ceramic through polarization effect, thereby causing a substantial physical decrease in its charge absorption rate and electromechanical conversion efficiency. The control group, lacking probe detection methods, completely lost this underlying physical state change; while the experimental group accurately captured this decay curve, providing an accurate and reliable data source for subsequent calculations of the "admittance offset coefficient" and "state compensation factor".

[0123] according to Figure 9 It can be seen that the system suddenly introduces a high-voltage step change of 2000V at the 50th cycle of the continuous actuation sequence (e.g. Figure 9 As shown by the dashed step line on the upper right Y-axis), the mechanical displacement consistency of the two fluid outputs showed a significant difference. (Control group) Figure 9 (The dotted line with squares in the middle) is not correct. Figure 10 The confirmed stiffness surge, without any compensation and compounded by time loss due to communication congestion, resulted in a severe step-like drop in single effective mechanical displacement, with an output attenuation of up to 21.5%, directly leading to abnormal microparticle pumping within the flow channel; while the experimental group ( Figure 9 The solid line with a circle in the middle) uses an algorithm to obtain the "reconstructed equivalent pulse width" that includes time compensation and state compensation, thus reasonably extending the execution time of the driving pulse (e.g., Figure 8 (The solid frame is longer than the dashed frame), perfectly equivalent to injecting additional charge to counteract stiffness resistance. Ultimately, the mechanical displacement curve of the experimental group remained flat and continuous throughout the entire high-voltage interference range, with fluctuation deviations strictly controlled within an extremely narrow range of ±1.8%. This fully demonstrates that, without changing the amplitude of the hardware driving voltage, this invention achieves excellent anti-interference physical closed-loop control of the piezoelectric microfluidic system solely through time-width dynamic reconstruction in the pure digital domain.

[0124] Although embodiments of the invention have been shown and described, it will be understood by those skilled in the art that various changes, modifications, substitutions and alterations can be made to these embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the appended claims and their equivalents.

Claims

1. A method for dynamic obstacle avoidance and synchronization control of piezoelectric actuators under high-voltage microfluidic conditions, characterized in that, Includes the following steps: Extract transient characteristic parameters and hardware link parameters of the high-voltage electric field generating unit, calculate the attenuation time domain of common-mode transient interference, and delineate transient no-entry zones on the global time axis; The initial control sequence is time-domain overlap with the transient restricted area. If there is a time-domain intersection, a dynamic phase shift operation is performed to avoid the transient restricted area and the resulting time shift is recorded. The local controller sends a probe detection command to the isolation unit, which in turn sends a probe electrical signal to the piezoelectric actuator and obtains the real-time equivalent admittance data to calculate the admittance offset coefficient. Using the time shift and the admittance offset coefficient as input variables, state calculations are performed to reconstruct the equivalent pulse width while keeping the upper limit of the driving voltage peak constant, and a reconstruction control command is generated. The reconfiguration control command is sent to the local controller, which controls the drive circuit to output electrical energy to complete the mechanical drive of the piezoelectric actuator.

2. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 1, characterized in that, Extract transient characteristic parameters and hardware link parameters of the high-voltage electric field generating unit, calculate the attenuation time domain of common-mode transient interference, and delineate transient no-entry zones on the global time axis, specifically including: When a pre-trigger signal indicating that the voltage state of the high-voltage electric field generating unit is about to switch is received, or before the high-voltage electric field generating unit is about to switch voltage state according to the preset schedule, the slew rate parameter is extracted as the transient characteristic parameter, and the capacitance time constant is extracted as the hardware link parameter. Establish a common-mode transient current decay model equation describing the process across isolation units; Substitute the system's preset safe communication threshold current parameter into the common-mode transient current decay model equation, and solve in reverse the time span required for the transient common-mode current prediction value to dissipate to within the safe limit. Define the solution result as the electromagnetic interference dissipation time. Combining the propagation delay of the digital signal of the isolation unit with the preset timing protection margin, a time closed interval consisting of the start time boundary and the end time boundary of the restricted area is determined on the global time axis as the transient restricted area.

3. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 2, characterized in that, The step of determining the temporal overlap between the initial control sequence and the transient restricted area, and performing a dynamic phase-shifting operation when a temporal intersection exists, specifically includes: Extract the initial control sequence containing the initial trigger phase and the initial reference pulse width, establish a basic working interval representing the original execution cycle, and determine whether the start node of the basic working interval is earlier than or equal to the end time boundary of the restricted area, and whether the end node of the basic working interval is later than or equal to the start time boundary of the restricted area. If the determination result is true, then it is determined that there is a temporal intersection, and the forward advance translation node and the backward delay translation node are calculated respectively; Select the node with the smaller absolute value of the time deviation from the initial trigger phase as the target safe time domain for translation, calculate the difference between the target safe time domain and the initial trigger phase as the time translation amount, and translate the initial trigger phase to the target safe time domain for translation as the new trigger phase.

4. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 3, characterized in that, The step of causing the local controller to output a probe electrical signal to the piezoelectric actuator specifically includes: The test pulse width of the probe electrical signal is limited to the lower limit of the basic mechanical resonance period ratio of the piezoelectric actuator; The amount of charge injected in a single probe signal is limited to be lower than the minimum charge threshold required for the piezoelectric actuator to generate a measurable macroscopic displacement, and a charge release interval is set between adjacent probe detections.

5. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 4, characterized in that, The calculation of the admittance offset coefficient from the real-time equivalent admittance data obtained from the return process specifically includes: The real-time equivalent admittance data is derived by performing a definite integral operation in the time dimension on discrete transient response current data and combining it with fixed supply voltage amplitude parameters. Calculate the ratio of the preset reference steady-state admittance parameter to the real-time equivalent admittance data to obtain the admittance offset coefficient; When calculating the ratio, the system-preset lower limit threshold of the dead zone admittance is compared with the real-time equivalent admittance data, and the maximum value is taken as the denominator for the ratio calculation.

6. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 3, characterized in that, The process of reconstructing the equivalent pulse width by using the time shift and the admittance offset coefficient as input variables, while maintaining a constant upper limit for the peak value of the driving voltage, specifically includes: Based on the second-order equivalent electromechanical inertia, the time compensation factor for correcting the time displacement deviation is calculated using the time translation amount and the damping coefficient of the equivalent electromechanical system. The state compensation factor for correcting the power efficiency decay is calculated using the admittance offset coefficient and the piezoelectric strain correction gain. Multiply the time compensation factor by the state compensation factor, and then multiply by the initial reference pulse width to obtain the reconstructed equivalent pulse width; The information containing the reconstructed equivalent pulse width and the new trigger phase is encapsulated to generate the reconstructed control command.

7. The dynamic avoidance and synchronization control method for piezoelectric actuators in a high-voltage microfluidic environment according to claim 6, characterized in that, After obtaining the reconstructed equivalent pulse width, and before generating the reconstructed control command, the process also includes execution cycle timing out-of-bounds verification and compensation margin accumulation: The calculated reconstructed equivalent pulse width is compared with the upper limit tolerance of the currently set PWM running cycle. If the reconstructed equivalent pulse width exceeds the maximum available time slot of the period, it is forcibly truncated to the specific value of the maximum available time slot; Calculate the difference between the reconstructed equivalent pulse width before truncation and the maximum available time slot, store the difference as an uncompensated time margin in the accumulator register, and in the reconstruction calculation of the next PWM running cycle, add the time margin to the newly calculated reconstructed equivalent pulse width.

8. The dynamic avoidance and synchronization control method for piezoelectric actuators in a high-voltage microfluidic environment according to claim 7, characterized in that, After performing the periodic timing out-of-bounds check and before generating the reconstruction control instruction, a secondary check is also performed: The reconstruction working range is re-established based on the currently calculated equivalent pulse width. A secondary check is performed on the reconstructed working interval and the transient restricted area. If the reconstructed working interval and the transient restricted area overlap, the reconstructed equivalent pulse width is truncated according to the maximum safe available time slot. If the truncated reconstructed equivalent pulse width is less than the minimum effective driving pulse width, then the reconstructing control instruction for this cycle will be set to the zero pulse width safety instruction.

9. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 3, characterized in that, Before sending the reconfiguration control command to the local controller, the method further includes: Periodically send synchronization data frames containing timestamps to establish time synchronization relationships across isolation units, and convert the new trigger phase into a local counter trigger value that can be recognized by the local controller; Determine whether the sum of the current time, data frame transmission time, isolation unit propagation delay, and local controller register loading time is earlier than the new trigger phase; if the condition is not met, abandon the current cycle drive instruction or generate a zero-pulse-width safety control instruction.

10. The dynamic avoidance and synchronization control method for piezoelectric actuators under high-voltage microfluidic environment according to claim 1, characterized in that, After the reconfiguration control command is issued to the local controller, the following is also included: The local controller extracts the cyclic redundancy check code appended to the end of the data frame and performs cyclic redundancy check verification. If the verification result contains an error, the abnormal data frame is discarded, the preset safe pulse width output is maintained, and the internal continuous error counter is incremented. When the accumulated value of the continuous error counter exceeds the preset safety shutdown threshold, a low-level hardware interrupt is triggered, forcibly cutting off the power output of the drive circuit; if the verification passes, the reconstructed equivalent pulse width is written into the shadow comparison register of the timer, and automatically loaded into the active comparison register by the hardware at the moment the update event is triggered.

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