Three-phase single-stage isolated bidirectional converter, control method and storage medium

Abstract

This invention provides a three-phase single-stage isolated bidirectional converter, its control method, and a storage medium, relating to the field of power electronic conversion technology. The converter eliminates the traditional intermediate DC bus capacitor and includes a three-phase LC input filter, a three-phase matrix converter, a high-frequency transformer, a full-bridge converter, and a frequency conversion control module. The transformer uses a nanocrystalline iron core, and parameter optimization is performed based on a dynamic loss model combined with the minimum operating frequency and an improved AP method. The control module reconstructs the three-phase voltage envelope and determines the operating sector. Under the assumption of frequency decoupling, it first solves for the duty cycle and phase shift time simultaneously. A normalized constant is constructed based on the piecewise definite integral of the inductor current, and the target switching frequency is analytically derived using power command feedforward. This invention proposes a multi-degree-of-freedom continuous frequency conversion control method, effectively reducing the average switching loss under all operating conditions, further improving the overall conversion efficiency, and achieving efficient and smooth bidirectional energy transmission.

Description

Technical Field

[0001] This invention relates to the field of power electronic conversion technology, and in particular to a three-phase single-stage isolated bidirectional converter, its control method, and its storage medium. Background Technology

[0002] Currently, three-phase isolated bidirectional converter systems generally adopt a two-stage topology of "front-stage PWM rectifier cascaded with rear-stage bidirectional DC / DC converter". However, this type of two-stage architecture has the following technical bottlenecks in practical applications: First, power density is limited and system reliability is insufficient. To achieve energy decoupling and dynamic voltage regulation between the front and rear stages, a large-capacity electrolytic capacitor is usually required on the intermediate DC bus. Large-capacity electrolytic capacitors are bulky and sensitive to operating temperature and high-frequency ripple current, easily becoming a weak link that limits the lifespan and reliability of the entire converter system; Second, the overall energy conversion efficiency is limited. System energy needs to pass through two independent power switching networks in the front and rear stages, resulting in a large number of power semiconductor devices and a significant superposition effect of conduction and switching losses; Third, global system optimization is difficult. The front-stage rectification and rear-stage DC conversion usually adopt mutually decoupled independent control loops, resulting in hysteresis in the system's dynamic response and making it difficult to perform multi-variable global coordinated control for overall system operating conditions (such as input current distortion and transformer losses under wide power fluctuations).

[0003] Therefore, how to eliminate the large and easily failed intermediate DC bus electrolytic capacitor in traditional topologies, design a minimalist single-stage converter topology, and solve the efficiency, loss, and power quality problems of the converter system under wide power fluctuation conditions through control methods are technical challenges that urgently need to be solved in this field. Summary of the Invention

[0004] To address the technical bottlenecks of existing two-stage bidirectional converters, such as large DC bus electrolytic capacitors, limited lifespan, limited overall energy conversion efficiency, and difficulty in achieving global system optimization, this invention provides a three-phase single-stage isolated bidirectional converter, its control method, and storage medium to achieve efficient, high-power-density, and highly reliable bidirectional energy transmission.

[0005] To achieve the above objectives, the present invention adopts the following technical solution:

[0006] In a first aspect, the present invention provides a three-phase single-stage isolated bidirectional converter, including an AC side interface, a three-phase LC input filter, a three-phase matrix converter, a high-frequency transformer, a full-bridge converter, a DC side interface, and a frequency conversion control module that is communicatively connected to both the three-phase matrix converter and the full-bridge converter;

[0007] The AC side interface is used to connect to a three-phase AC power grid; the DC side interface is used to connect to DC side equipment.

[0008] The input terminal of the three-phase LC input filter is connected to the AC side interface, and the output terminal of the three-phase LC input filter is connected to the AC input side of the three-phase matrix converter.

[0009] The primary winding of the high-frequency transformer is connected to the AC output side of the three-phase matrix converter, and the secondary winding of the high-frequency transformer is connected to the AC side of the full-bridge converter.

[0010] The DC side of the full-bridge converter is connected to the DC side interface;

[0011] The frequency conversion control module is configured to: acquire the real-time constant transmission power demand during the bidirectional operation of the system; and based on the real-time constant transmission power demand, dynamically calculate and coordinately adjust the phase shift angle, duty cycle and switching frequency of the switching devices in the three-phase matrix converter and the full-bridge converter, so as to achieve smooth bidirectional energy transmission through multi-degree-of-freedom continuous frequency conversion modulation.

[0012] Furthermore, the specific timing and logic for the frequency conversion control module to perform multi-degree-of-freedom continuous frequency conversion modulation include:

[0013] Based on the current voltage operating conditions of the converter, calculate and determine the phase shift angle and duty cycle of the switching devices;

[0014] Based on the obtained phase shift angle and duty cycle, the switching frequency is used as a global variable to analyze and calculate the overall transmission power level of the matching system.

[0015] Through the coordination of the phase shift angle, duty cycle, and switching frequency, the discontinuous chopping current generated on the AC input side of the three-phase matrix converter is buffered and filtered by the capacitor module of the three-phase LC input filter, and then outputs a continuous and smooth grid-side current to the AC side interface.

[0016] Furthermore, the three-phase LC input filter includes a filter inductor module and a filter capacitor module; the single-phase capacitance value C of the filter capacitor module satisfies the following constraint formula:

[0017]

[0018] In the formula, P N λ represents the rated active power of the converter system; λ represents the ratio of the maximum allowable fundamental reactive power to the rated active power; ω e Let ω represent the fundamental angular frequency of a three-phase AC power grid. e = 2*π*f e fe U is the fundamental frequency of the power grid; rms This represents the effective value of the phase voltage in a three-phase AC power grid.

[0019] The single-phase inductance value L of the filter inductor module satisfies the following constraint formula:

[0020]

[0021] In the formula, U dc The rated DC voltage for operation of the DC-side interface; f s_min The lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module; ΔI max To meet the maximum allowed peak-to-peak value of high-frequency ripple current in accordance with the power quality standards for grid connection.

[0022] Furthermore, the core of the high-frequency transformer is made of nanocrystalline soft magnetic material, and the geometric parameters of the core and the number of winding turns of the high-frequency transformer are optimized and configured through a dynamic loss separation model based on frequency conversion conditions.

[0023] The dynamic loss separation model defines the total core loss P under variable frequency non-sinusoidal excitation. core Physically decomposed into hysteresis loss P h Eddy current loss P e and residual loss P r ;

[0024] The core geometry parameters and winding turns of the high-frequency transformer are determined by optimization using the area AP method combined with the dynamic loss separation model.

[0025] Furthermore, the effective cross-sectional area A of the core of the high-frequency transformer... e With window area A w The product of these values ​​is the AP value, which is designed based on the minimum operating frequency set by the system and satisfies the following constraint formula:

[0026]

[0027] In the formula, P t K represents the maximum apparent power that the high-frequency transformer needs to transmit. f K represents the waveform coefficients of the non-sinusoidal excitation voltage generated by the matrix converter under high-frequency chopping. u The window utilization factor of the transformer winding; J is the set rated current density of the conductor; f s_min B is the lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module. m The maximum operating magnetic flux density of the high-frequency transformer is determined by ensuring the total core loss P. core And, under the premise that the temperature rise is within the allowable range of safe heat dissipation, the optimal upper limit of magnetic flux density is obtained iteratively;

[0028] The equivalent series inductance parameter L of the high-frequency transformer f The leakage inductance is achieved through the equivalent leakage inductance generated by the primary and secondary winding structure design of the high-frequency transformer.

[0029] Furthermore, when the frequency conversion control module calculates the phase shift angle and duty cycle, it first acquires the actual operating voltage u on the DC side of the full-bridge converter in real time. p Real-time instantaneous phase voltage values ​​on the three-phase AC grid side and the equivalent leakage inductance parameter L of the high-frequency transformer. f The instantaneous values ​​of the three-phase AC phase voltages are sorted by amplitude to reconstruct the maximum envelope voltage vv. max Intermediate envelope voltage vv mid Minimum envelope voltage vv min And the maximum line voltage U is calculated. max =vv max -vv min First intermediate line voltage U mid1 =vv max -vv mid and the second intermediate line voltage U mid2 =vv mid -vv min ;

[0030] Based on the instantaneous voltage difference across the equivalent leakage inductance of a high-frequency transformer, a piecewise linear inductor current recursive model is constructed within half a high-frequency switching cycle T / 2. The initial node current within the half-cycle is set as i1, and the final node current is set as i4, combined with the DC-side high-frequency square wave voltage u. p The set of discrete node recursive formulas for inductor current is as follows:

[0031]

[0032] In the formula, t1 is the duty cycle time, t2 is the phase shift time, T is the high-frequency switching period, and U is the phase shift time. max To measure the maximum line voltage of the converter in AC mode, U mid The first intermediate line voltage of the AC converter is measured, i2 is the second node current, i3 is the third node current, and L is the inductance value of the series inductor.

[0033] By performing definite integral operations on the instantaneous current function of each segmented interval within half a high-frequency switching cycle, the charge integral IP of each interval is obtained. j1 IP j2 IP j3 The corresponding system of definite integral equations is:

[0034]

[0035] In the formula, t represents time.

[0036] Furthermore, the frequency conversion control module compares the first intermediate line voltage U mid1 With the second intermediate line voltage U mid2 Based on the relationship between the magnitudes, determine the current operating sector of the system, and simultaneously solve for the duty cycle time t1 and phase shift time t2 under the corresponding sector;

[0037] When determining U mid1 >=U mid2 At that time, the frequency converter control module invokes a preset first set of steady-state equations, which are specifically:

[0038]

[0039] In the formula, i max1 i mid1 i min1 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically:

[0040]

[0041] When determining U mid1 mid2 At that time, the frequency converter control module calls the preset second steady-state equation set for replacement calculation, the second steady-state equation set being as follows:

[0042]

[0043] In the formula, i max2 i mid2 i min2 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically:

[0044] .

[0045] Furthermore, the frequency conversion control module has an internal mapping model between the system transmission power and the switching frequency. The mapping model achieves the inverse solution of the target switching frequency through a feedforward analytical step.

[0046] Based on the charge integral, the frequency conversion control module derives the equivalent average current i on the DC side. d Its derivation formula is:

[0047]

[0048] The equivalent average current i on the DC side d It is directly proportional to the high-frequency switching period T and related to the target switching frequency f. s ​They are inversely proportional;

[0049] The frequency converter control module acquires the bidirectional operation real-time constant transmission power demand command P issued by the external system. ref And combined with the DC side high-frequency square wave voltage u p Construct a method for directly solving the target switching frequency f. s Analytical power equation:

[0050]

[0051] The frequency conversion control module substitutes the calculated duty cycle time t1 and phase shift time t2 into the analytical power equation, and replaces T with 1 / f. s The unique target switching frequency f is determined through a feedforward analytical step. s ; and combined with t1, t2 and the target switching frequency f s Generate PWM pulses to drive the three-phase matrix converter and the full-bridge converter.

[0052] The feedforward analysis step specifically includes:

[0053] The system obtains the current DC side actual voltage, three-phase AC side actual voltage, and high-frequency transformer leakage inductance parameters, and calculates the current phase shift angle and duty cycle parameters of the modulation strategy based on the system operating conditions.

[0054] Based on the actual voltage of the three-phase AC side, the actual voltage of the DC side, the leakage inductance parameters, and the current phase shift angle and duty cycle parameters, a frequency-normalized current constant that is completely decoupled from the switching frequency is constructed.

[0055] Multiply the frequency-normalized current constant by the actual DC-side voltage to obtain the system's frequency-normalized power constant;

[0056] Using the frequency-normalized power constant as the numerator and the real-time constant transmission power requirement as the denominator, a division operation is performed to obtain the target switching frequency that enables the system to achieve the real-time constant transmission power requirement through feedforward analytical solution.

[0057] Secondly, the present invention provides a multi-degree-of-freedom continuous frequency conversion control method, applied to the three-phase single-stage isolated bidirectional converter as described above. The control method is executed by a frequency conversion control module and includes:

[0058] Obtain the real-time constant transmission power requirement during the bidirectional operation of the system;

[0059] Based on the real-time constant transmission power requirement, the state of the switching devices in the three-phase matrix converter and the full-bridge converter is dynamically calculated and coordinated in the order of first determining the phase shift angle and duty cycle, and then analyzing the target switching frequency.

[0060] The converter is driven by multi-degree-of-freedom continuous frequency modulation, so that the discontinuous current on the input side of the converter is filtered to achieve smooth bidirectional energy transmission.

[0061] Thirdly, the present invention provides a computer-readable storage medium having a computer program stored thereon, characterized in that the computer program, when executed by a processor, implements the multi-degree-of-freedom continuous frequency conversion control method as described above.

[0062] The three-phase single-stage isolated bidirectional converter, control method, and storage medium provided by this invention have at least the following beneficial effects:

[0063] (1) Topology physical level: The present invention adopts a single-stage isolated conversion architecture, which eliminates the large and easily failed intermediate DC bus electrolytic capacitor in the traditional two-stage topology, significantly improving the power density and long-term operational reliability of the system; at the same time, it reduces the number of power conversion stages and reduces multi-stage conduction losses.

[0064] (2) Control Mechanism Level: The multi-degree-of-freedom continuous frequency conversion modulation strategy proposed in this invention, by first determining the phase shift angle and duty cycle and then analyzing the execution logic of the switching frequency, enables the discontinuous chopping current on the input side to be made continuous only by passing through a conventional three-phase LC filter, effectively suppressing grid-side current distortion (THD) and simplifying the hardware complexity of the input filter. At the same time, since the switching frequency is dynamically adjusted within a set range according to the actual power demand, it avoids the system from maintaining the highest switching frequency for a long time under non-full load conditions, thereby effectively reducing the average switching loss under all operating conditions and further improving the overall conversion efficiency. In addition, this multi-degree-of-freedom continuous frequency conversion control method has the ability to adjust the phase of the grid-side AC voltage and current, and can achieve high power factor operation of the system.

[0065] (3) Parameter calculation level: A feedforward analytical solution model based on the frequency normalization constant was constructed, which transforms the multivariable coupled transmission power equation into a direct division operation for inverse solution. This feedforward analytical method avoids the time delay caused by frequency iteration in conventional closed-loop control and greatly improves the converter's dynamic tracking speed to power command changes.

[0066] (4) Magnetic component level: The high-frequency transformer uses nanocrystalline soft magnetic material and combines dynamic loss separation model to optimize parameters under non-sinusoidal frequency conversion conditions, effectively suppressing core loss in a wide frequency range and ensuring high conversion efficiency of the system under all operating conditions.

[0067] Additional aspects and advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. Attached Figure Description

[0068] The accompanying drawings, which are incorporated in and form part of this specification, illustrate embodiments consistent with the invention and, together with the description, serve to explain the principles of the invention.

[0069] Figure 1 A schematic diagram of the system topology of a three-phase single-stage isolated bidirectional converter provided in an embodiment of the present invention;

[0070] Figure 2 A schematic diagram of the multi-degree-of-freedom continuous frequency conversion control system provided in an embodiment of the present invention;

[0071] Figure 3 This is a flowchart of the analytical reverse solution for the target switching frequency feedforward provided in an embodiment of the present invention;

[0072] Figure 4 A schematic diagram of the modulation pulse timing for the multi-degree-of-freedom continuous frequency conversion control method provided in this embodiment of the invention;

[0073] Figure 5 This is a schematic diagram of the soft-switching operation of the system under rated power provided in an embodiment of the present invention; wherein, (a) is S 3a (a) is a schematic diagram of soft switching operation at the turn-on time; (b) is S 3b Schematic diagram of soft switching operation at the start-up time; (c) is S 3c A schematic diagram of soft switching operation at the start-up time; (d) is S 3d A schematic diagram of the soft switch operation at the activation moment;

[0074] Figure 6 The voltage / current waveform of phase A of the system provided in this embodiment of the invention at rated power;

[0075] Figure 7 The three-phase current waveform diagram of the system under rated power is provided for an embodiment of the present invention.

[0076] The accompanying drawings have illustrated specific embodiments of the invention, which will be described in more detail below. These drawings and descriptions are not intended to limit the scope of the invention in any way, but rather to illustrate the concept of the invention to those skilled in the art through reference to specific embodiments. Detailed Implementation

[0077] Exemplary embodiments will now be described in detail, examples of which are illustrated in the accompanying drawings. When the following description relates to the drawings, unless otherwise indicated, the same numerals in different drawings denote the same or similar elements. The embodiments described in the following exemplary embodiments do not represent all embodiments consistent with the present invention. Rather, they are merely examples of apparatuses and methods consistent with some aspects of the invention as detailed in the appended claims.

[0078] It should be noted that in the embodiments of the present invention, certain software, components, models and other existing solutions in the industry may be mentioned. These should be regarded as exemplary and are only intended to illustrate the feasibility of implementing the technical solution of the present invention. However, they do not mean that the inventor has used or necessarily used the solution.

[0079] The technical solution of the present invention and how the technical solution of the present invention solves the above-mentioned technical problems are described in detail below with specific embodiments. These specific embodiments can be combined with each other, and the same or similar concepts or processes may not be described again in some embodiments. The embodiments of the present invention will now be described with reference to the accompanying drawings.

[0080] This invention provides a three-phase single-stage isolated bidirectional converter. Figure 1 This is a schematic diagram of the system topology of a three-phase single-stage isolated bidirectional converter provided in an embodiment of the present invention. Figure 1 As shown, this three-phase single-stage isolated bidirectional converter eliminates the bulky and easily failed intermediate DC bus electrolytic capacitor found in traditional two-stage topologies, adopting a single-stage isolated converter architecture. The converter mainly includes: an AC side interface 10, a three-phase LC input filter 20, a three-phase matrix converter 30, a high-frequency transformer 40, a full-bridge converter 50, a DC side interface 60, and a frequency conversion control module 70. The AC side interface is used to connect to the three-phase AC power grid to obtain real-time AC voltage and current. The input terminal of the three-phase LC input filter is connected to the AC side interface for energy buffering and continuous filtering of the input current. The AC input side of the three-phase matrix converter is connected to the output terminal of the three-phase LC input filter. This matrix converter consists of 6 sets of bidirectional switch arrays, allowing any two phases of AC power to be directly connected to the subsequent stage at high frequency. The primary winding of the high-frequency transformer is connected to the AC output side of the three-phase matrix converter, and its secondary winding is connected to the AC side of the full-bridge converter to achieve electrical isolation and voltage matching between the AC and DC sides. The DC side of the full-bridge converter is connected to the DC side interface, and then connected to the DC side equipment. The frequency conversion control module is communicatively connected to the three-phase matrix converter and the full-bridge converter, respectively, to send multi-degree-of-freedom continuous frequency conversion control signals to drive the converter operation.

[0081] Furthermore, Figure 1 S in 1a -S 5d These are all identifiers for the switching transistors and their corresponding drivers, used to represent the corresponding switching transistors, their drive signals, and drive units in the three-phase matrix converter 30 and the full-bridge converter 50; HFT represents the high-frequency transformer 40; C represents the AC side filter capacitor in the three-phase LC input filter 20; L fThis represents the AC-side filter inductor in the three-phase LC input filter 20. Thus, the three-phase AC power input from the AC-side interface 10 passes through L... f The AC-side LC filter network formed by C then enters the three-phase matrix converter 30. The three-phase matrix converter 30 and the full-bridge converter 50 are then processed by S. 1a -S 5d The identified switching transistor operates according to multi-degree-of-freedom continuous frequency conversion PWM pulses under the corresponding drive action, and completes isolated energy transfer through HFT.

[0082] In some implementations, the key parameters of the three-phase LC input filter and high-frequency transformer in this three-phase single-stage isolated bidirectional converter are specifically designed based on the characteristics of the multi-degree-of-freedom continuous frequency conversion control method. Specifically, the three-phase LC input filter includes a filter inductor module and a filter capacitor module. Its parameter configuration is based on the minimum operating frequency of the system set when the switching frequency follows dynamic power adjustment, and is matched with the power quality requirements of the three-phase AC grid. The upper limit of the single-phase capacitance value C of the filter capacitor module is constrained by the fundamental reactive power of the system. To ensure a high power factor for the converter under light load or no-load conditions, the capacitance value must satisfy the following constraint formula:

[0083]

[0084] In the formula, P N λ represents the rated active power of the converter system; λ represents the ratio of the maximum allowable fundamental reactive power to the rated active power; ω e Let ω represent the fundamental angular frequency of a three-phase AC power grid. e = 2*π*f e f e U is the fundamental frequency of the power grid; rms This represents the effective value of the phase voltage in a three-phase AC power grid.

[0085] In some implementations, the lower limit of the single-phase inductance value L of the filter inductor module is mainly limited by the high-frequency harmonic current ripple injected into the power grid by the three-phase matrix converter under the worst operating conditions. Furthermore, since this invention employs a multi-degree-of-freedom continuous frequency conversion control method, and the switching frequency is dynamically changing, it is necessary to use the lowest operating switching frequency f set within the frequency conversion range. s_min As a basis for calculation, the inductance value must satisfy the following constraint formula:

[0086]

[0087] In the formula, U dc The rated DC voltage for operation of the DC-side interface; f s_minThe lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module; ΔI max To meet the maximum allowed peak-to-peak value of high-frequency ripple current in the grid-connected power quality standards, the above parameters are configured in a coordinated manner to ensure that the discontinuous chopper current generated on the AC input side of the three-phase matrix converter, after being buffered by the filter capacitor module of the three-phase LC input filter, outputs a continuous and smooth grid-side current to the AC side interface.

[0088] In some implementations, the core of the high-frequency transformer uses nanocrystalline soft magnetic material. Because the multi-degree-of-freedom continuous frequency conversion control method of this invention causes the high-frequency transformer to operate under variable-frequency and non-sinusoidal high-frequency voltage excitation conditions for extended periods, traditional soft magnetic materials struggle to balance high saturation magnetic flux density with low high-frequency losses. Nanocrystalline soft magnetic materials possess high permeability and excellent low-frequency loss characteristics, effectively adapting to this complex electromagnetic condition. The core geometry and winding turns of this high-frequency transformer are determined through optimization using a dynamic loss separation model under variable-frequency conditions. Specifically, the dynamic loss separation model determines the total core loss P under variable-frequency non-sinusoidal excitation. core Physically decomposed into hysteresis loss P h Eddy current loss P e and residual loss P r Its mathematical expression is: P core =P h +P e +P r During the transformer parameter optimization process, this dynamic loss separation model is used to accurately evaluate the actual loss within a wide frequency range, and the product of the magnetic circuit cross-sectional area and window area of ​​the high-frequency transformer is iteratively adjusted accordingly. Furthermore, to further improve the overall power density of the converter system, the equivalent series inductance parameter L used for energy transfer in the control model in this embodiment... f The leakage inductance is achieved directly through the equivalent leakage inductance generated by the primary and secondary winding structure design of the high-frequency transformer, thereby eliminating the need for an external physical series inductor in terms of size and weight.

[0089] In some embodiments, based on the accurate loss assessment using the aforementioned dynamic loss separation model, a design formula for the high-frequency transformer area AP method adapted to the multi-degree-of-freedom continuous frequency conversion control method is further given. Since the switching frequency of this converter is dynamically adjusted following the power, to prevent core saturation of the high-frequency transformer under the worst combination of voltage and frequency, the effective cross-sectional area A of its core is... e With window area A w The product (i.e., the AP value) needs to be designed based on the minimum operating frequency set by the system. The specific constraint formula is as follows:

[0090]

[0091] In the above formula, P t K represents the maximum apparent power that the high-frequency transformer needs to transmit. f K represents the waveform coefficients of the non-sinusoidal excitation voltage generated by the matrix converter under high-frequency slicing. u The window utilization factor of the transformer winding; J is the set rated current density of the conductor; f s_min This refers to the lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module. Specifically, B in the formula... m The maximum operating magnetic flux density of the high-frequency transformer is not a fixed empirical value, but is limited by the aforementioned dynamic loss separation model: that is, while ensuring the total core loss P of the transformer... core And, under the premise that the temperature rise is within the allowable range of safe heat dissipation, the optimal upper limit of magnetic flux density is obtained iteratively. Through this improved AP method formula, the physical properties of nanocrystalline soft magnetic materials can be accurately matched with the wide-frequency operation requirements of variable frequency multi-degree-of-freedom control, so as to achieve high power density and lightweight of the converter system while avoiding magnetic saturation under extreme operating conditions.

[0092] In some implementations, based on the aforementioned hardware topology and device parameter configuration, the frequency converter control module employs a multi-degree-of-freedom continuous frequency conversion control method to overcome the problems of excessive circulating current and limited soft-switching range in traditional fixed-frequency control under wide voltage and light load conditions. The core logic of this control method is executed sequentially, first calculating the phase shift angle and duty cycle based on the steady-state model, and then analyzing the target switching frequency using feedforward.

[0093] For example, the frequency conversion control module integrates a multi-degree-of-freedom continuous frequency conversion control system, which is used to execute a multi-degree-of-freedom continuous frequency conversion control method. Figure 2 The schematic diagram of a multi-degree-of-freedom continuous frequency conversion control system is shown below. Figure 2 As shown, the multi-degree-of-freedom continuous frequency conversion control system includes a voltage extreme value reconstruction module, a sector determination module, a duty cycle and phase shift angle calculation module, and a frequency analysis module; the voltage extreme value reconstruction module is used to obtain the three-phase input voltage U. a U b U c The voltage extreme value reconstruction module outputs the maximum line voltage U. max First intermediate line voltage U mid1 Second intermediate line voltage U mid2 To the sector determination module; the sector determination module outputs U max and U mid The duty cycle and phase shift angle calculation module combines the input u pThe duty cycle time t1 and phase shift time t2 are output to the frequency analysis module; the frequency analysis module receives the external real-time constant transmission power demand command and outputs the duty cycle time t1, phase shift time t2, and target switching frequency f. s The signal is then sent to the PWM signal generator; the PWM signal generator ultimately outputs a drive signal to the switching devices of the three-phase matrix converter and the full-bridge converter.

[0094] like Figure 3 The diagram shown is a flowchart of the analytical reverse solution process for the target switching frequency feedforward provided in an embodiment of the present invention. An exemplary analytical reverse solution process for the target switching frequency feedforward is as follows: The first step is to sample the three-phase AC voltage U. a U b U c DC side actual voltage u p The second step is to calculate the maximum line voltage U. max First intermediate line voltage U mid1 Second intermediate line voltage U mid2 The third step is to execute the judgment condition: U mid1 >=U mid2 If the judgment result is negative, the second steady-state equation set is invoked; if the judgment result is positive, the first steady-state equation set is invoked. The fourth step is to calculate the duty cycle time t1 and phase shift time t2. The fifth step is to calculate the target switching frequency f based on the given power. s .

[0095] Specifically, the frequency converter control module first acquires the actual operating voltage u on the DC side of the full-bridge converter in real time. p Real-time instantaneous phase voltage values ​​on the three-phase AC grid side and the equivalent leakage inductance parameter L of the high-frequency transformer. f Within each high-frequency control cycle, the frequency converter control module sorts the instantaneous values ​​of the acquired three-phase AC phase voltages by amplitude, extracts the maximum phase voltage, intermediate phase voltage, and minimum phase voltage at the current moment, and reconstructs three envelope extreme value variables for modulation calculation: the maximum envelope voltage vv. max Intermediate envelope voltage vv mid and minimum envelope voltage vv min Based on this, the control module further calculates three voltage difference variables characterizing the three-phase input line voltage features, namely: maximum line voltage U max =vv max -vv min First intermediate line voltage U mid1 =vv max -vv mid and the second intermediate line voltage U mid2 =vv mid -vv min .

[0096] After extracting and reconstructing the real-time voltage variables, the frequency converter control module enters the analytical solution stage for the duty cycle time t1 and phase shift time t2. In some implementations, to establish an accurate analytical model, the control module is based on the equivalent leakage inductance L of a high-frequency transformer. f The instantaneous voltage difference between the two ends is used to construct a piecewise linear inductor current recursive model within half a high-frequency switching cycle T / 2. The initial node current within the half-cycle is set as i1, and the final node current as i4, combined with the DC-side high-frequency square wave voltage u. p The set of discrete node recursive formulas for the inductor current is expressed as follows:

[0097]

[0098] In the formula, t1 is the duty cycle time, t2 is the phase shift time, T is the high-frequency switching period, and U is the phase shift time. max To measure the maximum line voltage of the converter in AC mode, U mid The first intermediate line voltage of the AC converter is measured, i2 is the second node current, i3 is the third node current, and L is the inductance value of the series inductor.

[0099] Based on the half-wave symmetry constraint of the steady-state operation of the AC system, satisfying the boundary condition i1 + i4 = 0, the analytical expression for the inductor current can be obtained.

[0100] After obtaining the inductor currents of each discrete node, in order to accurately calculate the equivalent average current of the three-phase side and the DC side, the frequency converter control module performs definite integral calculations on the instantaneous current functions of each segmented interval within half a high-frequency switching cycle, and obtains the charge integral IP of each interval. j1 To IP j3 The specific system of integral equations is listed below:

[0101]

[0102] Due to the intermediate phase voltage vv of the three-phase AC system mid The voltage will alternately deflect towards the maximum or minimum value according to the phase of the power grid. The frequency converter control module compares the first intermediate line voltage U mid1 With the second intermediate line voltage U mid2 The system determines the current operating sector based on the magnitude of the equations and calls the corresponding steady-state analytical equations for that sector.

[0103] When judging U mid1 >=U mid2 At this point, when the intermediate phase voltage is closer to the minimum phase voltage, the control module invokes the preset first steady-state equation set. This equation set is composed of the three-phase power / charge balance equation and the transformer primary and secondary volt-second balance equation, specifically:

[0104]

[0105] Among them, i max1 i mid1 i min1 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically:

[0106]

[0107] Conversely, when judging U mid1 mid2 When the intermediate phase voltage is closer to the maximum phase voltage, the control module calls the preset second steady-state equation set for replacement calculation. This equation set is as follows:

[0108]

[0109] Among them, i max2 i mid2 i min2 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically:

[0110]

[0111] The frequency converter control module solves the above equations simultaneously to obtain the duty cycle time t1 and phase shift time t2 for the current sector.

[0112] After accurately acquiring the time variables t1 and t2, the control module enters the target switching frequency f. s The feedforward analysis steps are as follows. First, based on the aforementioned charge integral, the control module derives the equivalent average current i on the DC side. d Its derivation formula is:

[0113]

[0114] In the above model, due to the integral of each charge IP jn All of these are obtained by integrating over time t, and their internal components necessarily contain a squared term of the high-frequency switching period T (i.e., T). 2 Therefore, i d After removing and simplifying T from the formula, the equivalent average current i on the DC side d Mathematically, it has a strict direct proportional relationship with the switching period T, and can be equivalently converted to a relationship with the switching frequency f. s They are inversely proportional (i.e., T=1 / f) s ).

[0115] Based on this characteristic, the frequency converter control module obtains the bidirectional operation real-time constant transmission power demand command P issued by the external system. ref And combined with the actual DC side voltage, i.e., the DC side high-frequency square wave voltage u​p Construct a method for directly solving the target switching frequency f. s Analytical power equation:

[0116]

[0117] By substituting the obtained t1 and t2 (normalized time ratios) into the above formula, and replacing T with 1 / f s The control module can then directly feedforward to determine the unique target switching frequency f. s Ultimately, the control module combines the time variables t1 and t2 with the target switching frequency f. s PWM pulses are generated to drive the three-phase matrix converter and the full-bridge converter, thereby achieving smooth single-stage bidirectional power transmission between the AC and DC sides under frequency conversion conditions.

[0118] Furthermore, Figures 4 to 7 This paper illustrates the typical implementation effect of a three-phase single-stage isolated bidirectional converter system according to an embodiment of the present invention when employing a multi-degree-of-freedom continuous frequency conversion cooperative control method. The system refers to a converter system composed of an AC side interface 10, a three-phase LC input filter 20, a three-phase matrix converter 30, a high-frequency transformer 40, a full-bridge converter 50, a DC side interface 60, and a frequency conversion control module 70. Its operating waveforms are used to illustrate the coordination relationship between the above topology and the control method.

[0119] like Figure 4 As shown, the modulation pulse timing of multi-degree-of-freedom continuous frequency conversion coordinated control is based on the target switching frequency f. s The high-frequency switching period T is determined, and within this period, the duty cycle time t1 and phase shift time t2 are used to jointly determine the turn-on and turn-off times of each switch and its drive in the three-phase matrix converter 30 and the full-bridge converter 50. This timing sequence is consistent with the aforementioned timing sequence based on U. max U mid1 U mid2 u p and P ref The t1, t2 and f obtained from the analysis s Correspondingly, this indicates that the frequency converter control module 70 can convert the steady-state analysis results into an executable switching drive sequence, thereby realizing single-stage isolated bidirectional power transmission.

[0120] like Figure 5 As shown, the four sub-figures respectively illustrate the operation of the switching transistor S under rated power conditions. 3a S 3b S 3c and S 3d The waveforms of the soft switch driven by the corresponding driver are shown, where the upper left, upper right, lower left, and lower right sub-figures correspond to S respectively. 3a S 3bS 3c and S 3d In each sub-diagram, the dashed circles indicate the commutation range of the corresponding switching transistor. As can be seen from the diagram, in S... 3a S 3b S 3c and S 3d Before the drive signal effectively flips, the voltage across the corresponding switching transistor has dropped to zero or near zero. Then, a drive pulse is applied to turn the transistor on, indicating that the transistor can be turned on under near-zero voltage conditions. Simultaneously, during commutation, the current flowing through the switching transistor transitions continuously without significant current abrupt changes or spikes. Specifically, Figure 5 (a) indicates S 3a It meets the zero-voltage turn-on condition at its commutation moment. Figure 5 (b) indicates that S 3b Near-zero voltage turn-on is also achieved in the corresponding converter section. Figure 5 (c) indicates that S 3c The voltage and current are smoothly connected before and after the driving action. Figure 5 (d) indicates that S 3d It can maintain soft-switching commutation characteristics even at rated power. Therefore, it can be seen that in the three-phase matrix converter 30, S... 3a -S 3d Both the switching transistor and its corresponding driver can achieve soft-switching operation under the action of multi-degree-of-freedom continuous frequency conversion and PWM pulse.

[0121] like Figure 6 As shown, the waveform phase relationship between phase A voltage and phase A current is stable under rated power. The phase A current, after being processed by the three-phase LC input filter 20, exhibits a continuous and smooth change, indicating that even after removing the intermediate DC bus electrolytic capacitor, the converter can still generate a grid-connectable continuous current on the AC side. Furthermore, as... Figure 7 As shown, the amplitudes of the three-phase current waveforms under rated power are basically balanced and the phases differ by about 120 degrees sequentially. This indicates that the single-stage isolation topology and multi-degree-of-freedom continuous frequency conversion coordinated control method of the present invention can maintain the symmetry and stability of the three-phase input current. This verifies the feasibility of the present invention under rated power operation and demonstrates the technical effects of expanding the soft switching range, improving energy transmission efficiency, and reducing the size of intermediate energy storage devices.

[0122] Based on the same inventive concept, this invention also provides a multi-degree-of-freedom continuous frequency conversion control method applied to the aforementioned three-phase single-stage isolated bidirectional converter. This method is executed by the aforementioned frequency conversion control module (such as a digital signal processor (DSP) or microcontroller (MCU), and its overall control timing strictly follows the multi-degree-of-freedom decoupling logic of "first determining the phase shift angle and duty cycle, then analyzing the target switching frequency." Specifically, the method mainly includes the following steps: First, the control module acquires the constant transmission power demand command during the bidirectional operation of the system in real time, and simultaneously collects the actual DC side voltage and the real-time instantaneous phase voltage values ​​of the three-phase AC grid side; second, the instantaneous phase voltage values ​​of the three-phase AC are sorted by amplitude to reconstruct the maximum, intermediate, and minimum envelope voltage variables, and the first intermediate line voltage U is calculated accordingly. mid1 With the second intermediate line voltage U mid2 Next, by comparing U mid1 with U mid2 The relative magnitudes of the current and current sectors are used to determine the current real-time operating sector of the system. Then, based on the sector determination result, the matching system steady-state analytical equations (i.e., the aforementioned constraint equations including charge balance and volt-second balance) are invoked. Under the assumption of frequency decoupling and normalization, the duty cycle time t1 and phase shift time t2 required by the switching devices in the three-phase matrix converter and the full-bridge converter are first solved simultaneously. Based on this, the equivalent average current function on the DC side is derived from the piecewise definite integral of the inductor current within the high-frequency cycle, and substituted into the analytical power equation containing the constant transmission power demand command, directly feedforward to analytically determine the target switching frequency f. s Finally, based on the dynamically calculated t1, t2, and f... s This method generates a PWM modulation signal to drive the converter. Through this method, the discontinuous chopping current on the converter input side can be smoothly connected to the grid after filtering, while effectively expanding the soft-switching operating range of the system under wide voltage range and light load conditions, thus improving the overall energy transfer efficiency.

[0123] Based on the same inventive concept, embodiments of the present invention also provide a computer-readable storage medium storing a computer program (or instructions). The computer-readable storage medium may be a read-only memory (ROM), random access memory (RAM), electrically erasable programmable read-only memory (EEPROM), optical disc storage, or magnetic storage device, etc. When the computer program is read and executed by the processor of a computer device, it can realize the multi-degree-of-freedom continuous frequency conversion control method and its underlying mathematical logic described in the above embodiments of the present invention. For example, by executing the program, the processor can complete various appropriate actions and processes such as real-time voltage variable reconstruction, intelligent determination of operating sectors, simultaneous solution of steady-state equations, and feedforward analytical calculation of the target switching frequency. Through this hardware and software combined implementation method, the present invention embeds complex nonlinear frequency conversion analytical algorithms into the underlying execution logic, thereby ensuring high dynamic response and high reliability of the converter system during grid-connected and off-grid, rectification and inversion bidirectional operation.

[0124] The above embodiments are only used to illustrate the present invention and are not intended to limit the present invention. Those skilled in the art can make various changes and modifications without departing from the spirit and scope of the present invention. Therefore, all equivalent technical solutions also fall within the scope of the present invention, and the patent protection scope of the present invention should be defined by the claims.

Claims

1. A three-phase single-stage isolated bidirectional converter, characterized in that, It includes an AC side interface, a three-phase LC input filter, a three-phase matrix converter, a high-frequency transformer, a full-bridge converter, a DC side interface, and a frequency conversion control module that is communicatively connected to both the three-phase matrix converter and the full-bridge converter; The AC side interface is used to connect to a three-phase AC power grid; the DC side interface is used to connect to DC side equipment. The input terminal of the three-phase LC input filter is connected to the AC side interface, and the output terminal of the three-phase LC input filter is connected to the AC input side of the three-phase matrix converter. The primary winding of the high-frequency transformer is connected to the AC output side of the three-phase matrix converter, and the secondary winding of the high-frequency transformer is connected to the AC side of the full-bridge converter. The DC side of the full-bridge converter is connected to the DC side interface; The frequency conversion control module is configured to: acquire the real-time constant transmission power demand during the bidirectional operation of the system; and based on the real-time constant transmission power demand, dynamically calculate and coordinately adjust the phase shift angle, duty cycle and switching frequency of the switching devices in the three-phase matrix converter and the full-bridge converter, so as to achieve smooth bidirectional energy transmission through a multi-degree-of-freedom continuous frequency conversion control method.

2. The three-phase single-stage isolated bidirectional converter according to claim 1, characterized in that, The specific timing and logic for the frequency conversion control module to perform multi-degree-of-freedom continuous frequency conversion modulation include: Based on the current voltage operating conditions of the converter, calculate and determine the phase shift angle and duty cycle of the switching devices; Based on the obtained phase shift angle and duty cycle, the switching frequency is used as a global variable to analyze and calculate the overall transmission power level of the matching system. Through the coordination of the phase shift angle, duty cycle, and switching frequency, the discontinuous chopping current generated on the AC input side of the three-phase matrix converter is buffered and filtered by the capacitor module of the three-phase LC input filter, and then outputs a continuous and smooth grid-side current to the AC side interface.

3. The three-phase single-stage isolated bidirectional converter according to claim 1, characterized in that, The three-phase LC input filter includes a filter inductor module and a filter capacitor module; the single-phase capacitance value C of the filter capacitor module satisfies the following constraint formula: In the formula, P N λ represents the rated active power of the converter system; λ represents the ratio of the maximum allowable fundamental reactive power to the rated active power; ω e Let ω represent the fundamental angular frequency of a three-phase AC power grid. e = 2*π*f e f e U is the fundamental frequency of the power grid. rms This represents the effective value of the phase voltage in a three-phase AC power grid. The single-phase inductance value L of the filter inductor module satisfies the following constraint formula: In the formula, U dc The rated DC voltage for operation of the DC-side interface; f s_min The lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module; ΔI max To meet the maximum allowed peak-to-peak value of high-frequency ripple current in accordance with the power quality standards for grid connection.

4. The three-phase single-stage isolated bidirectional converter according to claim 1, characterized in that, The core of the high-frequency transformer is made of nanocrystalline soft magnetic material, and the geometric parameters of the core and the number of winding turns of the high-frequency transformer are optimized and configured through a dynamic loss separation model based on the frequency conversion condition. The dynamic loss separation model defines the total core loss P under variable frequency non-sinusoidal excitation. core Physically decomposed into hysteresis loss P h Eddy current loss P e and residual loss P r ; The core geometry parameters and winding turns of the high-frequency transformer are determined by optimization using the area AP method combined with the dynamic loss separation model.

5. The three-phase single-stage isolated bidirectional converter according to claim 4, characterized in that, The effective cross-sectional area A of the core of the high-frequency transformer e With window area A w The product of these values ​​is the AP value, which is designed based on the minimum operating frequency set by the system and satisfies the following constraint formula: In the formula, P t K represents the maximum apparent power that the high-frequency transformer needs to transmit. f K represents the waveform coefficients of the non-sinusoidal excitation voltage generated by the matrix converter under high-frequency chopping. u is the window utilization factor of the transformer winding; J is the set rated current density of the conductor; f s_min B is the lowest operating switching frequency within the dynamic adjustment range of the frequency converter control module. m The maximum operating magnetic flux density of the high-frequency transformer is determined by ensuring the total core loss P. core And, under the premise that the temperature rise is within the allowable range of safe heat dissipation, the optimal upper limit of magnetic flux density is obtained iteratively; The equivalent series inductance parameter L of the high-frequency transformer f The leakage inductance is achieved through the equivalent leakage inductance generated by the design of the primary and secondary windings of the high-frequency transformer.

6. The three-phase single-stage isolated bidirectional converter according to claim 2, characterized in that, When the frequency converter control module calculates the phase shift angle and duty cycle, it first collects the actual operating voltage u on the DC side of the full-bridge converter in real time. p Real-time instantaneous phase voltage values ​​on the three-phase AC grid side and the equivalent leakage inductance parameter L of the high-frequency transformer. f The instantaneous values ​​of the three-phase AC phase voltages are sorted by amplitude to reconstruct the maximum envelope voltage vv. max Intermediate envelope voltage vv mid Minimum envelope voltage vv min And the maximum line voltage U is calculated. max =vv max -vv min First intermediate line voltage U mid1 =vv max -vv mid and the second intermediate line voltage U mid2 =vv mid -vv min ; Based on the instantaneous voltage difference across the equivalent leakage inductance of a high-frequency transformer, a piecewise linear inductor current recursive model is constructed within half a high-frequency switching cycle T / 2. The initial node current within the half-cycle is set as i1, and the final node current is set as i4, combined with the DC-side high-frequency square wave voltage u. p The set of discrete node recursive formulas for inductor current is as follows: In the formula, t1 is the duty cycle time, t2 is the phase shift time, T is the high-frequency switching period, and U is the phase shift time. max To measure the maximum line voltage of the converter in AC mode, U mid The first intermediate line voltage of the converter is measured in AC mode, i2 is the second node current, i3 is the third node current, and L is the inductance value of the series inductor. By performing definite integral operations on the instantaneous current function of each segmented interval within half a high-frequency switching cycle, the charge integral IP of each interval is obtained. j1 IP j2 IP j3 The corresponding system of definite integral equations is: In the formula, t represents time.

7. The three-phase single-stage isolated bidirectional converter according to claim 6, characterized in that, The frequency conversion control module compares the first intermediate line voltage U... mid1 With the second intermediate line voltage U mid2 Based on the relationship between the magnitudes, determine the current operating sector of the system, and simultaneously solve for the duty cycle time t1 and phase shift time t2 under the corresponding sector; When determining U mid1 >=U mid2 At that time, the frequency converter control module invokes a preset first set of steady-state equations, which are specifically: In the formula, i max1 i mid1 i min1 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically: When determining U mid1 mid2 At that time, the frequency converter control module calls the preset second steady-state equation set for replacement calculation, the second steady-state equation set being as follows:​ In the formula, i max2 i mid2 i min2 These are the three-phase equivalent average currents derived by integration under the current operating sector of the system, specifically: 。 8. The three-phase single-stage isolated bidirectional converter according to claim 7, characterized in that, The frequency conversion control module has an internal mapping model between the system transmission power and the switching frequency. The mapping model achieves the inverse solution of the target switching frequency through a feedforward analytical step. Based on the charge integral, the frequency conversion control module derives the equivalent average current i on the DC side. d Its derivation formula is: The equivalent average current i on the DC side d It is directly proportional to the high-frequency switching period T and related to the target switching frequency f. s They are inversely proportional; The frequency converter control module acquires the bidirectional operation real-time constant transmission power demand command P issued by the external system. ref And combined with the DC side high-frequency square wave voltage u p Construct a method for directly solving the target switching frequency f. s Analytical power equation: The frequency conversion control module substitutes the calculated duty cycle time t1 and phase shift time t2 into the analytical power equation, and replaces T with 1 / f. s The unique target switching frequency f is determined through a feedforward analytical step. s ; and combine t1, t2 with the target switching frequency f s Generate PWM pulses to drive the three-phase matrix converter and the full-bridge converter. The feedforward analysis step specifically includes: The system obtains the current DC side actual voltage, three-phase AC side actual voltage, and high-frequency transformer leakage inductance parameters, and calculates the current phase shift angle and duty cycle parameters of the modulation strategy based on the system operating conditions. Based on the actual voltage of the three-phase AC side, the actual voltage of the DC side, the leakage inductance parameters, and the current phase shift angle and duty cycle parameters, a frequency-normalized current constant that is completely decoupled from the switching frequency is constructed. Multiply the frequency-normalized current constant by the actual DC-side voltage to obtain the system's frequency-normalized power constant; Using the frequency-normalized power constant as the numerator and the real-time constant transmission power requirement as the denominator, a division operation is performed to obtain the target switching frequency that enables the system to achieve the real-time constant transmission power requirement through feedforward analytical solution.

9. A multi-degree-of-freedom continuous frequency conversion control method, characterized in that, The control method, applied to any one of claims 1 to 8, is executed by a frequency converter control module and includes: Obtain the real-time constant transmission power requirement during the bidirectional operation of the system; Based on the real-time constant transmission power requirement, the state of the switching devices in the three-phase matrix converter and the full-bridge converter is dynamically calculated and coordinated in the order of first determining the phase shift angle and duty cycle, and then analyzing the target switching frequency. The converter is driven by multi-degree-of-freedom continuous frequency conversion modulation, so that the discontinuous current on the input side of the converter is filtered to achieve smooth bidirectional energy transmission.

10. A computer-readable storage medium having a computer program stored thereon, characterized in that, When the computer program is executed by the processor, it implements the multi-degree-of-freedom continuous frequency conversion control method as described in claim 9.

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