Low-cost anti-interference built-in antenna and design method thereof

Abstract

The application relates to the field of wireless communication technology, in particular to a low-cost anti-interference built-in antenna and a design method thereof, which comprises a PCB (printed circuit board), an antenna radiator, a duplexer, a band-pass filter, a phase compensation open-circuit microstrip stub and a matching circuit; the duplexer divides signals at a preset cutoff frequency, the band-pass filter suppresses out-of-band interference, and the phase compensation stub eliminates time delay mismatch of cascaded devices; the design method establishes a mapping relationship between return loss fluctuation frequency and group time delay, and determines the optimal compensation length of the stub through parameterized scanning. The application can strongly suppress signals outside the application frequency band under the premise of high integration, multiple frequency bands and low cost.

Description

Technical Field

[0001] This invention relates to the field of wireless communication technology, and in particular to a low-cost anti-interference built-in antenna and its design method. Background Technology

[0002] With the widespread adoption of wireless communication technology, Wi-Fi devices are now widely used in consumer electronics, smart homes, industrial control, and other fields. Modern Wi-Fi devices generally need to support both 2.4GHz and 5GHz frequency bands simultaneously to meet the varying requirements for transmission speed and coverage in different scenarios. To this end, built-in antennas typically employ a broadband design strategy. By adjusting the size and shape of the antenna radiator and introducing impedance matching networks, the operating bandwidth of the antenna is maximized, enabling a single antenna to cover both operating frequency bands simultaneously. These broadband built-in antennas typically use PCBs as a carrier and conventional substrates such as FR-4 to fabricate the radiator. Their compact structure and low cost have made them the mainstream implementation for built-in antennas in current Wi-Fi devices.

[0003] While broadband antennas extend the operating bandwidth, they also possess strong reception capabilities for out-of-band signals outside the operating frequency band. However, they cannot effectively suppress interference signals between two operating frequency bands or outside the band at the antenna end, allowing a large number of out-of-band signals to enter the RF link of the terminal equipment along with the useful signal. To address this issue, existing technologies typically employ anti-interference measures at the active circuit end of the terminal equipment, including adding metal shielding covers above key components, connecting chip filters in series on power and signal lines, and conducting dedicated EMC wiring planning for sensitive signals. These measures not only significantly increase the design complexity and material costs of the overall circuitry but also require engineers to possess high electromagnetic compatibility design capabilities, extending the product development cycle. More importantly, the aforementioned overall anti-interference designs all rely on externally powered active circuits. Their anti-interference effectiveness is affected by multiple factors such as overall layout, component selection, and assembly processes, making it difficult to establish stable and replicable design specifications. This results in significant uncertainty when products meet EMC testing standards in various countries, increasing the time cost and risk of product certification.

[0004] Chinese Patent Publication No. CN204577588U discloses an antenna and a tablet computer for use in tablet computers. The antenna adopts a dual-radiating stub structure. The first radiating stub is placed on a supporting medium by bending and folding, making full use of the internal space of the phone to increase the effective radiating area and improve the antenna radiation efficiency. One section of the first radiating stub is arranged above the supporting medium, leaving the metal ground below clear, which serves both to fix the antenna and to reduce the number of radiating stubs. The antenna design utilizes single-stub dual-mode technology. The first radiating stub generates two resonant modes in the GSM 850MHz band and the UMTS 1900MHz band, while the second radiating stub generates one resonant mode at 2400MHz. The frequency is then adjusted by using a matching circuit. However, the antenna and tablet computer described above have the following problems: multi-band coverage is achieved only through radiating stubs and matching circuits, without effective suppression of out-of-band signals outside the operating frequency band, requiring additional anti-interference design at the circuit level for the terminal device; furthermore, the antenna cannot achieve physical isolation of multi-band signals, posing a risk of inter-band crosstalk. Summary of the Invention

[0005] Therefore, this invention provides a low-cost anti-interference built-in antenna and its design method to overcome the problems in the prior art, such as the inability of broadband built-in antennas to suppress out-of-band interference signals at the antenna end and the high design cost and unstable effect caused by the reliance on the active circuit of the whole machine for anti-interference design.

[0006] To achieve the above objectives, the present invention provides a low-cost anti-interference built-in antenna and its design method, comprising: PCB circuit board; An antenna radiator is disposed on the front side of the PCB circuit board. The radiation size of the antenna radiator is determined based on a quarter wavelength and covers the first operating frequency band and the second operating frequency band. A duplexer is connected to the signal output terminal of the antenna radiator. The duplexer uses a preset cutoff frequency between the first operating frequency band and the second operating frequency band as the branching boundary to divide the signal output by the antenna radiator into a low-frequency link and a high-frequency link. A bandpass filter is connected in series with the low-frequency link. The passband of the bandpass filter covers the first operating frequency band and suppresses out-of-band signals outside the first operating frequency band. A phase-compensated open-circuit microstrip stub is connected to the microstrip transmission line between the low-frequency output of the duplexer and the input of the bandpass filter, so that the oscillation frequency of the return loss curve of the antenna radiator port is less than a preset oscillation frequency threshold in the transition band between the first operating frequency band and the second operating frequency band after the duplexer and the bandpass filter are cascaded. A matching circuit, connected to the end of the low-frequency link, is used to tune the impedance of the low-frequency link so that the standing wave ratio (SWR) in both the first and second operating frequency bands is less than a preset SWR threshold. The antenna ground layer is located on the back of the PCB circuit board and is connected to the ground terminals of the duplexer, the bandpass filter, the phase-compensated open-circuit microstrip stub, and the matching circuit, respectively. The inner conductor pad and the outer conductor pad of the feeder are both disposed on the PCB circuit board. The inner conductor pad of the feeder is connected to the signal output terminal of the matching circuit, and the outer conductor pad of the feeder is connected to the antenna ground layer. A coaxial cable, wherein the inner conductor is soldered to the inner conductor pad of the feed line, and the outer conductor is soldered to the outer conductor pad of the feed line. A connector, which is connected to the end of the coaxial cable away from the PCB circuit board, is used to interface with the radio frequency interface of the terminal device.

[0007] Furthermore, the matching circuit includes a matching capacitor connected in parallel between the signal line of the low-frequency link and the antenna ground layer.

[0008] This invention also provides a low-cost, anti-interference built-in antenna design method, comprising: Step S1: Establish a parameterized simulation model of the low-cost anti-interference built-in antenna in the electromagnetic simulation software, and set the electrical length of the phase-compensated open-circuit microstrip stub as the variable to be determined, with the initial value set to zero. Step S2: Based on the parameterized simulation model, extract the extreme point frequency coordinates of the return loss curve of the antenna radiator port in the transition band between the first operating frequency band and the second operating frequency band, and determine the fluctuation frequency of the return loss curve in the transition band. Step S3: Determine the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude; and determine the fluctuation frequency threshold based on the mapping relationship and the preset group delay fluctuation amplitude threshold. Step S4: Based on the determination result of the fluctuation frequency and the fluctuation frequency threshold, determine the optimal compensation length of the phase compensation open-circuit microstrip stub. Step S5: Substitute the optimal compensation length into the parameterized simulation model for verification. After successful verification, convert the optimal compensation length into the physical dimensions of the phase-compensated open-circuit microstrip stub and output it as the plate-making parameter.

[0009] Further, step S2 includes: Step S21: In the parameterized simulation model, the nominal value of the dielectric constant of the PCB circuit board is used for simulation to obtain the return loss curve of the antenna radiator port in the full frequency band. Step S22: On the return loss curve, locate the transition band region between the cutoff frequency of the first operating frequency band and the start frequency of the second operating frequency band, and extract the frequency coordinates of all local maxima and local minima of the return loss curve within the transition band region. Step S23: Based on the frequency coordinates of all the local maxima and minima, calculate the frequency interval between adjacent extreme points in order of frequency from low to high, and take the average value of all the frequency intervals to obtain the average frequency interval. Step S24: Determine the fluctuation frequency of the return loss curve within the transition band based on the reciprocal of the average frequency interval.

[0010] Furthermore, in step S3, the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude is determined in the following way: In the parametric simulation model, the component parameters of the duplexer and the component parameters of the bandpass filter are each taken into several different sets of values ​​within their respective manufacturing tolerance ranges. Simulation is performed on each set of values ​​to obtain the corresponding fluctuation frequency and the corresponding group delay fluctuation amplitude, forming several sets of corresponding data points of fluctuation frequency and group delay fluctuation amplitude. Based on all the corresponding data points, curve fitting is performed to obtain the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude.

[0011] Further, step S4 includes: Step S41: Compare the fluctuation frequency obtained in step S2 with the fluctuation frequency threshold obtained in step S3; Step S42: Based on the comparison result that the fluctuation frequency is less than or equal to the fluctuation frequency threshold, the optimal compensation length of the phase compensation open-circuit microstrip stub is determined to be zero, and step S4 ends; based on the comparison result that the fluctuation frequency is greater than the fluctuation frequency threshold, step S43 is executed. Step S43: Perform parametric scanning on the phase-compensated open-circuit microstrip stub, substitute each scan value into the parametric simulation model for simulation, and obtain the corresponding fluctuation frequency; based on the relationship between the fluctuation frequency and the electrical length, determine the electrical length corresponding to the minimum value that makes the fluctuation frequency less than the fluctuation frequency threshold as the optimal compensation length of the phase-compensated open-circuit microstrip stub.

[0012] Further, step S43 includes: Step S431: The dielectric constant of the PCB circuit board and the electrical length of the phase compensation open-circuit microstrip stub are simultaneously set as scanning variables. The dielectric constant takes several discrete values ​​within the batch tolerance range of the PCB circuit board substrate, and the electrical length takes several discrete values ​​within a preset range with the optimal compensation length as the center. Simulation is performed on all combinations of the dielectric constant values ​​and the electrical length values ​​to obtain the fluctuation frequency corresponding to each combination. Step S432: Based on the fluctuation frequencies of all combinations obtained in step S431, determine the range of electrical length values ​​in which the fluctuation frequencies are all less than the fluctuation frequency threshold under all values ​​of the dielectric constant, and denot it as the compensation length tolerance range. Step S433: Update the optimal compensation length based on the positional relationship between the optimal compensation length and the compensation length tolerance interval.

[0013] Furthermore, in step S433, if the optimal compensation length is located in the middle of the compensation length tolerance interval, the optimal compensation length remains unchanged; if the optimal compensation length is biased towards the edge of the compensation length tolerance interval, the median of the compensation length tolerance interval is updated to the optimal compensation length.

[0014] Furthermore, in step S431, the upper limit of the scanning length of the phase-compensated open-circuit microstrip stub is the effective quarter wavelength corresponding to the center frequency of the transition band in the PCB substrate.

[0015] Further, the invention is characterized in that, in step S3, the mapping relationship is a polynomial fitting relationship between the fluctuation frequency and the group delay fluctuation amplitude, and the coefficients of each term of the polynomial fitting relationship are determined by fitting calculation based on all corresponding data points. On the other hand, the present invention also provides a processing system, comprising: Compared with the prior art, the beneficial effect of the present invention is that by integrating a duplexer, a bandpass filter and a phase-compensated open-circuit microstrip stub at the antenna end, the present invention simultaneously achieves multi-band signal separation and out-of-band suppression under passive conditions, so that the terminal equipment can meet the electromagnetic compatibility testing requirements without the need to add anti-interference design.

[0016] Furthermore, this invention uses a duplexer to divide the signal into a low-frequency link and a high-frequency link with a preset cutoff frequency as the branch boundary, thereby achieving physical isolation between the two operating frequency band signals and eliminating crosstalk between frequency bands from the source.

[0017] Furthermore, the present invention filters out out-of-band signals outside the operating frequency band directly at the antenna end by connecting a bandpass filter in series on the low-frequency link, so that the signal received by the RF link of the terminal device contains only the effective operating frequency band components.

[0018] Furthermore, by introducing a phase-compensated open-circuit microstrip stub between the low-frequency output of the duplexer and the input of the bandpass filter, this invention eliminates the group delay mismatch generated when the duplexer and the bandpass filter are cascaded, ensuring the stability of the overall filtering performance after cascading and that the VSWR in the operating frequency band meets the design requirements.

[0019] Furthermore, this invention establishes a mapping relationship between the fluctuation frequency of the transition band return loss curve and the fluctuation amplitude of the group delay through parametric simulation, transforming the group delay mismatch, which is difficult to measure directly, into a quantitative index that can be directly extracted from the simulation curve. This enables the determination of the optimal physical size of the phase-compensated open-circuit microstrip stub without the need for physical testing of finished products.

[0020] Furthermore, by introducing a two-dimensional parametric scan of the batch tolerance of dielectric constant, this invention determines the compensation length tolerance range and takes the midpoint of the range as the board-making parameter, so that the antenna design results can maintain performance indicators within the batch fluctuation range of PCB substrate, thus ensuring the consistency of mass production. Attached Figure Description

[0021] Figure 1 This is a front view of the low-cost anti-interference built-in antenna according to an embodiment of the present invention; Figure 2 This is a rear view of the low-cost anti-interference built-in antenna according to an embodiment of the present invention; Figure 3 This is a flowchart illustrating the design method of the low-cost anti-interference built-in antenna of the present invention; Figure 4 This is a flowchart of step S2 of the design method for a low-cost anti-interference built-in antenna according to the present invention; In the diagram, PCB board-1; WIFI antenna line-2; WIFI antenna to splitter / combiner connection line-3; splitter / combiner-4; 2.4G filter-5; 0.75pF capacitor-6; splitter / microstrip line-7; antenna ground-8; inner conductor pad of feeder-9; outer conductor pad of feeder-10; RF1.37 cable-11; insulating sleeve-12; MMCX-J connector-13; back of PCB board-14; antenna fixing adhesive-15; adhesive tear-off opening-16. Detailed Implementation

[0022] To make the objectives and advantages of the present invention clearer, the present invention will be further described below with reference to embodiments; it should be understood that the specific embodiments described herein are merely for explaining the present invention and are not intended to limit the present invention.

[0023] Preferred embodiments of the present invention will now be described with reference to the accompanying drawings. Those skilled in the art should understand that these embodiments are merely illustrative of the technical principles of the present invention and are not intended to limit the scope of protection of the present invention.

[0024] It should be noted that in the description of this invention, the terms "upper", "lower", "left", "right", "inner", "outer", etc., which indicate directions or positional relationships, are based on the directions or positional relationships shown in the accompanying drawings. This is only for the convenience of description and is not intended to indicate or imply that the device or element must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, it should not be construed as a limitation of this invention.

[0025] Furthermore, it should be noted that, in the description of this invention, unless otherwise explicitly specified and limited, the terms "installation," "connection," and "linking" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art can understand the specific meaning of the above terms in this invention according to the specific circumstances.

[0026] Please see Figure 1 and Figure 2 The figures shown are structural diagrams of a low-cost anti-interference built-in antenna according to an embodiment of the present invention and a rear structural diagram of a low-cost anti-interference built-in antenna according to an embodiment of the present invention. The present invention provides a low-cost anti-interference built-in antenna, comprising: PCB circuit board; An antenna radiator is disposed on the front side of the PCB circuit board. The radiation size of the antenna radiator is determined based on a quarter wavelength and covers the first operating frequency band and the second operating frequency band. A duplexer is connected to the signal output terminal of the antenna radiator. The duplexer uses a preset cutoff frequency between the first operating frequency band and the second operating frequency band as the branching boundary to divide the signal output by the antenna radiator into a low-frequency link and a high-frequency link. A bandpass filter is connected in series with the low-frequency link. The passband of the bandpass filter covers the first operating frequency band and suppresses out-of-band signals outside the first operating frequency band. A phase-compensated open-circuit microstrip stub is connected to the microstrip transmission line between the low-frequency output of the duplexer and the input of the bandpass filter, so that the oscillation frequency of the return loss curve of the antenna radiator port is less than a preset oscillation frequency threshold in the transition band between the first operating frequency band and the second operating frequency band after the duplexer and the bandpass filter are cascaded. A matching circuit, connected to the end of the low-frequency link, is used to tune the impedance of the low-frequency link so that the standing wave ratio (SWR) in both the first and second operating frequency bands is less than a preset SWR threshold. The antenna ground layer is located on the back of the PCB circuit board and is connected to the ground terminals of the duplexer, the bandpass filter, the phase-compensated open-circuit microstrip stub, and the matching circuit, respectively. The inner conductor pad and the outer conductor pad of the feeder are both disposed on the PCB circuit board. The inner conductor pad of the feeder is connected to the signal output terminal of the matching circuit, and the outer conductor pad of the feeder is connected to the antenna ground layer. A coaxial cable, wherein the inner conductor is soldered to the inner conductor pad of the feed line, and the outer conductor is soldered to the outer conductor pad of the feed line. A connector, which is connected to the end of the coaxial cable away from the PCB circuit board, is used to interface with the radio frequency interface of the terminal device.

[0027] Specifically, the matching circuit includes a matching capacitor connected in parallel between the signal line of the low-frequency link and the antenna ground layer.

[0028] In a specific embodiment, the specific implementation of each structural component of the present invention is described below with reference to the accompanying drawings. The relationships between the reference numerals in the drawings and the terms in the claims are as follows: WIFI antenna line 2 corresponds to the antenna radiator in the claims; WIFI antenna and splitter / combiner connection line 3 corresponds to the connection trace between the signal output terminal of the antenna radiator and the input terminal of the duplexer in the claims; splitter / combiner 4 corresponds to the duplexer in the claims; 2.4G filter 5 corresponds to the bandpass filter in the claims; 0.75pF capacitor 6 corresponds to the matching capacitor in the matching circuit in the claims; splitter / microstrip line 7 corresponds to the microstrip transmission line between the output terminal of the duplexer and the input terminals of each link device in the claims; antenna ground 8 corresponds to the antenna ground layer in the claims; RF1.37 cable 11 corresponds to the coaxial cable in the claims; MMCX-J connector 13 corresponds to the connector in the claims.

[0029] The PCB circuit board 1 is made of FR-4 material with a nominal dielectric constant of 4.4 and a preferred board thickness of 0.8mm. Its front side is used to support the antenna radiator and all passive components, while the back side is used to arrange the antenna ground 8 and the antenna fixing adhesive 15.

[0030] The WIFI antenna line 2 is fabricated on the front side of the PCB board 1 using copper foil traces. Its radiation size is determined according to the quarter-wavelength formula, with the first operating frequency band being 2400MHz to 2480MHz and the second operating frequency band being 5150MHz to 5850MHz. The WIFI antenna to splitter / combiner connection line 3 is also fabricated on the front side of the PCB board 1 using copper foil traces, connecting the signal output terminal of the WIFI antenna line 2 to the input terminal of the splitter / combiner 4.

[0031] The 1-to-2 combiner 4 employs a lumped-parameter LC-type passive duplexer, surface-mounted on the front of the PCB board 1. The splitting boundary is set at 3.3GHz, with the low-frequency channel having a passband of 0 to 3.3GHz and the high-frequency channel having a passband of 3.3GHz to 6GHz. It splits the full-band signal from the WIFI antenna line 2 into two independent transmission paths: a low-frequency link and a high-frequency link. The low-frequency and high-frequency outputs of the 1-to-2 combiner 4 are connected to subsequent devices via a 1-to-2 microstrip line 7.

[0032] The phase-compensated open-circuit microstrip stub is fabricated on the front side of PCB circuit board 1 in the form of copper foil traces. One end is connected to the low-frequency link segment of the 1-to-2 microstrip line 7, located between the low-frequency output terminal of the 1-to-2 combiner 4 and the input terminal of the 2.4G filter 5. The other end is left open. Its electrical length is given by the optimal compensation length determined by the design method of this invention and is directly reflected in the Gerber file of PCB fabrication.

[0033] The 2.4G filter 5 is a lumped parameter LC type passive bandpass filter, which is surface-mounted on the front of the PCB circuit board 1 and connected in series to the low frequency link. The passband is 2400MHz to 2480MHz, and the suppression of out-of-band signals outside the passband is greater than or equal to 20dB.

[0034] A 0.75pF capacitor 6 is surface-mounted to the front of the PCB circuit board 1, forming the core component of the matching circuit described in the claims of this invention. It is connected in parallel between the low-frequency link signal line and the antenna ground 8 to perform final tuning of the impedance at the end of the low-frequency link, so that the VSWR in both the first and second operating frequency bands is less than 2.0.

[0035] Antenna ground 8 is covered on the back of the PCB board 14 in the form of copper foil, and is connected to the ground terminals of the 1-to-2 combiner 4, 2.4G filter 5, phase compensation open microstrip stub and 0.75pF capacitor 6 through vias to provide a unified RF reference ground.

[0036] The inner conductor pad 9 and outer conductor pad 10 of the feeder are located at the ends of the PCB board 1. The inner conductor pad 9 is connected to the signal output terminal of the 0.75pF capacitor 6, and the outer conductor pad 10 is connected to the antenna ground 8. The inner conductor of the RF1.37 cable 11 is soldered to the inner conductor pad 9, and the outer conductor is soldered to the outer conductor pad 10. An insulating sleeve 12 is fitted between the RF1.37 cable 11 and the PCB board 1 for insulation protection. The MMCX-J connector 13 is fixed to the end of the RF1.37 cable 11 away from the PCB board 1 for interfacing with the RF interface of the terminal equipment.

[0037] The antenna fixing adhesive 15 is pasted on the back of the PCB board 14 to attach and fix the PCB circuit board 1 to the preset installation position on the inner wall of the terminal device. The adhesive tear-off opening 16 is set on the edge of the antenna fixing adhesive 15 to facilitate peeling off the adhesive during installation.

[0038] Understandably, the overall working principle of the aforementioned antenna structure is as follows: After receiving the wireless signal in space via WIFI antenna line 2, a broadband signal is formed containing a first operating frequency band, a second operating frequency band, and a large number of out-of-band components. This signal is transmitted via WIFI antenna and splitter / combiner connection line 3 to splitter / combiner 4. Splitter / combiner 4 uses 3.3GHz as the splitting boundary, rigidly dividing the signal into two independent paths: a low-frequency link and a high-frequency link. This ensures that the first operating frequency band signal enters the low-frequency link, and the second operating frequency band signal enters the high-frequency link, thus physically isolating the two frequency bands and preventing interference. The signal entering the low-frequency link first propagates along splitter / microstrip line 7, passing through a phase-compensated open-circuit microstrip stub. This stub does not directly filter the signal but rather compensates for the signal phase and simultaneously improves the VSWR within the operating frequency band. After passing through the phase-compensated open-circuit microstrip stub, the signal enters the 2.4G filter 5. The 2.4G filter 5 suppresses all out-of-band signals outside the 2400MHz to 2480MHz range, allowing only signals in the first operating frequency band to pass. Subsequently, the signal undergoes final impedance tuning via a 0.75pF capacitor 6 and is output to the terminal equipment through the inner conductor pad 9 of the feeder, the RF1.37 cable 11, and the MMCX-J connector 13. All components in the entire solution are passive, requiring no external power supply. Multi-band fusion and out-of-band suppression are completed at the antenna end, eliminating the need for any additional anti-interference design in the terminal equipment.

[0039] In one specific embodiment, the 0.75pF capacitor 6 in the matching circuit is surface-mounted to the front of the PCB circuit board 1 and connected in parallel between the signal line of the low-frequency link and the antenna ground 8. The capacitance value is preferably 0.75pF. The 0.75pF capacitor 6 compensates for the residual inductive impedance component presented at the output of the 2.4G filter 5 by adjusting the equivalent parallel admittance at the end of the low-frequency link, bringing the output impedance of the first and second operating frequency bands closer to the 50Ω standard impedance, so that the VSWR in both operating frequency bands is less than 2.0.

[0040] Understandably, this invention constructs a complete signal processing closed loop by highly integrating multiple levels of passive components at the antenna end, fundamentally changing the implementation path of traditional anti-interference design. After the antenna radiator receives broadband signals in space, the duplexer physically separates the signal into two independent links—low-frequency and high-frequency—with a preset cutoff frequency as the boundary, avoiding inter-band crosstalk from the source. The bandpass filter connected in series on the low-frequency link directly filters out interference signals outside the operating frequency band, so that the out-of-band suppression function is completed at the antenna end. The phase-compensated open-circuit microstrip stub eliminates the group delay mismatch between cascaded devices through precise phase tuning, ensuring the stability of filtering performance. The end matching circuit performs final impedance calibration to ensure efficient signal transmission. This design, which integrates the entire process of frequency band separation, out-of-band suppression, phase compensation, and impedance matching into the antenna module, allows the terminal device to receive the processed clean signal only through a connector, completely eliminating the need for traditional anti-interference measures such as adding shielding covers, chip filters, and EMC-specific wiring at the device circuit end. Since all functions are implemented using passive components, no external power supply circuit is required. This reduces the complexity of the antenna itself and transforms the overall anti-interference design cost into the benefits of large-scale production of the antenna module. Ultimately, the antenna simultaneously achieves efficient transmission of multi-band signals and deep suppression of out-of-band interference under passive conditions, forming a "plug-and-play" anti-interference solution for terminal devices.

[0041] Please see Figure 3 The diagram shows a flowchart of the design method for a low-cost anti-interference built-in antenna according to the present invention; the present invention also provides a design method for a low-cost anti-interference built-in antenna, comprising: Step S1: Establish a parameterized simulation model of the low-cost anti-interference built-in antenna in the electromagnetic simulation software, and set the electrical length of the phase-compensated open-circuit microstrip stub as the variable to be determined, with the initial value set to zero. In one specific embodiment, step S1 is implemented in the HFSS electromagnetic simulation software, and specifically includes the following operations: Establish the basic model: Import the 3D CAD model of the antenna structure described in claim 1, including all structural components such as PCB circuit board 1, WIFI antenna line 2, WIFI antenna and splitter connection line 3, splitter 4, 2.4G filter 5, 0.75pF capacitor 6, splitter microstrip line 7, phase compensation open-circuit microstrip stub 17, etc. Set material parameters: The PCB substrate material is set to FR-4, and its dielectric constant εr is set as a variable, with a value range of 4.2 to 4.6 and a nominal value of 4.4. The conductivity of the copper foil conductor is set to 5.8 × 10⁻⁶. 7 S / m; Device parameterization: The lumped LC element value of the 1-to-2 combiner 4 is set as a variable, allowing a tolerance fluctuation of ±5%; The lumped LC element values ​​of the 2.4G filter 5 are set as variables, allowing a tolerance fluctuation of ±5%; Key variable settings: The electrical length Lstub of phase-compensated open-circuit microstrip stub 17 is set as a variable to be determined, and the initial value is set to zero (i.e., physical length 0mm). The capacitance of capacitor 6 is fixed at 0.75pF. Boundary conditions: Antenna ground 8 is set as an ideal electrical conductor boundary; The size of the radiation bounding box is set to 1 / 4 of the operating wavelength λmax (λmax = 6GHz corresponding wavelength). Port settings: The excitation port is located at interface 13 of the MMCX-J connector; The reference impedance is set to 50Ω.

[0042] Step S2: Based on the parameterized simulation model, extract the extreme point frequency coordinates of the return loss curve of the antenna radiator port in the transition band between the first operating frequency band and the second operating frequency band, and determine the fluctuation frequency of the return loss curve in the transition band. Please continue reading. Figure 4 The diagram shows a flowchart of step S2 in the design method of the low-cost anti-interference built-in antenna of the present invention; specifically, step S2 includes: Step S21: In the parameterized simulation model, the nominal value of the dielectric constant of the PCB circuit board is used for simulation to obtain the return loss curve of the antenna radiator port in the full frequency band. Step S22: On the return loss curve, locate the transition band region between the cutoff frequency of the first operating frequency band and the start frequency of the second operating frequency band, and extract the frequency coordinates of all local maxima and local minima of the return loss curve within the transition band region. Step S23: Based on the frequency coordinates of all the local maxima and minima, calculate the frequency interval between adjacent extreme points in order of frequency from low to high, and take the average value of all the frequency intervals to obtain the average frequency interval. Step S24: Determine the fluctuation frequency of the return loss curve within the transition band based on the reciprocal of the average frequency interval.

[0043] In one specific embodiment, step S2 is implemented in the post-processing module of the HFSS electromagnetic simulation software, and specifically includes the following operations: In the parametric simulation model, the dielectric constant of the PCB circuit board is fixed at the nominal value of 4.4, and the component parameters of the duplexer and bandpass filter are all taken at their nominal values. The electrical length of the phase-compensated open-circuit microstrip stub remains at its initial value of zero. The frequency scan range is set to 2GHz to 6GHz, the scan type is discrete fast scan, and the step size is set to 1MHz. After the simulation is executed, the return loss S11 curve data of the antenna radiator port is exported, which is a frequency-amplitude value pair sequence.

[0044] On the exported S11 curve data, the cutoff frequency f1 = 2480MHz of the first working frequency band and the starting frequency f2 = 5150MHz of the second working frequency band are located, and the frequency interval [f1, f2] is defined as the transition band region. Within this region, the amplitude value sequence of the S11 curve is traversed to identify all local maxima and local minima. Local maxima satisfy the condition that their amplitude value is greater than the amplitude values ​​of the preceding and following points; local minima satisfy the condition that their amplitude value is less than the amplitude values ​​of the preceding and following points. The frequency coordinates corresponding to all extreme points are recorded to form the extreme point frequency sequence {F}. i}, i=1,2,...,N, where N is the total number of extreme points.

[0045] The extreme point frequency sequence {F i Sort by frequency value from lowest to highest. Calculate the frequency interval Δf between two adjacent extreme points. j =F{j+1}-Fj, j=1,2,...,N-1. The arithmetic mean of all N-1 frequency intervals is taken to obtain the average frequency interval Δf. avg .

[0046] Calculate the average frequency interval Δf avg The reciprocal of the product of the two products yields the oscillation frequency F of the return loss curve within the transition band. ripple That is, Fripple =1 / Δf avg F ripple The unit is (cycles / GHz).

[0047] Understandably, the reason for calculating the average frequency interval in step S23 instead of directly using the minimum interval is that the fluctuations of the S11 curve within the transition band may be uneven, and the average interval can more stably characterize the overall fluctuation rate; step S24 takes the reciprocal to obtain F. ripple This is because the definition of fluctuation frequency is the number of fluctuation cycles per unit frequency, which is inversely proportional to the average interval.

[0048] Understandably, when a duplexer is cascaded with a bandpass filter, the signal undergoes multiple reflections at the interface due to impedance mismatch. These reflected waves superimpose with the incident wave to form a standing wave, which manifests as amplitude oscillations in the S11 curve within the transition band in the frequency domain. The intensity of these oscillations directly reflects the severity of the group delay mismatch. The denser the oscillations (i.e., the smaller the frequency interval between adjacent extreme points), the faster the phase relationship between the reflected and incident waves changes with frequency, and the greater the group delay fluctuation. By extracting all the extreme points of the S11 curve within the transition band and calculating the reciprocal of their average interval, an index F can be obtained to quantify the intensity of the oscillations. ripple This indicator is strongly correlated with the group delay fluctuation amplitude and can be used as a target control variable for subsequent compensation optimization. This characterization method based on the extreme point analysis of the S11 curve avoids the high-precision phase detection required for direct measurement of group delay, significantly reducing the optimization complexity.

[0049] Step S3: Determine the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude; and determine the fluctuation frequency threshold based on the mapping relationship and the preset group delay fluctuation amplitude threshold. Specifically, in step S3, the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude is determined in the following way: In the parametric simulation model, the component parameters of the duplexer and the component parameters of the bandpass filter are each taken into several different sets of values ​​within their respective manufacturing tolerance ranges. Simulation is performed on each set of values ​​to obtain the corresponding fluctuation frequency and the corresponding group delay fluctuation amplitude, forming several sets of corresponding data points of fluctuation frequency and group delay fluctuation amplitude. Based on all the corresponding data points, curve fitting is performed to obtain the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude.

[0050] Specifically, in step S3, the mapping relationship is a polynomial fitting relationship between the fluctuation frequency and the group delay fluctuation amplitude, and the coefficients of each term of the polynomial fitting relationship are determined by fitting calculation based on all corresponding data points.

[0051] In one specific embodiment, step S3 is implemented in the MATLAB data processing platform, and specifically includes the following operations: Five sets of different values ​​(nominal value × 0.95, 0.97, 1.0, 1.03, 1.05) are uniformly taken for the parameters of the duplexer components within a tolerance range of ±5% of the nominal value; five sets of different values ​​are also uniformly taken for the parameters of the bandpass filter components within a tolerance range of ±5% of the nominal value; the five sets of values ​​for the duplexer are combined with the five sets of values ​​for the filter to form 25 independent parameter combinations. For each parameter combination, perform a full-band simulation (2-6 GHz) in HFSS; calculate the corresponding fluctuation frequency F for each combination according to the method in step S2. ripple Extract the group delay curve in the post-simulation processing, calculate the difference between the maximum and minimum group delay values ​​in the first operating frequency band (2400-2480MHz) and the second operating frequency band (5150-5850MHz), and take the larger of the two differences as the group delay fluctuation amplitude Δτ of this set of parameters; 25 groups (F ripple Import the data (Δτ) into MATLAB; call the polyfit function to perform a quadratic polynomial fitting: Δτ=a×F ripple 2 +b×F ripple +c; Where, a is the coefficient of the quadratic term, dimensionless, with a value range of -0.5 to 0.5, determined by fitting calculation; b is the coefficient of the linear term, dimensionless, with a value range of -2.0 to 2.0, determined by fitting calculation; c is the constant term, in nanoseconds (ns), with a value range of -1.0 to 1.0, determined by fitting calculation; Based on the tolerance requirements for inter-symbol interference in the 802.11ac standard, the group delay fluctuation amplitude threshold Δτmax is set to 0.5 ns. Substituting Δτmax into the fitting polynomial, we solve the equation: 0.5 = a × F threshold 2 +b×F threshold +c; The minimum real root value that is greater than zero is taken as the fluctuation frequency threshold F. threshold ; Understandably, the choice of the quadratic polynomial fitting form is based on the statistical characteristics of the electromagnetic simulation data: when F ripple When it is small, Δτ varies with F ripple Approximately linear growth; when F ripple When the value is large, the growth of Δτ accelerates, exhibiting quadratic characteristics. The signs and magnitudes of the coefficients a, b, and c reflect the degree of nonlinearity of the impedance mismatch between the duplexer and the filter in the specific antenna design. Negative values ​​indicate that the group delay fluctuation caused by the mismatch exhibits anomalous saturation characteristics.

[0052] Understandably, when a duplexer and filter are cascaded, the interface impedance mismatch leads to multiple signal reflections. The phase interference between these reflected waves and the incident wave manifests as amplitude oscillations of the S11 curve in the frequency domain and transforms into group delay ripples in the time domain. By systematically changing the device parameters and collecting data points, we can reveal the F... ripple The inherent statistical regularity between Δτ and the group delay index. This mapping relationship based on statistical fitting transforms the group delay index, which is difficult to measure directly, into a fluctuation frequency index that is easy to extract from the S11 curve, providing a quantifiable control target for subsequent compensation optimization. The preset group delay threshold originates from the fundamental limitation of communication protocol on signal distortion. After converting it into a fluctuation frequency threshold, accurate phase compensation can be implemented under purely passive conditions.

[0053] Step S4: Based on the determination result of the fluctuation frequency and the fluctuation frequency threshold, determine the optimal compensation length of the phase compensation open-circuit microstrip stub. Specifically, step S4 includes: Step S41: Compare the fluctuation frequency obtained in step S2 with the fluctuation frequency threshold obtained in step S3; Step S42: Based on the comparison result that the fluctuation frequency is less than or equal to the fluctuation frequency threshold, the optimal compensation length of the phase compensation open-circuit microstrip stub is determined to be zero, and step S4 ends; based on the comparison result that the fluctuation frequency is greater than the fluctuation frequency threshold, step S43 is executed. Step S43: Perform parametric scanning on the phase-compensated open-circuit microstrip stub, substitute each scan value into the parametric simulation model for simulation, and obtain the corresponding fluctuation frequency; based on the relationship between the fluctuation frequency and the electrical length, determine the electrical length corresponding to the minimum value that makes the fluctuation frequency less than the fluctuation frequency threshold as the optimal compensation length of the phase-compensated open-circuit microstrip stub.

[0054] Specifically, step S43 includes: Step S431: The dielectric constant of the PCB circuit board and the electrical length of the phase compensation open-circuit microstrip stub are simultaneously set as scanning variables. The dielectric constant takes several discrete values ​​within the batch tolerance range of the PCB circuit board substrate, and the electrical length takes several discrete values ​​within a preset range with the optimal compensation length as the center. Simulation is performed on all combinations of the dielectric constant values ​​and the electrical length values ​​to obtain the fluctuation frequency corresponding to each combination. Specifically, in step S431, the upper limit of the scanning length of the phase-compensated open-circuit microstrip stub is the effective quarter wavelength of the transition band center frequency in the PCB substrate.

[0055] Step S432: Based on the fluctuation frequencies of all combinations obtained in step S431, determine the range of electrical length values ​​in which the fluctuation frequencies are all less than the fluctuation frequency threshold under all values ​​of the dielectric constant, and denot it as the compensation length tolerance range. Step S433: Update the optimal compensation length based on the positional relationship between the optimal compensation length and the compensation length tolerance interval.

[0056] Specifically, in step S433, if the optimal compensation length is located in the middle of the compensation length tolerance interval, the optimal compensation length remains unchanged; if the optimal compensation length is biased towards the edge of the compensation length tolerance interval, the median of the compensation length tolerance interval is updated to the optimal compensation length.

[0057] In a specific embodiment, step S41 directly compares the fluctuation frequency obtained in step S2 with the fluctuation frequency threshold obtained in step S3. If the fluctuation frequency is less than or equal to the fluctuation frequency threshold, it indicates that the cascaded group delay mismatch between the duplexer and the bandpass filter is within an acceptable range under uncompensated conditions, and the optimal compensation length of the phase-compensated open-circuit microstrip stub is determined to be 0, meaning that there is no need to fabricate a physical trace for this stub on the PCB, and step S4 ends. If the fluctuation frequency is greater than the fluctuation frequency threshold, then step S43 is executed.

[0058] In a specific embodiment, in step S43, the electrical length of the phase-compensated open-circuit microstrip stub is used as the scanning variable, and a parameterized scan is performed in HFSS. The scanning start point is 0, the scanning end point is the upper limit of the scan Lmax, and the scanning step size is preferably Lmax divided by 20, that is, the range from zero to Lmax is divided into twenty scanning steps.

[0059] Wherein, Lmax is the upper limit of the scan, in millimeters (mm). It is the physical length of the effective quarter-wavelength corresponding to the center frequency of the transition band in the PCB substrate 1, determined by the center frequency of the transition band, the nominal relative permittivity of the FR-4 substrate, and the speed of light in free space. Specifically, the physical length of the effective quarter-wavelength is equal to the square root of the free space quarter-wavelength divided by the nominal relative permittivity of the FR-4 substrate. Preferably, under the conditions of a transition band center frequency of approximately 3.8 GHz and a nominal FR-4 permittivity of 4.4, the physical length of Lmax is approximately 10 mm, calculated and calibrated based on the effective wavelength of the transition band center frequency in FR-4.

[0060] For each electrical length scan value, the parameterized simulation model is substituted into the simulation. The fluctuation frequency corresponding to the scan value is calculated according to the method in step S2, resulting in a discrete data sequence of fluctuation frequency as a function of electrical length. In this discrete data sequence, the discrete electrical length value corresponding to the first fluctuation frequency value being less than the fluctuation frequency threshold is found. This value is recorded as the initial optimal compensation length Lopt, and then step S431 is executed.

[0061] In a specific embodiment, in step S431, the dielectric constant of the PCB circuit board 1 and the electrical length of the phase-compensated open-circuit microstrip stub are simultaneously set as scanning variables for two-dimensional parametric scanning. The dielectric constant is uniformly selected from 5 discrete values ​​within the batch tolerance range of 4.2 to 4.6 for the FR-4 substrate, with an interval of 0.1 between adjacent discrete values. The electrical length is uniformly selected from 9 discrete values ​​within a range extending one-quarter of Lmax to both sides of Lopt as the center, with an interval of Lmax divided by 16 between adjacent discrete values. All 45 combinations of dielectric constant and electrical length are substituted into the parametric simulation model for simulation. The fluctuation frequency corresponding to each combination is calculated according to the method in step S2, forming a five-row, nine-column two-dimensional fluctuation frequency numerical matrix. The rows of the matrix correspond to the 5 values ​​of the dielectric constant, and the columns correspond to the 9 values ​​of the electrical length.

[0062] In a specific embodiment, in step S432, the above two-dimensional fluctuation frequency matrix is ​​checked column by column: for a certain column (i.e., a certain electrical length value), if the fluctuation frequency corresponding to all 5 dielectric constant values ​​in the column is less than the fluctuation frequency threshold, then the electrical length value is a valid value; the minimum value among all valid values ​​is denoted as Ltolmin, and the maximum value is denoted as Ltolmax, thereby determining the compensation length tolerance interval as [Ltolmin, Ltolmax], the tolerance interval width is denoted as Wtol, Wtol is equal to Ltolmax minus Ltolmin, and the midpoint of the interval Ltolmid is equal to half of the sum of Ltolmin and Ltolmax.

[0063] In a specific embodiment, in step S433, the absolute value D of the difference between Lopt and Ltolmid is calculated, and D is compared with one-quarter of Wtol to determine the position of Lopt in the compensation length tolerance interval: if D is less than one-quarter of Wtol, it is determined that Lopt is located in the middle of the compensation length tolerance interval, Lopt is kept unchanged, and Lopt is directly used as the final optimal compensation length of the phase compensation open-circuit microstrip stub; if D is greater than or equal to one-quarter of Wtol, it is determined that Lopt is biased towards the edge of the compensation length tolerance interval, and Ltolmid is updated as the final optimal compensation length of the phase compensation open-circuit microstrip stub.

[0064] It is understandable that the reason why step S433 above uses one-quarter of Wtol as the quantization boundary for determining whether Lopt is biased towards the edge is because if the deviation of Lopt from the midpoint of the interval is less than one-quarter of the interval width, then the distance from Lopt to both ends of the interval is greater than one-quarter of the interval width. When the dielectric constant shifts in either direction, the actual electrical length deviates from Lopt by a margin of not less than one-quarter of the interval width within the compensation length tolerance range, which has sufficient manufacturing tolerance. Conversely, if the deviation is greater than or equal to one-quarter of the interval width, then Lopt is biased towards a certain side boundary. When the dielectric constant shifts in that side direction, the margin is insufficient. Taking the midpoint of the interval can distribute the margin evenly to both sides, making the tolerance in either direction equal.

[0065] Understandably, phase-compensated open-circuit microstrip stubs generate a reflected signal with a specific phase and amplitude by introducing controlled open-circuit terminal reflections into the transmission line. This reflected signal is vector-superimposed on the original multiple reflected signals at the interface of the duplexer and bandpass filter cascade. When the stub electrical length is appropriately chosen, the phase and amplitude relationships of the compensated reflection and the original multiple reflections at various frequency points tend to cancel each other out, making the equivalent group delay of each frequency component in the transition band tend to be consistent, thereby eliminating the amplitude oscillation of the return loss curve and reducing the fluctuation frequency to below the threshold. The upper limit of the stub electrical length scan is taken as the effective quarter wavelength corresponding to the center frequency of the transition band because the input impedance of the open-circuit microstrip stub changes from an open-circuit state to a short-circuit state as the electrical length increases. At the quarter wavelength, the stub is equivalent to a short circuit, and its reflected phase rotation has covered the entire effective compensation phase range. After exceeding this length, the compensation characteristics of the stub begin to repeat periodically. Continuing to increase the length does not produce new compensation capability, but instead increases the physical size and radiation loss of the stub. The introduction of two-dimensional scanning of dielectric constant tolerance is due to batch-to-batch fluctuations in the actual dielectric constant of the FR-4 substrate. An excess or deficiency in the dielectric constant causes the actual effective electrical length of the stub to shift in the same direction relative to the design value. If the optimal compensation length only holds true at the nominal dielectric constant value, batch deviations will cause the actual electrical length to deviate from the optimal value, causing the fluctuation frequency to rise above the threshold again. By verifying each stub within the entire dielectric constant tolerance range, the electrical length range that maintains the fluctuation frequency below the threshold under any dielectric constant deviation is identified. The median of this range is then used as the final fabrication parameter. This ensures that when the dielectric constant shifts in any direction, the actual stub performance maintains a symmetrical margin within the tolerance range, allowing the antenna design results to directly meet the consistency requirements of mass production without relying on batch-by-batch physical testing.

[0066] Step S5: Substitute the optimal compensation length into the parameterized simulation model for verification. After successful verification, convert the optimal compensation length into the physical dimensions of the phase-compensated open-circuit microstrip stub and output it as the plate-making parameter.

[0067] In a specific embodiment, step S5 first substitutes the final optimal compensation length determined in step S4 into the parametric simulation model, performs a full-band simulation under the condition that the dielectric constant of the PCB circuit board is a nominal value of 4.4, and verifies the simulation results. The verification indicators include the following three items: the standing wave ratio (SWR) in both the first and second operating frequency bands is less than the preset SWR threshold; the group delay fluctuation amplitude in the transition band is less than the preset group delay fluctuation amplitude threshold set in step S3; and the out-of-band signal suppression outside the cutoff frequency of the first operating frequency band and outside the starting frequency of the second operating frequency band is greater than or equal to the preset out-of-band suppression threshold. When all three indicators are met simultaneously, the verification is deemed successful. If any indicator is not met, the process returns to step S43 to re-execute the parametric scan, reduce the scan step size, and redetermine the optimal compensation length.

[0068] After successful verification, the final optimal compensation length will be converted from the electrical length to the physical length of the phase-compensated open-circuit microstrip stub. The conversion formula is as follows: ; Wherein, Lphys is the physical length of the phase-compensated open-circuit microstrip stub, in millimeters (mm); θopt is the electrical length corresponding to the final optimal compensation length, in degrees (°), ranging from 0° to 90°, determined by step S4; c is the speed of light in free space; fc is the center frequency of the transition band, in GHz, and is the arithmetic mean of the cutoff frequency of the first operating band and the starting frequency of the second operating band. Preferably, under the conditions of the first operating band cutoff frequency of 2480MHz and the second operating band starting frequency of 5150MHz, fc is approximately 3.8GHz; εeff is the effective dielectric constant of the FR-4 substrate at the center frequency of the transition band, dimensionless, ranging from 3.8 to 4.2, preferably 4.0, and is calibrated based on the correction relationship of the microstrip linewidth to substrate thickness on the relative dielectric constant.

[0069] Understandably, the conversion formula introduces the effective dielectric constant instead of directly using the nominal value of the relative dielectric constant 4.4 because the electromagnetic field in the microstrip transmission line is not completely confined inside the substrate. Part of the electric field is distributed in the air layer above the substrate, making the actual equivalent dielectric constant of the transmission line lower than the relative dielectric constant of the substrate. This correction allows the converted physical length to more accurately reflect the actual electrical behavior of the stub on the PCB, avoiding systematic deviations between the board size and simulation expectations.

[0070] The Lphys conversion result is rounded to the smallest unit of PCB fabrication precision (preferably 0.01mm), and together with the stub linewidth (consistent with the 7-linewidth of the 1-to-2 microstrip line), it is written into the PCB fabrication Gerber file as the fixed fabrication parameters for the phase compensation open-circuit microstrip stub. This parameter can be directly used for subsequent PCB prototyping and mass production without relying on actual finished product test feedback for adjustment.

[0071] Understandably, simulation verification and physical dimension conversion are necessary transitional steps in transforming purely digital design results into a manufacturable entity. Simulation verification performs a final consistency check on the design results in the digital domain, confirming that the three indicators—VSWR, group delay ripple, and out-of-band rejection—are simultaneously valid in the same simulation model, eliminating overall performance deviations caused by error accumulation between sub-steps S3 and S4. Physical dimension conversion transforms the stub parameters, expressed as electrical lengths in the simulation, into geometric dimensions expressed in millimeters in PCB manufacturing. The introduction of effective dielectric constant correction during the conversion process is because the actual electrical behavior of microstrip stubs is determined by their equivalent propagation environment, not solely by the bulk dielectric constant of the substrate. An accurate effective dielectric constant allows the conversion results to directly correspond to the phase response of the stubs on the actual PCB, thereby ensuring that the optimal compensation effect obtained in the simulation can be reproduced on the physical PCB, achieving a complete closed loop of design-to-manufacture.

[0072] The technical solution of the present invention has been described above with reference to the preferred embodiments shown in the accompanying drawings. However, it will be readily understood by those skilled in the art that the scope of protection of the present invention is obviously not limited to these specific embodiments. Without departing from the principles of the present invention, those skilled in the art can make equivalent changes or substitutions to the relevant technical features, and the technical solutions after these changes or substitutions will all fall within the scope of protection of the present invention.

Claims

1. A low-cost, anti-interference built-in antenna, characterized in that, include: PCB circuit board; An antenna radiator is disposed on the front side of the PCB circuit board. The radiation size of the antenna radiator is determined based on a quarter wavelength and covers the first operating frequency band and the second operating frequency band. A duplexer is connected to the signal output terminal of the antenna radiator. The duplexer uses a preset cutoff frequency between the first operating frequency band and the second operating frequency band as the branching boundary to divide the signal output by the antenna radiator into a low-frequency link and a high-frequency link. A bandpass filter is connected in series with the low-frequency link. The passband of the bandpass filter covers the first operating frequency band and suppresses out-of-band signals outside the first operating frequency band. A phase-compensated open-circuit microstrip stub is connected to the microstrip transmission line between the low-frequency output of the duplexer and the input of the bandpass filter, so that the oscillation frequency of the return loss curve of the antenna radiator port is less than a preset oscillation frequency threshold in the transition band between the first operating frequency band and the second operating frequency band after the duplexer and the bandpass filter are cascaded. A matching circuit, connected to the end of the low-frequency link, is used to tune the impedance of the low-frequency link so that the standing wave ratio (SWR) in both the first and second operating frequency bands is less than a preset SWR threshold. The antenna ground layer is located on the back of the PCB circuit board and is connected to the ground terminals of the duplexer, the bandpass filter, the phase-compensated open-circuit microstrip stub, and the matching circuit, respectively. The inner conductor pad and the outer conductor pad of the feeder are both disposed on the PCB circuit board. The inner conductor pad of the feeder is connected to the signal output terminal of the matching circuit, and the outer conductor pad of the feeder is connected to the antenna ground layer. A coaxial cable, wherein the inner conductor is soldered to the inner conductor pad of the feed line, and the outer conductor is soldered to the outer conductor pad of the feed line. A connector, which is connected to the end of the coaxial cable away from the PCB circuit board, is used to interface with the radio frequency interface of the terminal device.

2. The low-cost anti-interference built-in antenna according to claim 1, characterized in that, The matching circuit includes a matching capacitor connected in parallel between the signal line of the low-frequency link and the antenna ground layer.

3. A design method for a low-cost anti-interference built-in antenna, applied to the low-cost anti-interference built-in antenna described in claims 1-2, characterized in that, include: Step S1: Establish a parameterized simulation model of the low-cost anti-interference built-in antenna in the electromagnetic simulation software, and set the electrical length of the phase-compensated open-circuit microstrip stub as the variable to be determined, with the initial value set to zero. Step S2: Based on the parameterized simulation model, extract the extreme point frequency coordinates of the return loss curve of the antenna radiator port in the transition band between the first operating frequency band and the second operating frequency band, and determine the fluctuation frequency of the return loss curve in the transition band. Step S3: Determine the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude; and determine the fluctuation frequency threshold based on the mapping relationship and the preset group delay fluctuation amplitude threshold. Step S4: Based on the determination result of the fluctuation frequency and the fluctuation frequency threshold, determine the optimal compensation length of the phase compensation open-circuit microstrip stub. Step S5: Substitute the optimal compensation length into the parameterized simulation model for verification. After successful verification, convert the optimal compensation length into the physical dimensions of the phase-compensated open-circuit microstrip stub and output it as the plate-making parameter.

4. The design method for a low-cost anti-interference built-in antenna according to claim 3, characterized in that, Step S2 includes: Step S21: In the parameterized simulation model, the nominal value of the dielectric constant of the PCB circuit board is used for simulation to obtain the return loss curve of the antenna radiator port in the full frequency band. Step S22: On the return loss curve, locate the transition band region between the cutoff frequency of the first operating frequency band and the start frequency of the second operating frequency band, and extract the frequency coordinates of all local maxima and local minima of the return loss curve within the transition band region. Step S23: Based on the frequency coordinates of all the local maxima and minima, calculate the frequency interval between adjacent extreme points in order of frequency from low to high, and take the average value of all the frequency intervals to obtain the average frequency interval. Step S24: Determine the fluctuation frequency of the return loss curve within the transition band based on the reciprocal of the average frequency interval.

5. The design method for a low-cost anti-interference built-in antenna according to claim 4, characterized in that, In step S3, the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude is determined in the following way: In the parametric simulation model, the component parameters of the duplexer and the component parameters of the bandpass filter are each taken into several different sets of values ​​within their respective manufacturing tolerance ranges. Simulation is performed on each set of values ​​to obtain the corresponding fluctuation frequency and the corresponding group delay fluctuation amplitude, forming several sets of corresponding data points of fluctuation frequency and group delay fluctuation amplitude. Curve fitting is performed on all the corresponding data points to obtain the mapping relationship between the fluctuation frequency and the group delay fluctuation amplitude.

6. The design method for a low-cost anti-interference built-in antenna according to claim 5, characterized in that, Step S4 includes: Step S41: Compare the fluctuation frequency obtained in step S2 with the fluctuation frequency threshold obtained in step S3; Step S42: Based on the comparison result that the fluctuation frequency is less than or equal to the fluctuation frequency threshold, the optimal compensation length of the phase compensation open-circuit microstrip stub is determined to be zero, and step S4 ends; based on the comparison result that the fluctuation frequency is greater than the fluctuation frequency threshold, step S43 is executed. Step S43: Perform parametric scanning on the phase-compensated open-circuit microstrip stub, substitute each scan value into the parametric simulation model for simulation, and obtain the corresponding fluctuation frequency; based on the relationship between the fluctuation frequency and the electrical length, determine the electrical length corresponding to the minimum value that makes the fluctuation frequency less than the fluctuation frequency threshold as the optimal compensation length of the phase-compensated open-circuit microstrip stub.

7. The design method for a low-cost anti-interference built-in antenna according to claim 6, characterized in that, Step S43 includes: Step S431: The dielectric constant of the PCB circuit board and the electrical length of the phase compensation open-circuit microstrip stub are simultaneously set as scanning variables. The dielectric constant takes several discrete values ​​within the batch tolerance range of the PCB circuit board substrate, and the electrical length takes several discrete values ​​within a preset range with the optimal compensation length as the center. Simulation is performed on all combinations of the dielectric constant values ​​and the electrical length values ​​to obtain the fluctuation frequency corresponding to each combination. Step S432: Based on the fluctuation frequencies of all combinations obtained in step S431, determine the range of electrical length values ​​in which the fluctuation frequencies are all less than the fluctuation frequency threshold under all values ​​of the dielectric constant, and denot it as the compensation length tolerance range. Step S433: Update the optimal compensation length based on the positional relationship between the optimal compensation length and the compensation length tolerance interval.

8. The design method for a low-cost anti-interference built-in antenna according to claim 7, characterized in that, In step S433, if the optimal compensation length is located in the middle of the compensation length tolerance interval, the optimal compensation length remains unchanged; if the optimal compensation length is biased towards the edge of the compensation length tolerance interval, the median of the compensation length tolerance interval is updated to the optimal compensation length.

9. The design method for a low-cost anti-interference built-in antenna according to claim 8, characterized in that, In step S431, the upper limit of the scanning length of the phase-compensated open-circuit microstrip stub is the effective quarter wavelength of the transition band center frequency in the PCB substrate.

10. The design method for a low-cost anti-interference built-in antenna according to claim 9, characterized in that, In step S3, the mapping relationship is a polynomial fitting relationship between the fluctuation frequency and the group delay fluctuation amplitude, and the coefficients of each term of the polynomial fitting relationship are determined by fitting calculation based on all corresponding data points.

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