Reconfigurable antenna subarray, phased array antenna and millimeter wave communication system

Abstract

The application discloses a directional diagram reconfigurable antenna subarray, a phased array antenna and a millimeter wave communication system, and relates to the technical field of antennas. The subarray comprises a plurality of antenna units, a feed network, a plurality of parasitic units and a plurality of radio frequency switches. The feed network near-field couples the antenna units to excite a main radiation field; the parasitic units near-field couple the antenna units and are connected to loads to generate a secondary radiation field; the first port of the radio frequency switch is connected to the parasitic units; by adjusting the voltage polarity combination of the direct current voltage signals connected to the first and second control ends of the radio frequency switch, the switching of the load impedance between the open circuit state and the short circuit state is realized, so as to control the phase of the secondary radiation field to reconfigure the radiation directional diagram of the directional diagram reconfigurable antenna subarray, thereby solving the problems of serious grating lobes of the existing W-wave band large-interval subarray, large scanning loss, limited applicability of the traditional reconfigurable scheme and high series loss of the radio frequency switch, and improving the wide-angle scanning performance and link efficiency.

Description

Technical Field

[0001] This invention relates to the field of antenna technology, and in particular to a pattern-reconfigurable antenna subarray, a phased array antenna, and a millimeter-wave communication system. Background Technology

[0002] The W-band (75-110 GHz) is widely used in millimeter-wave radar, satellite communications, and precision guidance due to its abundant spectrum resources and high-resolution imaging capabilities. To overcome the significant transmission loss in millimeter-wave space, phased array antennas often employ large-scale arrays and beamforming techniques. However, achieving high-power, wide-angle scanning phased arrays in the W-band presents a serious challenge. A phased array antenna consists of multiple antenna elements arranged at a certain spacing, with beam scanning achieved by controlling the feed phase of each antenna element. According to phased array theory, to ensure no grating lobes are generated within the ±θ_max scanning range, the antenna element spacing d must satisfy d < λ / (1 + |sinθ_max|). For ±30° scanning, d < 0.67λ0 is required. In the W-band, to obtain high transmit power (>20 dBm), advanced integrated RF front-end chips are typically required. These chips have physical dimensions much larger than the antenna element dimensions, making it difficult to achieve half-wavelength spacing. Therefore, a 2×2 subarray architecture is often used in engineering, feeding the signal from one RF channel to four antenna elements simultaneously via a power divider, thereby effectively increasing the control area of ​​each channel. This forces the antenna element spacing to increase to 1.3λ0. With this large spacing, traditional phased arrays exhibit significant grating lobes (>-5 dB) and scanning losses (>5 dB) during ±30° scanning. Furthermore, the applicability of traditional pattern reconfigurable schemes is limited in the W-band; active devices such as PIN diodes and varactors have excessively large parasitic parameters and low self-resonant frequencies in this band, making reliable operation difficult. In addition, traditional reconfigurable antennas often connect RF switches in series at the antenna feed, resulting in insertion losses greater than 2 dB and directly degrading link efficiency. Moreover, when multiple subarrays are spliced ​​into a large array, the effective spacing between subarrays further increases (approximately 2.6λ0), and without controlling the subarray pattern, grating lobes and gain drops become unacceptable. Summary of the Invention

[0003] The main objective of this invention is to propose a pattern-reconfigurable antenna subarray, a phased array antenna, and a millimeter-wave communication system. This invention aims to solve the problems of severe grating lobes and high scanning loss in existing W-band large-pitch subarrays, limited applicability of traditional reconfigurable schemes, and high series loss of RF switches, thereby improving wide-angle scanning performance and link efficiency.

[0004] To achieve the above objectives, the present invention proposes a pattern-reconfigurable antenna subarray, comprising: Multiple antenna elements are arranged in a rectangular array, and the distance between two adjacent antenna elements is not less than 1.3 times the operating wavelength; A feeding network is connected to the antenna elements in the near field, and is used to feed the incoming radio frequency signal into the antenna elements to excite the antenna elements to generate the main radiation field. Multiple parasitic units are configured and connected in near field to each of the multiple antenna units, and are used to sense the main radiation field of the corresponding antenna unit and generate a secondary radiation field. Each parasitic unit is connected to a load. Multiple radio frequency switches, each radio frequency switch having a first port, a second port, a third port, a first control terminal, and a second control terminal, the multiple first ports being connected one-to-one with the multiple parasitic units, the second port being connected to a short-circuit structure, the third port being connected to an open-circuit structure, the first control terminal and the second control terminal being used to receive DC voltage signals of the same magnitude but opposite polarity; By changing the voltage polarity combination of the DC voltage signals connected to the first and second control terminals of the multiple radio frequency switches, the load impedance state connected to each parasitic unit is switched between an open-circuit state and a short-circuit state, thereby regulating the phase of the secondary radiation field to reconstruct the radiation pattern of the pattern-reconstructable antenna subarray.

[0005] In one embodiment, the number of the antenna element and the number of the radio frequency switch are four; When the first control terminals of the four radio frequency switches are all connected to DC voltage signals of the same polarity, the load impedance connected to the four parasitic units is in an open circuit state, so that the radiation pattern does not deflect and points to the normal. When the first control terminals of the two RF switches in the first row of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals of the two RF switches in the second row is opposite, the load impedance states of the two parasitic units in the same row are the same, and the load impedance states of the parasitic units in the two rows are opposite, so that the composite beam of the pattern reconfigurable antenna subarray is deflected on the H-plane. When the first control terminals of the two RF switches in the first column of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals of the two RF switches in the second column is opposite, the load impedance states of the two parasitic units in the same column are the same, and the load impedance states of the parasitic units in the two columns are opposite, so that the synthesized beam is deflected on the E plane.

[0006] In one embodiment, when the first control terminal of the two radio frequency switches in the first row is connected to a negative DC voltage signal, the load impedance state of the two parasitic units in the first row is open circuit, so that the synthesized beam will be positively deflected on the H plane. When the first control terminals of the two RF switches in the first row are connected to a positive DC voltage signal, the load impedance of the two parasitic units in the first row is in a short-circuit state, so that the synthesized beam will be deflected in the reverse direction on the H plane. When the first control terminals of the two RF switches in the first column are connected to a negative DC voltage signal, the load impedance connected to the two parasitic units in the first column is in an open circuit state, so that the synthesized beam will be positively deflected on the E plane. When the first control terminals of the two RF switches in the first column are connected to a positive DC voltage signal, the load impedance of the two parasitic units in the first column is in a short-circuit state, so that the synthesized beam is deflected in the reverse direction on the E plane.

[0007] In one embodiment, the pattern-reconfigurable antenna subarray further includes: The packaging carrier includes a first metal layer, a second metal layer, a third metal layer, a fourth metal layer, a fifth metal layer, a sixth metal layer and a seventh metal layer stacked sequentially from bottom to top; Multiple quasi-coaxial vertical structures, each of the quasi-coaxial vertical structures passing through the seventh metal layer, the sixth metal layer, the fifth metal layer, the fourth metal layer and the third metal layer and extending to the second metal layer, one end of each quasi-coaxial vertical structure being electrically connected to a parasitic unit and the other end being electrically connected to a radio frequency switch; Among them, a plurality of the parasitic units are disposed in the seventh metal layer; Multiple antenna elements are disposed on the sixth metal layer and are configured one-to-one with multiple parasitic elements; The fifth metal layer is a ground plane, and the fifth metal layer is provided with metallized vias. The antenna unit is connected to the fifth metal layer through the metallized vias. The feed network is disposed on the fourth metal layer, and the feed network passes through the fifth metal layer and extends to the sixth metal layer to be connected to the antenna element in the near field. The third metal layer is a floor layer. The third metal layer has a through groove on the side opposite to the fourth metal layer that passes through the second metal layer and the first metal layer. A plurality of radio frequency switches are disposed in the through groove, and the material of the peripheral wall of the through groove is an insulating material. The open-circuit structure and the short-circuit structure are disposed on the first metal layer.

[0008] In one embodiment, a plurality of the radio frequency switches are located between the third metal layer and the second metal layer.

[0009] In one embodiment, the power supply network includes: The first type of coaxial vertical transition structure penetrates the fifth metal layer and extends to the sixth metal layer to be near-field coupled to the antenna element; The second type of coaxial vertical transition structure penetrates the third metal layer, the second metal layer and the first metal layer. The second type of coaxial vertical transition structure is used to connect to external radio frequency circuits. A power divider is disposed on the fourth metal layer, and the power divider connects the first type of coaxial vertical transition structure and the second type of coaxial vertical transition structure.

[0010] In one embodiment, each antenna element includes two electric dipoles symmetrically arranged about a first direction, each electric dipole includes two metal patches symmetrically arranged about a second direction, each metal patch is connected to the fifth metal layer through a metallized via, a gap is formed between the two metal patches of the same electric dipole, the two gaps constitute a magnetic dipole, and the second direction intersects the first direction; the end of the first type of coaxial vertical transition structure away from the power divider is located between the two electric dipoles to be near-field coupled to the two electric dipoles.

[0011] In one embodiment, each of the metal patches includes a first side and a second side disposed opposite to each other along a second direction, and a third side and a fourth side disposed opposite to each other along a first direction. The third side includes a bent segment and a straight segment. The bent segment extends from the first side toward the second side and is inclined toward a side away from the fourth side. The straight segment extends from the bent segment toward the second side.

[0012] The present invention also proposes a pattern reconfigurable phased array antenna, comprising multiple pattern reconfigurable antenna subarrays as described above, wherein the multiple pattern reconfigurable antenna subarrays are arranged in a rectangular array, and the spacing between two adjacent pattern reconfigurable antenna subarrays is not less than 1.3 times the operating wavelength.

[0013] The present invention also proposes a millimeter-wave communication system, including a pattern-reconfigurable phased array antenna as described above.

[0014] In the technical solution of this invention, by setting up a feeding network to be connected to multiple antenna elements in the near field, the incoming radio frequency signal is fed into multiple antenna elements to excite multiple antenna elements to generate the main radiation field, so that even when the distance between two adjacent antenna elements is not less than 1.3 times the working wavelength, it can still be adapted to radio frequency front-end chips with large physical size to obtain high transmission power. This invention sets up multiple parasitic units, each corresponding to a multiple antenna unit and connected in near-field coupling, to sense the main radiation field of the corresponding antenna unit and generate a secondary radiation field. Each parasitic unit is connected to a load, and multiple radio frequency (RF) switches are also provided. Each RF switch has a first port, a second port, a third port, a first control terminal, and a second control terminal. The multiple first ports are connected to the multiple parasitic units one-to-one, the second ports are connected to a short-circuit structure, and the third ports are connected to an open-circuit structure. The first and second control terminals are used to receive DC voltage signals of the same magnitude but opposite polarities. By changing the voltage polarity combination of the DC voltage signals received by the first and second control terminals of the multiple RF switches, the load impedance state connected to each parasitic unit is switched between an open-circuit state and a short-circuit state, thereby regulating the phase of the secondary radiation field and reconstructing the radiation pattern of the reconstructable antenna subarray. This configuration allows the pattern-reconfigurable antenna subarray to compensate for and correct the main radiation field using the secondary radiation field generated by parasitic elements, even with a large spacing of at least 1.3 times the operating wavelength between adjacent antenna elements. This effectively suppresses grating lobes and reduces scanning loss, alleviating the conflict between high-power transmission requirements and wide-angle scanning. Since the RF switch is not connected in series in the main feed link between the feed network and multiple antenna elements but is instead connected to the load end of the parasitic element, the additional insertion loss introduced by the series RF switch in traditional reconfigurable antennas, which degrades link efficiency, is avoided. Simultaneously, this approach avoids the problem of excessively large parasitic parameters and low self-resonance frequencies of active devices such as PIN diodes and varactors in the W-band, leading to unreliable operation and improving the applicability and operational stability of the pattern-reconfigurable antenna subarray in the W-band. Furthermore, by controlling the phase of the secondary radiation field, the radiation pattern of the pattern-reconfigurable antenna subarray can be reconstructed. This helps to control the subarray pattern when multiple pattern-reconfigurable antenna subarrays are spliced ​​into a large array and the equivalent spacing between the subarrays is about 2.6 times the operating wavelength, thereby reducing grating lobe and gain drop phenomena and improving the overall scanning performance of the large array. Attached Figure Description

[0015] To more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of the present invention. For those skilled in the art, other drawings can be obtained based on the structures shown in these drawings without creative effort.

[0016] Figure 1 This is a schematic diagram of the overall structure of the pattern-reconfigurable antenna subarray of the present invention; Figure 2 This is a circuit diagram of the pattern-reconfigurable antenna subarray of the present invention; Figure 3 This is a summary diagram of the five pattern states of the pattern-reconfigurable antenna subarray of the present invention and their corresponding control methods; Figure 4 for Figure 3 The radiation patterns of the reconfigurable antenna subarray in state 1 are: E-plane co-polarization (E Co-pol), E-plane cross-polarization (E X-pol), H-plane co-polarization (H Co-pol), and H-plane cross-polarization (H X-pol). Figure 5 for Figure 3 The radiation patterns of the reconfigurable antenna subarray in state 2 are H-plane co-polarized (H Co-pol) and H-plane cross-polarized (H X-pol) radiation patterns. Figure 6 for Figure 3 The radiation patterns of the reconfigurable antenna subarray in state 3 are H-plane co-polarized (H Co-pol) and H-plane cross-polarized (H X-pol) radiation patterns. Figure 7 for Figure 3 The radiation patterns of the reconfigurable antenna subarray in state 4 are E-plane co-polarized (E Co-pol) and E-plane cross-polarized (E X-pol). Figure 8 for Figure 3 The radiation patterns of the reconfigurable antenna subarray in state 5 are E-plane co-polarized (E Co-pol) and E-plane cross-polarized (E X-pol). Figure 9 This is a front view of the pattern-reconfigurable antenna subarray of the present invention; Figure 10 This is a schematic diagram of the overall structure of the antenna unit of the present invention; Figure 11 This is a planar structural diagram of the antenna element of the present invention; Figure 12 This is a graph showing the return loss of the pattern-reconfigurable antenna subarray of the present invention as a function of frequency. Figure 13 This is a graph showing the efficiency of the pattern-reconfigurable antenna subarray of the present invention as a function of frequency. Figure 14 This is a graph showing the gain of the pattern-reconfigurable antenna subarray of the present invention as a function of frequency. Figure 15The radiation patterns of the pattern reconfigurable antenna subarray of the present invention are the E-plane and H-plane co-polarized (Co-pol) and cross-polarized (X-pol) radiation patterns at an operating frequency of 85 GHz. Figure 16 The radiation patterns of the pattern reconfigurable antenna subarray of the present invention are the E-plane and H-plane co-polarized (Co-pol) and cross-polarized (X-pol) radiation patterns at the operating frequency of 94 GHz. Figure 17 The radiation patterns of the pattern reconfigurable antenna subarray of the present invention are the E-plane and H-plane co-polarized (Co-pol) and cross-polarized (X-pol) radiation patterns at an operating frequency of 100 GHz; Figure 18 To realize the pattern reconfigurable phased array antenna of the present invention in the H-plane Radiation gain pattern of continuous beam scanning from 40° to +40°; Figure 19 To realize the pattern-reconfigurable phased array antenna of the present invention in the E-plane Radiation gain pattern of continuous beam scanning from 30° to +30°.

[0017] Explanation of icon numbers: 100. Pattern-reconfigurable antenna subarray; 10. Antenna element; 11. Electric dipole; 111. Metal patch; 1111. First side; 1112. Second side; 1113. Third side; 11131. Bent section; 11132. Straight section; 1114. Fourth side; 12. Magnetic dipole; 20. Feed network; 21. Type I coaxial vertical transition structure; 22. Type II coaxial vertical transition structure; 23. Power divider; 30. Parasitic element; 40. RF switch; 41. First RF switch; 4 2. Second RF switch; 43. Third RF switch; 44. Fourth RF switch; 411. First port; 412. Second port; 413. Third port; V1. First control terminal; V2. Second control terminal; 50. Packaging carrier; 51. First metal layer; 52. Second metal layer; 53. Third metal layer; 54. Fourth metal layer; 55. Fifth metal layer; 56. Sixth metal layer; 57. Seventh metal layer; 501. Through slot; 60. Quasi-coaxial vertical structure; 70. Open circuit structure; 80. Short circuit structure.

[0018] The realization of the objective, functional features and advantages of the present invention will be further explained in conjunction with the embodiments and with reference to the accompanying drawings. Detailed Implementation

[0019] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative effort are within the scope of protection of the present invention.

[0020] The W-band (75-110 GHz) is widely used in millimeter-wave radar, satellite communications, and precision guidance due to its abundant spectrum resources and high-resolution imaging capabilities. To overcome the significant transmission loss in millimeter-wave space, phased array antennas often employ large-scale arrays and beamforming techniques. However, achieving high-power, wide-angle scanning phased arrays in the W-band presents a serious challenge. A phased array antenna consists of multiple antenna elements arranged at a certain spacing, with beam scanning achieved by controlling the feed phase of each antenna element. According to phased array theory, to ensure no grating lobes are generated within the ±θ_max scanning range, the antenna element spacing d must satisfy d < λ / (1 + |sinθ_max|). For ±30° scanning, d < 0.67λ0 is required. In the W-band, to obtain high transmit power (>20 dBm), advanced integrated RF front-end chips are typically required. These chips have physical dimensions much larger than the antenna element dimensions, making it difficult to achieve half-wavelength spacing. Therefore, a 2×2 subarray architecture is often used in engineering, feeding the signal from one RF channel to four antenna elements simultaneously via a power divider, thereby effectively increasing the control area of ​​each channel. This forces the antenna element spacing to increase to 1.3λ0. With this large spacing, traditional phased arrays exhibit significant grating lobes (>-5 dB) and scanning losses (>5 dB) during ±30° scanning. Furthermore, the applicability of traditional pattern reconfigurable schemes is limited in the W-band; active devices such as PIN diodes and varactors have excessively large parasitic parameters and low self-resonant frequencies in this band, making reliable operation difficult. In addition, traditional reconfigurable antennas often connect RF switches in series at the antenna feed, resulting in insertion losses greater than 2 dB and directly degrading link efficiency. Moreover, when multiple subarrays are spliced ​​into a large array, the effective spacing between subarrays further increases (approximately 2.6λ0), and without controlling the subarray pattern, grating lobes and gain drops become unacceptable.

[0021] This invention proposes a pattern-reconfigurable antenna subarray 100.

[0022] Please see Figure 1 and Figure 2 In one embodiment of the present invention, the pattern-reconfigurable antenna subarray 100 includes: Multiple antenna elements 10 are arranged in a rectangular array, and the spacing between two adjacent antenna elements 10 is not less than 1.3 times the operating wavelength; The feed network 20 is connected to multiple antenna elements 10 in the near field and is used to feed the incoming radio frequency signal into the antenna elements 10 to excite the multiple antenna elements 10 to generate the main radiation field. Multiple parasitic units 30 are configured one-to-one with multiple antenna units 10 and are connected in the near field. They are used to sense the main radiation field of the corresponding antenna unit 10 and generate a secondary radiation field. Each parasitic unit 30 is connected to a load. Multiple radio frequency switches 40, each radio frequency switch 40 having a first port 411, a second port 412, a third port 413, a first control terminal V1, and a second control terminal V2. The multiple first ports 411 are connected to multiple parasitic units 30 one by one. The second port 412 is connected to a short-circuit structure 80, and the third port 413 is connected to an open-circuit structure 70. The first control terminal V1 and the second control terminal V2 are used to receive DC voltage signals with the same voltage magnitude but opposite polarity. By changing the voltage polarity combination of the DC voltage signals connected to the first control terminal V1 and the second control terminal V2 of multiple RF switches 40, the load impedance state connected to each parasitic unit 30 is switched between an open circuit state and a short circuit state, thereby regulating the phase of the secondary radiation field to reconstruct the radiation pattern of the pattern-reconstructable antenna subarray 100.

[0023] In this embodiment, as mentioned earlier, in the application scenarios of W-band phased array antennas, in order to obtain a high transmit power greater than 20 dBm, it is usually necessary to use an advanced process integrated RF front-end chip. However, the physical size of such RF front-end chips is much larger than the size of antenna element 10, making it difficult to achieve the theoretical half-wavelength spacing. Therefore, in engineering, a 2×2 subarray architecture is often used to feed the signal of one RF channel to four antenna elements 10 simultaneously via a power divider 23. This forces the spacing between antenna elements 10 to increase to 1.3 times the operating wavelength. Under this large spacing condition, traditional phased arrays have severe grating lobes and large scanning losses when scanning at ±30°. This embodiment arranges multiple antenna elements 10 in a rectangular array with a spacing of not less than 1.3 times the operating wavelength between adjacent antenna elements 10. This is precisely to adapt to the physical size limitations of W-band RF front-end chips and the layout requirements of the 2×2 subarray architecture. This allows the pattern-reconfigurable antenna subarray 100 to maintain a normal array arrangement under the physical constraints of large spacing to meet high power requirements, and is compatible with the size specifications of existing W-band high-power RF front-end chips and the 2×2 subarray feeding topology.

[0024] Meanwhile, in traditional reconfigurable antenna schemes, the RF switch 40 is often connected in series at the antenna feed end, resulting in an insertion loss greater than 2 dB and directly degrading link efficiency. In this embodiment, the feed network 20 uses near-field coupling to transmit the RF signal to multiple antenna elements 10, eliminating the need for series active switching devices in the RF signal transmission path. This avoids the link efficiency degradation caused by the additional insertion loss introduced by the RF switch 40 connected in series with the main feed link in traditional schemes. Under the near-field coupling excitation of the feed network 20, multiple antenna elements 10 generate a main radiation field. This main radiation field serves as the fundamental radiation component of the pattern reconfigurable antenna subarray 100. Its energy is transferred from the feed network 20 to the multiple antenna elements 10 via non-contact near-field coupling, reducing ohmic loss and reflection loss in the conduction path. This ensures that each antenna element 10 can obtain uniform and low-loss signal excitation under a large-spacing layout, enabling the main radiation field of the pattern reconfigurable antenna subarray 100 to be effectively excited under low-loss conditions.

[0025] Furthermore, when multiple subarrays are spliced ​​into a large array, the equivalent spacing between the subarrays increases further. Without controlling the subarray radiation pattern, grating lobe and gain drop issues become unacceptable. To address this problem, in this embodiment, multiple parasitic elements 30 are configured one-to-one with multiple antenna elements 10 and are connected via near-field coupling. The multiple parasitic elements 30 do not directly receive the RF signal from the feed network 20, but instead sense the primary radiation field of the corresponding antenna element 10 through near-field coupling, thereby generating a secondary radiation field. The secondary radiation field and the primary radiation field are vector-superimposed in space to form the composite radiation field of the pattern-reconstructable antenna subarray 100. This composite radiation field can correct the array factor distortion caused by the large spacing by adjusting the amplitude and phase distribution of the secondary radiation field, thereby suppressing grating lobe levels and compensating for scanning gain loss during wide-angle scanning. That is, the pattern reconfigurable antenna subarray 100 uses the secondary radiation field generated by the parasitic element 30 to dynamically compensate the main radiation field, so that the synthesized beam can still maintain low sidelobes and stable gain when it deviates from the normal scan, thus alleviating the problem of deterioration of scanning performance inherent in large-pitch arrays.

[0026] Given the limited applicability of traditional pattern reconfigurable schemes in the W-band, active devices such as PIN diodes or varactors exhibit excessively large parasitic parameters and low self-resonant frequencies in this band, making reliable operation difficult. In this embodiment, multiple RF switches 40 operate on the load branch of the parasitic unit 30 rather than the main feed link between the feed network 20 and multiple antenna elements 10. Their insertion loss is not directly superimposed on the RF signal transmission path, thus further avoiding the deterioration of link efficiency caused by the additional insertion loss introduced by the series connection of RF switches 40 in traditional schemes. Furthermore, from an electromagnetic theory perspective, the amplitude and phase of the induced current on the parasitic unit 30 are jointly determined by three key factors: the mutual impedance between the antenna element 10 and the parasitic unit 30, the self-impedance of the parasitic unit 30 itself, and the load impedance connected to the parasitic unit 30. The mutual impedance is determined by their relative positions, while the self-impedance is determined by the geometry of the parasitic unit 30 itself. To ensure that the secondary radiation fields generated by each parasitic element 30 in the pattern-reconfigurable antenna subarray 100 are consistent and to avoid unexpected beam deflection effects, it is necessary to ensure that the electromagnetic environment of each parasitic element 30 is symmetrical and that the connected load impedance states are synchronized. Therefore, each RF switch 40 has a first port 411, a second port 412, a third port 413, a first control terminal V1, and a second control terminal V2. Multiple first ports 411 are connected one-to-one with multiple parasitic elements 30. The second port 412 is connected to a short-circuit structure 80, and the third port 413 is connected to an open-circuit structure 70. By applying DC voltage signals of the same magnitude but opposite polarity to the first control terminal V1 and the second control terminal V2, the internal path of the RF switch 40 is switched, thereby switching the load impedance state connected to the parasitic element 30 between an open-circuit state and a short-circuit state. When the load impedance is in an open-circuit state, the parasitic element 30 exhibits specific reactance characteristics. When the load impedance is in a short-circuit state, the reactance characteristics of the parasitic element 30 are reversed. These two drastically different boundary conditions cause a significant difference in the phase of the induced current in the parasitic element 30, thereby modulating the phase of the secondary radiation field to reconstruct the radiation pattern of the pattern-reconfigurable antenna subarray 100. This method avoids the limitations of active devices such as PIN diodes or varactors in the W-band, which are difficult to operate reliably due to excessive parasitic parameters and low self-resonant frequencies. This allows the pattern-reconfigurable antenna subarray 100 to achieve stable pattern reconstruction through pure switching discrete state switching in the W-band high-frequency environment, while maintaining low RF link loss.

[0027] In the technical solution of the present invention, by setting the feeding network 20 to be connected to multiple antenna elements 10 in the near field, the incoming radio frequency signal is fed into multiple antenna elements 10 to excite multiple antenna elements 10 to generate the main radiation field, so that when the distance between two adjacent antenna elements 10 is not less than 1.3 times the working wavelength, it can still adapt to radio frequency front-end chips with large physical size to obtain high transmission power. This invention sets up multiple parasitic units 30, which are correspondingly arranged and near-field coupled to multiple antenna units 10. These units are used to sense the main radiation field of the corresponding antenna unit 10 and generate a secondary radiation field. Each parasitic unit 30 is connected to a load. Multiple radio frequency switches 40 are also provided. Each radio frequency switch 40 has a first port 411, a second port 412, a third port 413, a first control terminal V1, and a second control terminal V2. The multiple first ports 411 are connected to the multiple parasitic units 30 in a one-to-one correspondence. The second port 412 is connected to a short-circuit structure 80, and the third port 413 is connected to an open-circuit structure 70. The first control terminal V1 and the second control terminal V2 are used to receive DC voltage signals of the same magnitude but opposite polarity. By changing the voltage polarity combination of the DC voltage signals received by the first control terminals V1 and the second control terminals V2 of the multiple radio frequency switches 40, the load impedance state connected to each parasitic unit 30 is switched between an open-circuit state and a short-circuit state, thereby regulating the phase of the secondary radiation field and reconstructing the radiation pattern of the reconstructable antenna subarray 100. This configuration allows the pattern-reconfigurable antenna subarray 100 to compensate for and correct the main radiation field using the secondary radiation field generated by the parasitic element 30, even with a large spacing of at least 1.3 times the operating wavelength between adjacent antenna elements 10. This effectively suppresses grating lobes and reduces scanning loss, alleviating the conflict between high-power transmission requirements and wide-angle scanning. Since the RF switch 40 is not connected in series in the main feed link between the feed network 20 and the multiple antenna elements 10 but is connected to the load end of the parasitic element 30, the additional insertion loss introduced by the series connection of the RF switch 40 in traditional reconfigurable antennas, which degrades link efficiency, is avoided. Simultaneously, this approach avoids the problem of excessively large parasitic parameters and low self-resonance frequencies of active devices such as PIN diodes and varactors in the W-band, leading to unreliable operation and improving the applicability and operational stability of the pattern-reconfigurable antenna subarray 100 in the W-band. Furthermore, by controlling the phase of the secondary radiation field, the radiation pattern of the pattern-reconfigurable antenna subarray 100 can be reconstructed. This helps to control the subarray pattern when multiple pattern-reconfigurable antenna subarrays 100 are spliced ​​into a large array and the equivalent spacing between the subarrays is about 2.6 times the operating wavelength, thereby reducing grating lobe and gain drop phenomena and improving the overall scanning performance of the large array.

[0028] like Figures 1 to 3 As shown, in one embodiment, there are four antenna elements 10 and four radio frequency switches 40. When the first control terminal V1 of the four RF switches 40 is connected to a DC voltage signal of the same polarity, the load impedance connected to the four parasitic units 30 is in an open circuit state so that the radiation pattern does not deflect and points to the normal. When the first control terminals V1 of the two RF switches 40 in the first row of the rectangular array are connected to DC voltage signals of the same polarity and the polarity of the DC voltage signals connected to the first control terminals V1 of the two RF switches 40 in the second row is opposite, the load impedance states of the two parasitic units 30 in the same row are the same, and the load impedance states of the parasitic units 30 in the two rows are opposite, so that the composite beam of the pattern reconfigurable antenna subarray 100 is deflected on the H plane. When the first control terminals V1 of the two RF switches 40 in the first column of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals V1 of the two RF switches 40 in the second column is opposite, the load impedance states of the two parasitic units 30 in the same column are the same, and the load impedance states of the parasitic units 30 in the two columns are opposite, so that the synthesized beam is deflected on the E plane.

[0029] In this embodiment, there are four antenna elements 10 and four radio frequency switches 40. This number configuration corresponds to a 2×2 subarray architecture, so that the four antenna elements 10 are arranged in a rectangular array and the spacing between two adjacent antenna elements 10 is not less than 1.3 times the operating wavelength. At the same time, the four radio frequency switches 40 are connected to the four parasitic units 30 one by one to realize independent control of the load impedance state of the four parasitic units 30.

[0030] When the first control terminal V1 of the four RF switches 40 is connected to a DC voltage signal of the same polarity, the load impedance connected to the four parasitic units 30 is in an open circuit state. At this time, the secondary radiation field generated by the four parasitic units 30 is in phase. The composite radiation field formed by the superposition of the secondary radiation field and the main radiation field is symmetrical about the normal, so that the radiation pattern does not deflect and points to the normal, which is suitable for detection or communication scenarios facing the target.

[0031] When the first control terminals V1 of the two RF switches 40 in the first row of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals V1 of the two RF switches 40 in the second row is opposite, the load impedance states of the two parasitic units 30 in the same row are the same, and the load impedance states of the parasitic units 30 in the first and second rows are alternately distributed as open circuit and short circuit states. As a result, a phase gradient along the row direction is introduced between the antenna units 10 in the first and second rows, so that the composite beam of the pattern reconfigurable antenna subarray 100 is deflected in the H plane. This deflection effect can be used to compensate for the grating lobe or gain drop caused by the increased spacing when the large array is scanned in the H plane.

[0032] When the first control terminals V1 of the two RF switches 40 in the first column of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals V1 of the two RF switches 40 in the second column is opposite, the load impedance states of the two parasitic units 30 in the same column are the same, and the load impedance states of the parasitic units 30 in the first and second columns also show an alternating distribution of open circuit and short circuit states. As a result, a phase gradient along the column direction is introduced between the antenna units 10 in the first and second columns, causing the synthesized beam to deflect on the E plane. This deflection effect can be used to compensate for the grating lobes or gain drop caused by the increased spacing when the large array is scanned on the E plane.

[0033] By switching between the three voltage polarity combinations mentioned above, the pattern reconfigurable antenna subarray 100 can reconstruct the radiation pattern between the normal, H-plane deflection, and E-plane deflection states by utilizing the phase modulation of the secondary radiation field of the parasitic unit 30 without changing the physical structure of the feed network 20. This adapts to the dynamic control requirements of the subarray radiation pattern when multiple pattern reconfigurable antenna subarrays 100 are spliced ​​into a large array, which helps to alleviate the grating lobe and gain drop phenomena that occur when the equivalent spacing of the subarrays is about 2.6 times the operating wavelength, and improves the overall scanning performance of the large array.

[0034] like Figures 1 to 3 As shown, in one embodiment, when the first control terminal V1 of the two radio frequency switches 40 in the first row is connected to a negative DC voltage signal, the load impedance connected to the two parasitic units 30 in the first row is in an open circuit state, so that the synthesized beam will be positively deflected on the H plane. When the first control terminal V1 of the two RF switches 40 in the first row is connected to a positive DC voltage signal, the load impedance connected to the two parasitic units 30 in the first row is in a short-circuit state, so that the synthesized beam will be deflected in the reverse direction on the H plane. When the first control terminal V1 of the two RF switches 40 in the first column is connected to a negative DC voltage signal, the load impedance connected to the two parasitic units 30 in the first column is in an open circuit state, so that the synthesized beam will be positively deflected on the E plane. When the first control terminal V1 of the two RF switches 40 in the first column is connected to a positive DC voltage signal, the load impedance connected to the two parasitic units 30 in the first column is in a short-circuit state, so that the synthesized beam will be deflected in the reverse direction on the E plane.

[0035] In this embodiment, the four radio frequency (RF) switches 40 include a first RF switch 41, a second RF switch 42, a third RF switch 43, and a fourth RF switch 44. The first RF switch 41 and the second RF switch 42 are located in the first row, the third RF switch 43 and the fourth RF switch 44 are located in the second row, the first RF switch 41 and the third RF switch 43 are located in the first column, and the second RF switch 42 and the fourth RF switch 44 are located in the second column. Each of the first RF switch 41, the second RF switch 42, the third RF switch 43, and the fourth RF switch 44 has a first control terminal V1 and a second control terminal V2. By applying a combination of DC voltage signals of different magnitudes and polarities to the first control terminal V1 and the second control terminal V2, the internal channels of the RF switch 40 can be driven to switch to an on or off state. This causes the load impedance state connected to the corresponding parasitic unit 30 to switch between an open-circuit state and a short-circuit state, thereby regulating the phase of the secondary radiation field and reconstructing the radiation pattern of the reconstructable antenna subarray 100.

[0036] like Figure 3 As shown, when the first control terminal V1 of the first RF switch 41, the second RF switch 42, the third RF switch 43, and the fourth RF switch 44 are all connected to -2V and the second control terminal V2 is all connected to 2V, the reconfigurable pattern antenna subarray 100 is in state 1. At this time, the load impedance connected to the first RF switch 41, the second RF switch 42, the third RF switch 43, and the fourth RF switch 44 is in an open-circuit state, meaning that the four parasitic elements 30 all present a high-impedance open-circuit state. The resulting secondary radiation field is spatially in phase with the main radiation field and is symmetrically distributed about the normal direction. Therefore, the radiation pattern of the reconfigurable pattern antenna subarray 100 remains undeflected and points towards the normal direction. Figure 4 As shown, in state 1, the gain of the pattern reconfigurable antenna subarray 100 reaches a maximum of 11.5 dBi, indicating that the energy focusing effect is better at this time, which is suitable for application scenarios that require high-sensitivity detection of targets or long-distance communication. It also verifies that the 2×2 subarray architecture itself has good basic radiation efficiency without any deflection control intervention.

[0037] like Figure 3As shown, when the first control terminal V1 of the first RF switch 41 and the second RF switch 42 is set to -2V and the second control terminal V2 is set to 2V, and simultaneously, the third RF switch 43 and the fourth RF switch 44 are switched to the first control terminal V1 being set to 2V and the second control terminal V2 being set to -2V, that is, the reconfigurable antenna subarray 100 enters state 2. At this time, the load impedance state of the parasitic unit 30 connected to the first RF switch 41 and the second RF switch 42 is exactly opposite to the load impedance state of the parasitic unit 30 connected to the third RF switch 43 and the fourth RF switch 44. Specifically, the two parasitic units 30 in the first row are in an open circuit state while the two parasitic units 30 in the second row are in a short circuit state. The above-mentioned load impedance difference along the row direction will introduce a significant phase gradient between the first row and the second row antenna units 10, thereby causing the synthesized beam to be positively deflected on the H plane. Figure 5 As shown, in state 2, the forward deflection angle of the synthesized beam is approximately +23°, while the gain remains at approximately 9.6 dBi. Although there is a decrease of approximately 1.9 dB compared to state 1, it still remains at a high level. This indicates that by adjusting the voltage polarity combination of the first and second row RF switches 40, a forward beam scan at a large angle in the H-plane can be achieved without significantly sacrificing the gain, effectively compensating for the grating lobe problem that may be caused by the increased cell spacing when the large array is scanned in the H-plane.

[0038] Conversely, such as Figure 3 As shown, when the first RF switch 41 and the second RF switch 42 switch to the first control terminal V1 being 2V and the second control terminal V2 being -2V, while the third RF switch 43 and the fourth RF switch 44 maintain the first control terminal V1 being -2V and the second control terminal V2 being 2V, the reconfigurable antenna subarray 100 enters state 3. At this time, the two parasitic elements 30 in the first row become short-circuited while the two parasitic elements 30 in the second row return to open-circuit state. The phase gradient direction between rows is reversed accordingly, causing the synthesized beam to deflect in the opposite direction on the H-plane. Figure 6 As shown, in state 3, the reverse deflection angle of the synthesized beam is approximately -23°, and the gain is also approximately 9.6 dBi, which is highly consistent with the performance in state 2. This demonstrates that the forward and reverse deflection of the H-plane has good symmetry and stability, further confirming that the bidirectional beam reconstruction of the H-plane can be reliably achieved by simply switching the voltage polarity of the control terminals of the first and second row RF switches 40, and no obvious gain fluctuations or pattern distortions occurred during the entire deflection process.

[0039] Similarly, such as Figure 3As shown, by adjusting the voltage combination relationship between the first RF switch 41 and the third RF switch 43, for example, in states 4 and 5, the load impedance states connected to the parasitic units 30 of the first and second columns respectively present opposite distribution patterns. The pattern-reconfigurable antenna subarray 100 can also achieve forward and reverse deflection functions on the E-plane. Figure 7 and Figure 8 As shown, in the E-plane deflection state, the deflection angle of the synthesized beam is approximately ±10°, with a gain of approximately 9.9 dBi to 10.0 dBi. This gain value is slightly higher than the 9.6 dBi in the H-plane deflection state, reflecting that the phase compensation effect of the secondary radiation field of parasitic element 30 on the main radiation field is more sufficient and the energy loss is relatively smaller in the E-plane deflection mode. It is worth noting that the E-plane deflection angle is smaller than the H-plane deflection angle. This is related to the difference in physical dimensions in the row and column directions of the 2×2 subarray and the anisotropy of the coupling strength of parasitic element 30. However, even so, the ±10° E-plane deflection capability is sufficient to meet the basic requirements for E-plane scanning coverage in most large array applications.

[0040] All the aforementioned deflection state switching relies solely on the reconfiguration of the DC voltage polarity of the first control terminal V1 and the second control terminal V2 of the four RF switches 40. No changes are required to the physical structure of any feed network 20 or the addition of additional phase-shifting devices. This fully demonstrates the flexibility and practicality of the pattern reconfigurable antenna subarray 100 in W-band large-spacing application scenarios. It helps to dynamically suppress grating lobes and alleviate gain drop effects when multiple pattern reconfigurable antenna subarrays 100 are spliced ​​together to form a large-scale array, thereby significantly improving the scanning performance and environmental adaptability of the overall large array.

[0041] like Figure 9 As shown, in one embodiment, the pattern-reconfigurable antenna subarray 100 further includes: The encapsulation carrier 50 includes a first metal layer 51, a second metal layer 52, a third metal layer 53, a fourth metal layer 54, a fifth metal layer 55, a sixth metal layer 56 and a seventh metal layer 57 stacked sequentially from bottom to top. Multiple coaxial vertical structures 60, each coaxial vertical structure 60 penetrates the seventh metal layer 57, the sixth metal layer 56, the fifth metal layer 55, the fourth metal layer 54 and the third metal layer 53 and extends to the second metal layer 52. One end of each coaxial vertical structure 60 is electrically connected to a parasitic unit 30 and the other end is electrically connected to a radio frequency switch 40. Among them, multiple parasitic units 30 are disposed in the seventh metal layer 57; Multiple antenna elements 10 are disposed on the sixth metal layer 56 and are disposed in a one-to-one correspondence with multiple parasitic elements 30; The fifth metal layer 55 is a ground plane. The fifth metal layer 55 is provided with metallized vias. The antenna element 10 is connected to the fifth metal layer 55 through the metallized vias. The feed network 20 is disposed on the fourth metal layer 54, and the feed network 20 passes through the fifth metal layer 55 and extends to the sixth metal layer 56 to be connected to the antenna element 10 in the near field. The third metal layer 53 is a floor layer. The third metal layer 53 has a through groove 501 that passes through the second metal layer 52 and the first metal layer 51 on the side away from the fourth metal layer 54. Multiple radio frequency switches 40 are disposed in the through groove 501, and the material of the periphery of the through groove 501 is an insulating material. The open circuit structure 70 and the short circuit structure 80 are disposed on the first metal layer 51.

[0042] In this embodiment, the seventh metal layer 57 is used to house multiple parasitic units 30, and the sixth metal layer 56 is used to house multiple antenna units 10. The multiple antenna units 10 and multiple parasitic units 30 are arranged one-to-one in the vertical direction to achieve near-field coupling between the antenna units 10 and the parasitic units 30. The fifth metal layer 55 serves as a ground plane, providing a ground reference plane for the antenna units 10 and enabling reliable grounding of the antenna units 10 through metallized vias. The fourth metal layer 54 is used to house a feed network 20, which penetrates the fifth metal layer 55 and extends to the vicinity of the sixth metal layer 56. This feed network forms a near-field coupling connection with the antenna units 10 in a non-contact manner, reducing signal loss caused by direct electrical contact between the feed network 20 and the antenna units 10, and avoiding the process complexity associated with direct soldering between the feed network 20 and the antenna units 10.

[0043] On the side of the third metal layer 53 opposite to the fourth metal layer 54, there is a through slot 501 that penetrates the second metal layer 52 and the first metal layer 51. This through slot 501 is used to accommodate multiple RF switches 40, allowing the RF switches 40 to be embedded inside the packaging carrier 50 rather than being mounted on the surface of the packaging carrier 50. This helps to reduce the overall cross-sectional height of the pattern reconfigurable antenna subarray 100 and enhance structural stability. The peripheral wall of the through slot 501 is made of insulating material. This design reduces the electromagnetic interference of the second metal layer 52 to the RF switches 40, and avoids short circuits or parasitic coupling between the RF switches 40 and the second metal layer 52, ensuring the reliability of the RF switches 40 in switching states under different voltage polarity control. The third metal layer 53 also serves as a ground plane, providing an independent grounding shielding environment for the RF switches 40, reducing the electromagnetic influence of the parasitic unit 30, antenna unit 10, and feed network 20 on the RF switches 40, and improving the signal integrity and isolation of the pattern reconfigurable antenna subarray 100 in a multi-layer stacked structure.

[0044] Each type of coaxial vertical structure 60 has one end electrically connected to a parasitic unit 30, and the other end extending through the seventh metal layer 57, the sixth metal layer 56, the fifth metal layer 55, the fourth metal layer 54, and the third metal layer 53 to the second metal layer 52, so as to be electrically connected to an RF switch 40 by means of gold wire bonding or flip bonding. This type of coaxial vertical structure 60 forms a shielded transmission channel between the seventh metal layer 57 and the second metal layer 52, which can effectively suppress radiation loss and interlayer crosstalk of signals during vertical transmission. Since the quasi-coaxial vertical structure 60 extends to the second metal layer 52, the first metal layer 51 is provided so that the bottom end of the quasi-coaxial vertical structure 60 is not exposed to the external environment, thereby forming physical protection and electromagnetic enclosure for the bottom end of the quasi-coaxial vertical structure 60. At the same time, the open circuit structure 70 and short circuit structure 80 provided on the first metal layer 51 serve as the terminal load when the radio frequency switch 40 is switched, and work with the quasi-coaxial vertical structure 60 to limit the load impedance state of the parasitic unit 30 to an open circuit state or a short circuit state.

[0045] The dielectric layer between the sixth metal layer 56 and the fifth metal layer 55 is defined as the core layer. The core layer material is CCL-HL972LF type LD, with a dielectric constant εr=3.4, a loss tangent tanδ=0.004, and a thickness of 250um. All dielectric layers other than the core layer are defined as prepreg layers. For example, the dielectric layer between the seventh metal layer 57 and the sixth metal layer 56 is a prepreg layer, made of GHPL-970LF type LD, with a dielectric constant εr=3.4, a loss tangent tanδ=0.004, and a thickness of 100um. The first metal layer 51, second metal layer 52, third metal layer 53, fourth metal layer 54, fifth metal layer 55, sixth metal layer 56, and seventh metal layer 57 are all made of Copper, with a dielectric constant εr=1, a loss tangent tanδ=0, and a thickness of 17um. It can be noted that the thickness of the core layer is greater than that of the prepreg layer. This is from a processing perspective, so that the sixth metal layer 56 carrying the antenna element 10 has sufficient structural support strength, avoiding bending or deformation of the antenna element 10 during the pressing of the packaging carrier 50 and subsequent processing. This ensures the consistency of the spacing between the antenna element 10 and the parasitic element 30, maintains the stability of the near-field coupling connection and the repeatability of the pattern reconstruction performance.

[0046] like Figure 10 and Figure 11 As shown, in one embodiment, a plurality of radio frequency switches 40 are located between the third metal layer 53 and the second metal layer 52.

[0047] In this embodiment, since the first metal layer 51 has an open-circuit structure 70 and a short-circuit structure 80, if the RF switch 40 is too close to the first metal layer 51, the edge field or parasitic capacitance generated by the open-circuit structure 70 and the short-circuit structure 80 can easily couple to the RF switch 40, thereby affecting the state switching accuracy and response speed of the RF switch 40 under different voltage polarities. By placing the RF switch 40 between the third metal layer 53 and the second metal layer 52, and using the second metal layer 52 as an intermediate shielding reference surface, the distance between the RF switch 40 and the first metal layer 51 can be increased in physical space, weakening the electromagnetic coupling strength between them. At the same time, the third metal layer 53, as a ground layer, can also provide a stable ground reference for the RF switch 40, so that the RF switch 40 is in a relatively independent electromagnetic environment inside the packaging carrier 50. This is beneficial to maintaining the signal isolation and operational stability between the RF switch 40, parasitic element 30, antenna element 10 and feed network 20 in the pattern reconfigurable antenna subarray 100.

[0048] like Figures 9 to 11 As shown, in one embodiment, the power supply network 20 includes: The first type of coaxial vertical transition structure 21 penetrates the fifth metal layer 55 and extends to the sixth metal layer 56 to be connected to the antenna element 10 in the near field coupling. The second type of coaxial vertical transition structure 22 penetrates the third metal layer 53, the second metal layer 52 and the first metal layer 51. The second type of coaxial vertical transition structure 22 is used to connect external radio frequency circuits. The power divider 23 is located on the fourth metal layer 54. The power divider 23 connects the first type of coaxial vertical transition structure 21 and the second type of coaxial vertical transition structure 22.

[0049] In this embodiment, to achieve wide bandwidth and good impedance matching, the feed network 20 employs an L-shaped probe coupling feed method. The radio frequency signal originates from the power divider 23 of the fourth metal layer 54, is transmitted to the sixth metal layer 56 through a first-type coaxial vertical transition structure 21 with a characteristic impedance of 50 ohms, and then undergoes near-field coupling with the antenna element 10 via the horizontal portion of the L-shaped probe. The first-type coaxial vertical transition structure 21 consists of a central signal via and surrounding ground vias. This L-shaped probe, as an extension of the first-type coaxial vertical transition structure 21 in the sixth metal layer 56, forms a capacitive coupling region between its horizontal portion and the antenna element 10. Combined with the inductive characteristics of the first-type coaxial vertical transition structure 21, impedance compensation can be achieved over a wide frequency band, maintaining a high level of energy transmission efficiency between the feed network 20 and the antenna element 10. Meanwhile, the second type of coaxial vertical transition structure 22 passes through the third metal layer 53, the second metal layer 52 and the first metal layer 51 and is connected to the power divider 23. The signal introduced by the external radio frequency circuit is distributed by the power divider 23 and then transmitted to the antenna element 10 through the first type of coaxial vertical transition structure 21. The entire signal path relies on the metal layer and vias inside the packaging carrier 50 to complete the vertical interconnection, avoiding the radiation loss and parasitic effects caused by traditional surface traces. This is beneficial to maintaining the feeding stability and pattern purity of the pattern reconfigurable antenna subarray 100 under high frequency operation.

[0050] like Figures 9 to 11 As shown, in one embodiment, each antenna element 10 includes two electric dipoles 11 symmetrically arranged about a first direction, each electric dipole 11 includes two metal patches 111 symmetrically arranged about a second direction, each metal patch 111 is connected to a fifth metal layer 55 through a metallized via, a gap is formed between the two metal patches 111 of the same electric dipole 11, the two gaps constitute a magnetic dipole 12, the second direction is intersecting the first direction; the end of the first type of coaxial vertical transition structure 21 away from the power divider 23 is located between the two electric dipoles 11 to be near-field coupled to the two electric dipoles 11.

[0051] In this embodiment, each antenna element 10 utilizes two electric dipoles 11 symmetrically arranged about a first direction and a magnetic dipole 12 formed by two gaps to form a magnetoelectric dipole 11 antenna structure. Two metal patches 111 symmetrically arranged about a second direction are connected to the fifth metal layer 55 through metallized vias to obtain a stable ground reference potential. The second direction intersects with the first direction, causing the electric dipoles 11 and magnetic dipoles 12 to be spatially orthogonal, which helps to extend the impedance bandwidth of the antenna element 10 and improve the symmetry of the radiation pattern. The end of the first type of coaxial vertical transition structure 21 furthest from the power divider 23 is located between the two electric dipoles 11. This position is at the symmetry center region of the two electric dipoles 11, allowing the first type of coaxial vertical transition structure 21 to simultaneously form a balanced near-field coupling connection with both electric dipoles 11, avoiding the impact on the radiation performance of the antenna element 10 due to inconsistent excitation amplitude or phase of the two electric dipoles 11 caused by feed position offset. Meanwhile, the magnetic dipole 12 formed by the two gaps and the two electric dipoles 11 are coupled to each other in the near field region, so that the antenna element 10 maintains good impedance matching characteristics and stable gain level in a wide frequency band. Combined with the central feeding layout of the first type of coaxial vertical transition structure 21, the radiation efficiency and pattern purity of each antenna element 10 in the pattern reconfigurable antenna subarray 100 are further improved.

[0052] like Figures 9 to 11 As shown, in one embodiment, each metal patch 111 includes a first side 1111 and a second side 1112 disposed opposite to each other along a second direction, and a third side 1113 and a fourth side 1114 disposed opposite to each other along a first direction. The third side 1113 includes a bent section 11131 and a straight section 11132. The bent section 11131 extends from the first side 1111 toward the second side 1112 and is inclined toward the side away from the fourth side 1114. The straight section 11132 extends from the bent section 11131 toward the second side 1112.

[0053] In this embodiment, the third side 1113 of each metal patch 111 forms an irregular contour through a combination of a bent section 11131 and a straight section 11132. The bent section 11131 extends from the first side 1111 toward the second side 1112 and is inclined away from the fourth side 1114, causing the current path of each metal patch 111 in the region near the first side 1111 to deflect, thereby adjusting the current distribution and equivalent inductance-capacitance parameters on the surface of each metal patch 111. This, combined with the stable radiating arm structure formed by the straight section 11132 extending from the bent section 11131 toward the second side 1112, helps to broaden the operating bandwidth of each antenna element 10 and improve impedance matching characteristics. Under this structure, as... Figures 12 to 14As shown, electromagnetic simulation results indicate that each antenna element 10 exhibits a return loss of less than -10 dB in a wide bandwidth from 85 GHz to 104 GHz, with a relative bandwidth of 20.2%. At the center frequency of 94 GHz, the actual gain of each antenna element 10 is approximately 6.8 dBi, and the radiation efficiency is higher than 88%. Furthermore, as... Figures 15 to 17 As shown, each antenna element 10 exhibits good unidirectional radiation characteristics in both the E-plane and H-plane with a front-to-back ratio exceeding 40dB. The radiation pattern shape remains highly stable in the range of 85GHz to 100GHz. The above performance verifies the optimization effect of the geometric configuration of the bent segment 11131 and the straight segment 11132 in each metal patch 111 on the radiation characteristics of each antenna element 10, and also establishes a reliable unit-level performance reference for the overall design of the subsequent pattern-reconfigurable antenna subarray 100.

[0054] This invention also proposes a pattern-reconfigurable phased array antenna, which includes multiple pattern-reconfigurable antenna subarrays 100 as described above. The multiple pattern-reconfigurable antenna subarrays 100 are arranged in a rectangular array, and the spacing between two adjacent pattern-reconfigurable antenna subarrays 100 is not less than 1.3 times the operating wavelength. The specific structure of the pattern-reconfigurable antenna subarray 100 is as described in the above embodiments. Since this pattern-reconfigurable phased array antenna adopts all the technical solutions of all the above embodiments, it has at least all the beneficial effects brought about by the technical solutions of the above embodiments, which will not be described in detail here.

[0055] In this embodiment, the 8×8 pattern reconfigurable phased array antenna consists of 16 2×2 pattern reconfigurable antenna subarrays 100 to achieve phased array beam scanning. As the target scanning angle changes, each 2×2 pattern reconfigurable antenna subarray 100 can switch between states 1, 2, 3, 4, and 5. When scanning only in the H-plane, a combination of states 1, 2, and 3 is used; when scanning only in the E-plane, a combination of states 1, 4, and 5 is used. The introduction of the parasitic element 30 increases the size of each 2×2 pattern reconfigurable antenna subarray 100, ensuring that the spacing between adjacent pattern reconfigurable antenna subarrays 100 along the E-plane and H-plane directions is at least 1.3 times the operating wavelength. At the center frequency, when the grating lobe level is -10dB, the E-plane scanning angle range is ±20° and the H-plane scanning angle range is ±30°. Figure 18 and Figure 19As shown, when the scanning angle is 0°, all 2×2 pattern reconfigurable antenna subarrays 100 use state 1, at which point the 8×8 pattern reconfigurable phased array antenna reaches its maximum gain of 23.5 dBi. For H-plane scanning, states 2 and 3 are used. The gain decreases as the scanning angle increases. The gain is approximately 22.3 dBi when scanning to ±10°, approximately 21.0 dBi when scanning to ±20°, approximately 20.6 dBi when scanning to ±30°, and the grating lobe level increases significantly and the gain drops to 18.9 dBi when scanning to ±40°. Thanks to the pattern reconfigurability, even with a large spacing between two adjacent pattern reconfigurable antenna subarrays 100, the scanning loss remains below 3 dB when scanning to ±30° in the H-plane. For E-plane scanning, due to the smaller deflection angles in states 4 and 5, the gains at ±10°, ±20°, and ±30° are 22.2 dBi, 20.2 dBi, and 18.2 dBi, respectively. Because pattern reconfigurability is achieved in the W-band with low scanning loss, and the spacing between two adjacent pattern reconfigurable antenna subarrays 100 is increased, this 8×8 pattern reconfigurable phased array antenna is suitable for heterogeneous integrated phased arrays, and the saturated output power of the T / R component of a single channel can exceed 20dBm.

[0056] The present invention also proposes a millimeter-wave communication system, which includes a pattern-reconfigurable phased array antenna. The specific structure of the pattern-reconfigurable phased array antenna is as described in the above embodiments. Since this millimeter-wave communication system adopts all the technical solutions of all the above embodiments, it has at least all the beneficial effects brought about by the technical solutions of the above embodiments, which will not be described in detail here.

[0057] The above description is merely an exemplary embodiment of the present invention and does not limit the patent scope of the present invention. Any equivalent structural transformations made using the contents of the present invention specification and drawings under the technical concept of the present invention, or direct / indirect applications in other related technical fields, are included within the patent protection scope of the present invention.

Claims

1. A pattern-reconfigurable antenna subarray, characterized in that, include: Multiple antenna elements are arranged in a rectangular array, and the distance between two adjacent antenna elements is not less than 1.3 times the operating wavelength; A feeding network is connected to the antenna elements in the near field, and is used to feed the incoming radio frequency signal into the antenna elements to excite the antenna elements to generate the main radiation field. Multiple parasitic units are configured and connected in near field to each of the multiple antenna units, and are used to sense the main radiation field of the corresponding antenna unit and generate a secondary radiation field. Each parasitic unit is connected to a load. Multiple radio frequency switches, each radio frequency switch having a first port, a second port, a third port, a first control terminal, and a second control terminal, the multiple first ports being connected one-to-one with the multiple parasitic units, the second port being connected to a short-circuit structure, the third port being connected to an open-circuit structure, the first control terminal and the second control terminal being used to receive DC voltage signals of the same magnitude but opposite polarity; By changing the voltage polarity combination of the DC voltage signals connected to the first and second control terminals of the multiple radio frequency switches, the load impedance state connected to each parasitic unit is switched between an open-circuit state and a short-circuit state, thereby regulating the phase of the secondary radiation field to reconstruct the radiation pattern of the pattern-reconstructable antenna subarray.

2. The pattern-reconfigurable antenna subarray as described in claim 1, characterized in that, The number of the antenna unit and the number of the radio frequency switch are four each; When the first control terminals of the four radio frequency switches are all connected to DC voltage signals of the same polarity, the load impedance connected to the four parasitic units is in an open circuit state, so that the radiation pattern does not deflect and points to the normal. When the first control terminals of the two RF switches in the first row of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals of the two RF switches in the second row is opposite, the load impedance states of the two parasitic units in the same row are the same, and the load impedance states of the parasitic units in the two rows are opposite, so that the composite beam of the pattern reconfigurable antenna subarray is deflected on the H-plane. When the first control terminals of the two RF switches in the first column of the rectangular array are connected to DC voltage signals of the same polarity, and the polarity of the DC voltage signals connected to the first control terminals of the two RF switches in the second column is opposite, the load impedance states of the two parasitic units in the same column are the same, and the load impedance states of the parasitic units in the two columns are opposite, so that the synthesized beam is deflected on the E plane.

3. The pattern-reconfigurable antenna subarray as described in claim 2, characterized in that, When the first control terminal of the two RF switches in the first row is connected to a negative DC voltage signal, the load impedance of the two parasitic units in the first row is in an open circuit state, so that the synthesized beam will be positively deflected on the H plane. When the first control terminals of the two RF switches in the first row are connected to a positive DC voltage signal, the load impedance of the two parasitic units in the first row is in a short-circuit state, so that the synthesized beam will be deflected in the reverse direction on the H plane. When the first control terminals of the two RF switches in the first column are connected to a negative DC voltage signal, the load impedance connected to the two parasitic units in the first column is in an open circuit state, so that the synthesized beam will be positively deflected on the E plane. When the first control terminals of the two RF switches in the first column are connected to a positive DC voltage signal, the load impedance of the two parasitic units in the first column is in a short-circuit state, so that the synthesized beam is deflected in the reverse direction on the E plane.

4. The pattern-reconfigurable antenna subarray as described in claim 1, characterized in that, The pattern-reconfigurable antenna subarray also includes: The packaging carrier includes a first metal layer, a second metal layer, a third metal layer, a fourth metal layer, a fifth metal layer, a sixth metal layer and a seventh metal layer stacked sequentially from bottom to top; Multiple quasi-coaxial vertical structures, each of the quasi-coaxial vertical structures passing through the seventh metal layer, the sixth metal layer, the fifth metal layer, the fourth metal layer and the third metal layer and extending to the second metal layer, one end of each quasi-coaxial vertical structure being electrically connected to a parasitic unit and the other end being electrically connected to a radio frequency switch; Among them, a plurality of the parasitic units are disposed in the seventh metal layer; Multiple antenna elements are disposed on the sixth metal layer and are configured one-to-one with multiple parasitic elements; The fifth metal layer is a ground plane, and the fifth metal layer is provided with metallized vias. The antenna unit is connected to the fifth metal layer through the metallized vias. The feed network is disposed on the fourth metal layer, and the feed network passes through the fifth metal layer and extends to the sixth metal layer to be connected to the antenna element in the near field. The third metal layer is a floor layer. The third metal layer has a through groove on the side opposite to the fourth metal layer that passes through the second metal layer and the first metal layer. A plurality of radio frequency switches are disposed in the through groove, and the material of the peripheral wall of the through groove is an insulating material. The open-circuit structure and the short-circuit structure are disposed on the first metal layer.

5. The pattern-reconfigurable antenna subarray as described in claim 4, characterized in that, Multiple radio frequency switches are located between the third metal layer and the second metal layer.

6. The pattern-reconfigurable antenna subarray as described in claim 4, characterized in that, The power supply network includes: The first type of coaxial vertical transition structure penetrates the fifth metal layer and extends to the sixth metal layer to be near-field coupled to the antenna element; The second type of coaxial vertical transition structure penetrates the third metal layer, the second metal layer and the first metal layer. The second type of coaxial vertical transition structure is used to connect to external radio frequency circuits. A power divider is disposed on the fourth metal layer, and the power divider connects the first type of coaxial vertical transition structure and the second type of coaxial vertical transition structure.

7. The pattern-reconfigurable antenna subarray as described in claim 6, characterized in that, Each antenna element includes two electric dipoles symmetrically arranged about a first direction, and each electric dipole includes two metal patches symmetrically arranged about a second direction. Each metal patch is connected to the fifth metal layer through a metallized via. A gap is formed between the two metal patches of the same electric dipole, and the two gaps constitute a magnetic dipole. The second direction intersects the first direction. The end of the first type of coaxial vertical transition structure away from the power divider is located between the two electric dipoles to be near-field coupled to the two electric dipoles.

8. The pattern-reconfigurable antenna subarray as described in claim 7, characterized in that, Each of the metal patches includes a first side and a second side disposed opposite to each other along a second direction, and a third side and a fourth side disposed opposite to each other along a first direction. The third side includes a bent section and a straight section. The bent section extends from the first side toward the second side and is inclined toward a side away from the fourth side. The straight section extends from the bent section toward the second side.

9. A pattern-reconfigurable phased array antenna, characterized in that, It includes multiple pattern reconfigurable antenna subarrays as described in any one of claims 1 to 8, wherein the multiple pattern reconfigurable antenna subarrays are arranged in a rectangular array, and the spacing between two adjacent pattern reconfigurable antenna subarrays is not less than 1.3 times the operating wavelength.

10. A millimeter-wave communication system, characterized in that, This includes the pattern-reconfigurable phased array antenna as described in claim 9.

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