A three-level dual active bridge converter model predictive control method and device

Abstract

The application provides a three-level dual active bridge converter model predictive control method and device, which comprises the following steps: collecting the upper capacitor voltage and the lower capacitor voltage of the three-level dual active bridge converter at the current time, and calculating the capacitor voltage difference between the upper capacitor voltage and the lower capacitor voltage; substituting the control degree of the current operating point into the inductance current mathematical model corresponding to the power mode to calculate the inductance current parameter of the current operating point; calculating the virtual parameter at the current time according to the recursive least square algorithm; calculating the predicted correction phase angle amplitude according to the capacitor voltage difference, the inductance current parameter and the virtual parameter; multiplying the correction phase angle amplitude by the direction function to obtain the actual correction phase angle quantity; converting the actual correction phase angle quantity into a driving signal and outputting the driving signal to control the operation of the converter. The application realizes the capacitor voltage balance of the three-level dual active bridge converter, reduces the capacitor voltage ripple, and improves the control robustness.

Description

Technical Field

[0001] This invention relates to the field of power electronic converter control technology, and in particular to a model predictive control method and device for a three-level dual active bridge converter. Background Technology

[0002] Currently, grid-connected power generation from renewable energy sources such as photovoltaics and wind power has become a core development trend in the new power grid. Simultaneously, to mitigate the randomness, volatility, and intermittency of power generation brought by renewable energy, energy storage systems are also widely deployed in renewable energy power generation systems. For DC equipment such as photovoltaics and energy storage systems, the three-level dual-active bridge converter is a highly promising topology choice. In addition to the advantages of traditional two-level dual-active converters, such as electrical isolation, bidirectional power, and high power density, the three-level dual-active bridge also offers advantages such as stronger voltage isolation capability, wider application scenarios, and higher operating efficiency.

[0003] In a three-level dual active bridge topology, two clamping capacitors are introduced, one upper and one lower. During steady-state operation, the voltages of these clamping capacitors should remain consistent, both at half the DC bus voltage. However, in actual operation, due to factors such as drive signal delay and device parameter drift, the voltages of the clamping capacitors may deviate from their theoretical values. This increases the voltage stress on some power devices, potentially damaging the converter. Traditional capacitor voltage balancing methods introduce a PI (Proportional-Integral) controller loop to compensate for the voltage error between the upper and lower capacitors, correcting the phase angle of the drive signal to achieve active control and balancing of the clamping capacitor voltages. However, the compensation provided by the PI controller is difficult to accurately match the actual voltage error; therefore, the additional compensation increases the voltage ripple of the clamping capacitors. Furthermore, the PI controller's compensation also depends on the PI parameter tuning; inappropriate PI parameters will further increase the capacitor voltage ripple. Larger capacitor voltage ripple will shorten capacitor lifespan, increase capacitor power loss, and increase capacitor size, severely limiting the application prospects of three-level dual active bridge converters. Summary of the Invention

[0004] To address the aforementioned problems in the prior art, this invention provides a model predictive control method and apparatus for a three-level dual active bridge converter, which achieves capacitor voltage balance in the three-level dual active bridge converter while reducing capacitor voltage ripple and improving control robustness.

[0005] To achieve the above objectives, the technical solution adopted by the present invention is as follows: In a first aspect, the present invention provides a model predictive control method for a three-level dual active bridge converter, comprising: Step S1: Collect the upper capacitor voltage and lower capacitor voltage of the three-level dual active bridge converter at the current moment, and calculate the capacitor voltage difference between the upper capacitor voltage and the lower capacitor voltage. Step S2: Substitute the control degree of the current operating point into the mathematical model of the inductor current of the corresponding power mode to calculate the inductor current parameters at the current operating point. Step S3: Calculate the virtual parameters at the current time step using the recursive least squares algorithm; Step S4: Calculate the predicted corrected phase angle magnitude based on the capacitor voltage difference, the inductor current parameter, and the virtual parameter; Step S5: Multiply the magnitude of the corrected phase angle by the direction function to obtain the actual corrected phase angle. Step S6: Convert the actual corrected phase angle into a drive signal and output it to control the converter.

[0006] The beneficial effects of this invention are as follows: by collecting the capacitor voltage difference and using the recursive least squares algorithm to update the virtual parameters in real time, and combining the inductor current parameters and direction function to predict and correct the phase angle amplitude, accurate predictive control of the capacitor voltage of the three-level dual active bridge converter is achieved, effectively reducing capacitor voltage ripple, and maintaining control robustness under conditions such as parameter mismatch, avoiding performance degradation caused by improper parameter tuning of traditional PI controllers, thereby improving the operating stability of the converter and the service life of the capacitor.

[0007] Optionally, the power mode in step S2 includes: The operating range for power mode 1 is defined as {D0≤D1≤D2≤D1+D}; The operating range for power mode 2 is defined as {D1≤D0≤D2≤D1+D}; The operating range for power mode 3 is defined as {D1≤D2≤D0≤D1+D}; Where D0 is the inner phase shift angle of the primary-side full bridge, D1 is the inner phase shift angle between the drive signals of switching transistors Q1 and Q2 in the secondary-side three-level full bridge, D2 is the inner phase shift angle between the drive signals of switching transistors Q3 and Q4 in the secondary-side three-level full bridge, and D is the inner phase shift angle between the drive signals of Q2 and Q3 in the secondary-side three-level full bridge.

[0008] Optionally, the mathematical model of the inductor current in step S2 includes: Inductor current reference value i b The voltage transformation ratio k is defined as: ; ; In the formula, n is the transformer turns ratio, U2 is the converter output voltage, and f s L is the switching frequency. eq U1 is the phase-shifting inductance value, and U1 is the converter input voltage; In power mode 1, the expression for the inductor current at each moment is: ; In the formula, O represents the initial moment of a switching cycle, and A represents the secondary three-level full-bridge switch Q. a1 The conduction timing of the drive signal, B represents the secondary three-level full-bridge switch Q. a2 The conduction timing of the drive signal, C represents the secondary three-level full-bridge switch Q. b3 The turn-on timing of the drive signal, D represents the secondary three-level full-bridge switch Q. b4 The on-time of the drive signal; In power mode 2, the expression for the inductor current at each moment is as follows: ; In power mode 3, the expression for the inductor current at each moment is as follows: .

[0009] Optionally, in step S3, the virtual parameter Z V (k) is defined as: ; In the formula, n is the transformer turns ratio, C is the capacitance of the upper and lower capacitors of the secondary three-level full-bridge, and L... eq This is the inductance value of the phase-shifting inductor.

[0010] Optionally, the expression for the recursive least squares algorithm in step S3 is: ; ; In the formula, Z V (k 1) represents the virtual parameters calculated in the previous switching cycle, ε is the forgetting factor, and P(k) is the covariance matrix at the current time. 1) is the covariance matrix calculated for the previous switching cycle, and A(k) and B(k) are the variable matrix and constant matrix at the current time, respectively.

[0011] Optionally, in step S4, the magnitude m of the phase angle is corrected. a The formula for calculating (k) is: ; In the formula, f s For the switching frequency, ΔU c (k) represents the capacitor voltage difference, Z V (k) is a virtual parameter, U2 is the output voltage, and i LD ∧ (k) is the per-unit value of the inductor current at point D during the conduction time, i LB ∧(k) is the per-unit value of the inductor current at point B during the conduction time, K m This is the upper limit of the amplitude.

[0012] Optionally, the direction function in step S5 is defined as: ; ; ΔU c (k)= U C2 (k) U C1 (k); In the formula, s1(k), s2(k), s3(k), and s4(k) are Q, respectively. a1 Q a2 Q b3 Q b4 The direction function of the corrected phase shift angle applied by the drive signal, sgn() is the sign function, ΔU c (k) represents the capacitor voltage difference, i LD ∧ (k) and i LB ∧ (k) is the per-unit value of the inductor current, U C2 (k) represents the lower capacitor voltage, U C1 (k) represents the voltage across the upper capacitor.

[0013] Optionally, the per-unit values ​​of the inductor current at points B and D at the current moment are defined as follows: ; ; In the formula, i L (B) and i L (D) represent the instantaneous inductor current values ​​at points B and D respectively under the corresponding power modes, i b This is the reference value for the inductor current.

[0014] Optionally, the variable matrix A(k) and the constant matrix B(k) are defined as follows: ; ; In the formula, m a (k 1) The corrected phase angle magnitude calculated for the previous switching cycle, i LD ∧ (k-1) represents the inductor current related to point D at the previous conduction moment and i LB ∧ (k-1) represents the inductor current related quantity at point B during the previous conduction time.

[0015] In a second aspect, the present invention provides a model predictive control device for a three-level dual active bridge converter, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor. When the processor executes the computer program, it implements a model predictive control method for a three-level dual active bridge converter according to the first aspect.

[0016] The technical effects of the three-level dual active bridge converter model predictive control device provided in the second aspect are described in the relevant description of the three-level dual active bridge converter model predictive control method provided in the first aspect. Attached Figure Description

[0017] Figure 1 This is a schematic diagram of the main flow of a model predictive control method for a three-level dual active bridge converter according to an embodiment of the present invention; Figure 2 This is a schematic diagram of the flow framework of a model predictive control method for a three-level dual active bridge converter according to an embodiment of the present invention; Figure 3 This is a topology diagram of a three-level dual active bridge converter according to an embodiment of the present invention.

[0018] Figure 4 This is a schematic diagram of the control degree and modulation strategy of the three-level dual active bridge converter involved in the embodiments of the present invention.

[0019] Figure 5 This is a schematic diagram of phase angle correction for a model predictive control method for a three-level dual active bridge converter according to an embodiment of the present invention.

[0020] Figure 6 The diagram shows the capacitor voltage ripple of the three-level dual active bridge converter involved in the embodiments of the present invention and the prior art involved in this embodiment.

[0021] Figure 7 The diagram shows the capacitor voltage ripple test results of the prior art and this embodiment under parameter mismatch conditions in this invention.

[0022] Figure 8 This is a schematic diagram of the structure of a model predictive control device for a three-level dual active bridge converter according to an embodiment of the present invention.

[0023] Explanation of reference numerals in the attached figures: 1: A model predictive control device for a three-level dual active bridge converter; 2: Processor; 3: Memory. Detailed Implementation

[0024] To better understand the above technical solutions, exemplary embodiments of the present invention will be described in more detail below with reference to the accompanying drawings. Although exemplary embodiments of the present invention are shown in the drawings, it should be understood that the present invention can be implemented in various forms and should not be limited to the embodiments set forth herein. Rather, these embodiments are provided so that the present invention can be understood more clearly and thoroughly, and that the scope of the present invention can be fully conveyed to those skilled in the art.

[0025] Example 1 In existing technologies, the phase angle correction in the capacitor voltage control method of a three-level dual active bridge converter is based on a PI controller. However, the PI controller cannot accurately compensate for actual voltage errors, thus causing external voltage fluctuations and increasing capacitor voltage ripple. Furthermore, the performance of capacitor voltage control based on the PI controller depends on the tuning of the PI parameters; mismatched PI parameters will further worsen the capacitor voltage ripple.

[0026] In this embodiment, the upper and lower capacitor voltages of the three-level dual active bridge converter are acquired at the current moment, and the capacitor voltage difference between the upper and lower capacitor voltages is calculated. The control degree of the current operating point is substituted into the inductor current mathematical model of the corresponding power mode to calculate the inductor current parameters at the current operating point. The virtual parameters at the current moment are calculated according to the recursive least squares algorithm. Based on the capacitor voltage difference, inductor current parameters, and virtual parameters, the predicted corrected phase angle magnitude is calculated. The corrected phase angle magnitude is multiplied by the direction function to obtain the actual corrected phase angle. The actual corrected phase angle is converted into a drive signal and output to control the converter operation, thereby achieving capacitor voltage balance of the three-level dual active bridge converter, reducing capacitor voltage ripple, and improving control robustness. See the discussion below for details.

[0027] Please refer to Figures 1 to 7 A model predictive control method for a three-level dual active bridge converter, such as... Figure 1 and Figure 2 It can be seen that this includes: Step S1: Collect the upper capacitor voltage and lower capacitor voltage of the three-level dual active bridge converter at the current moment, and calculate the capacitor voltage difference between the upper capacitor voltage and the lower capacitor voltage.

[0028] In this embodiment, the capacitor voltage U on the three-level dual active bridge converter at the current time k is collected. C1 (k) and lower capacitor voltage U C2 (k), and calculate the difference ΔU C (k)=U C2 (k) U C1 (k).

[0029] The topology diagram of the three-level dual active bridge converter is as follows: Figure 3 It can be known that...

[0030] Step S2: Substitute the control degree of the current operating point into the mathematical model of the inductor current of the corresponding power mode to calculate the inductor current parameters at the current operating point.

[0031] The power modes in step S2 include: The operating range for power mode 1 is defined as {D0≤D1≤D2≤D1+D}; The operating range for power mode 2 is defined as {D1≤D0≤D2≤D1+D}; The operating range for power mode 3 is defined as {D1≤D2≤D0≤D1+D}; Where D0 is the inner phase shift angle of the primary-side full bridge, D1 is the inner phase shift angle between the drive signals of switching transistors Q1 and Q2 in the secondary-side three-level full bridge, D2 is the inner phase shift angle between the drive signals of switching transistors Q3 and Q4 in the secondary-side three-level full bridge, and D is the inner phase shift angle between the drive signals of Q2 and Q3 in the secondary-side three-level full bridge.

[0032] Therefore, in this embodiment, by limiting the above three power modes, all possible operating ranges of the converter are covered, and the mode can be determined by judging the relationship between the current internal phase shift angle.

[0033] The mathematical model for the inductor current in step S2 includes: Inductor current reference value i b The voltage transformation ratio k is defined as: ; ; In the formula, n is the transformer turns ratio, U2 is the converter output voltage, and f s L is the switching frequency. eq U1 is the phase-shifting inductance value, and U1 is the converter input voltage; In power mode 1, the expression for the inductor current at each moment is: ; In the formula, O represents the initial moment of a switching cycle, and A represents the secondary three-level full-bridge switch Q. a1 The conduction timing of the drive signal, B represents the secondary three-level full-bridge switch Q. a2 The conduction timing of the drive signal, C represents the secondary three-level full-bridge switch Q. b3 The turn-on timing of the drive signal, D represents the secondary three-level full-bridge switch Q. b4 The on-time of the drive signal; In power mode 2, the expression for the inductor current at each moment is as follows: ; In power mode 3, the expression for the inductor current at each moment is as follows: .

[0034] The per-unit values ​​of the inductor current at points B and D at the current moment are defined as follows: ; ; In the formula, i L (B) and i L (D) represent the instantaneous inductor current values ​​at points B and D respectively under the corresponding power modes, i b This is the reference value for the inductor current.

[0035] Therefore, the benchmark value i b The introduction of the voltage ratio k simplifies the expression and facilitates per-unit processing. Points O, A, B, C, and D in the inductor current expression correspond to key moments within a switching cycle, with the per-unit current values ​​at points B and D being directly used in subsequent phase angle correction calculations.

[0036] Step S3: Calculate the virtual parameters at the current time step using the recursive least squares algorithm.

[0037] In step S3, the virtual parameter Z V (k) is defined as: ; In the formula, n is the transformer turns ratio, C is the capacitance of the upper and lower capacitors of the secondary three-level full-bridge, and L... eq This is the inductance value of the phase-shifting inductor.

[0038] The expression for the recursive least squares algorithm in step S3 is as follows: ; ; In the formula, Z V (k 1) represents the virtual parameters calculated in the previous switching cycle, ε is the forgetting factor, and P(k) is the covariance matrix at the current time. 1) is the covariance matrix calculated for the previous switching cycle, and A(k) and B(k) are the variable matrix and constant matrix at the current time, respectively.

[0039] It should be noted that virtual parameters integrate multiple physical parameters into a single comprehensive parameter, so that the controller does not need to know the independent value of each parameter precisely.

[0040] In this embodiment, the forgetting factor ε controls the weight of historical data, and is usually taken as a value slightly less than 1, such as 0.93.

[0041] The variable matrix A(k) and the constant matrix B(k) are defined as follows: ; ; In the formula, m a (k 1) The corrected phase angle magnitude calculated for the previous switching cycle, i LD ∧ (k-1) represents the inductor current related to point D at the previous conduction moment and i LB ∧ (k-1) represents the inductor current related quantity at point B during the previous conduction time.

[0042] Step S4: Calculate the predicted corrected phase angle magnitude based on the capacitor voltage difference, inductor current parameters, and virtual parameters.

[0043] In the three-level dual active bridge converter, the voltages of the upper and lower clamping capacitors can be expressed as: ; In the formula, C X It is the capacitance value of the upper and lower clamping capacitors, i cx It is the current of the upper and lower clamping capacitors.

[0044] Considering the computational delay of one switching cycle in actual operation of data processors such as DSPs (Digital Signal Processors), the capacitor voltage can be further expressed as: ; In the formula, CX (k)> is the average value of the capacitor current over the period [k,k+1].

[0045] At this point, the goal of this embodiment is to minimize the capacitor voltage error; therefore, the cost function of this algorithm is: ; In actual operation, the capacitance values ​​of the upper and lower capacitors are basically close, so C1=C2=C can be taken.

[0046] Then, according to Kirchhoff's current law, the neutral point current i can be obtained. O =i C2 i C1 Based on the above conditions, the cost function can be derived as follows: ; In the formula, O ​​(k)> is the average value of the neutral point current over the period [k,k+1].

[0047] Since the magnitude of the phase angle correction is usually limited to a small range to avoid significant fluctuations in the inductor current, and the magnitude of the inductor current can be approximated as constant within this small range, then... O (k)> can be computed as: ; In the formula, s1(k), s2(k), s3(k), and s4(k) are respectively Q a1 Q a2 Q b3 Q b4 The direction function of the corrected phase shift angle applied by the drive signal. LB (k), i LD (k),i Lb (k),i Ld (k) represents the magnitude of the inductor current at points B, D, b, and d within a switching cycle.

[0048] Consider that the converter uses the same degree of control in the two switching cycles [k,k+1] and [k+1,k+2], meaning the inductor current is the same in both cycles. Simultaneously, due to the half-cycle symmetry of the inductor current, i... LB (k)= i Lb (k) and i LD (k)= i Ld (k). Combining the two conditions above, substitute into... O (k)> The calculation formula and the cost function can be further derived as follows: ; Since the goal of this embodiment is to minimize the cost function, that is, let g c =0. Therefore, after substituting the direction function, the magnitude m of the phase angle correction in step S4 is... a The formula for calculating (k) is: ; In the formula, f s For the switching frequency, ΔU c (k) represents the capacitor voltage difference, Z V (k) is a virtual parameter, U2 is the output voltage, and i LD ∧ (k) is the per-unit value of the inductor current at point D during the conduction time, i LB ∧ (k) is the per-unit value of the inductor current at point B during the conduction time, K m This is the upper limit of the amplitude. ​​

[0049] Therefore, this embodiment establishes the relationship between the capacitor voltage difference and the neutral point current, and simplifies it by utilizing half-cycle symmetry, ultimately obtaining an explicit formula that depends only on the current voltage error, virtual parameters, and per-unit value of the inductor current.

[0050] Step S5: Multiply the magnitude of the corrected phase angle by the direction function to obtain the actual corrected phase angle.

[0051] In step S5, the direction function is defined as follows: ; ; ΔU c (k)= U C2 (k) U C1 (k); In the formula, s1(k), s2(k), s3(k), and s4(k) are Q, respectively. a1 Q a2 Q b3 Q b4 The direction function of the corrected phase shift angle applied by the drive signal, sgn() is the sign function, ΔU c (k) represents the capacitor voltage difference, i LD ∧ (k) and i LB ∧ (k) is the per-unit value of the inductor current, U C2 (k) represents the lower capacitor voltage, U C1 (k) represents the voltage across the upper capacitor.

[0052] In this embodiment, the direction function determines whether the drive signal for each switch leads or lags, and its design principle is to enable the neutral point current to compensate for the current voltage imbalance. The use of the sign function ensures that the correction direction depends only on the polarity of the voltage error and the direction of the inductor current, and is independent of the specific numerical value, thereby simplifying the control logic.

[0053] Step S6: Convert the actual corrected phase angle into a drive signal and output it to control the converter.

[0054] The actual phase correction angle is added to Q according to the direction function. a1 Q a2 Q b3 Q b4 The signals of other complementary switches are automatically adjusted based on the drive signal. The entire control process is executed once in each switching cycle, forming a closed-loop regulation.

[0055] In this embodiment, refer to Figure 4It can be seen that the driving signal Q a1 With drive signal Q a3 Complementary conduction, driving signal Q a2 With drive signal Q a4 Complementary conduction, driving signal Q b1 With drive Q b3 Complementary conduction, driving signal Q b2 With drive signal Q b4 The drive signals are complementary and conduction is activated. Drive signals S1 and S3 are complementary and conduction is activated, as are drive signals S2 and S4. Simultaneously, D0 is the inner phase shift angle of H1, controlling the primary-side full-bridge output voltage U. ab The zero-level width. D1, D2, and D are the inner phase shift angles of H2, controlling the secondary full-bridge output voltage U. cd The zero level and ±1 / 2 level width.

[0056] like Figure 5 As shown, m1, m2, m3, and m4 are Q a1 Q a2 Q b3 Q b4 The corrected phase angle of the drive signal, the magnitude of which is m a .

[0057] like Figure 6 As shown, in this embodiment of the invention, the input voltage is 160V, the output voltage is 400V, the switching frequency is 20kHz, the phase-shifting inductor value is 100uH, the transformer turns ratio is 1:2, the clamping capacitor value is 470uF, the transmission power is 1.2kW, and the limiting K is... m Under a test condition of 0.01, the capacitor voltage ripple in the existing technology based on a traditional PI controller is 2.09V, where the control parameter is K. p =0.02, K i =1. In this embodiment, the capacitor voltage ripple of the capacitor voltage prediction control method based on virtual parameters is 1.82V, which achieves a reduction in capacitor voltage ripple.

[0058] like Figure 7 As shown, in this embodiment of the invention, the input voltage is 160V, the output voltage is 400V, the switching frequency is 20kHz, the phase-shifting inductor value is 100uH, the transformer turns ratio is 1:2, the clamping capacitor value is 470uF, the transmission power is 1.2kW, and the limiting K is... m Under the test condition of K = 0.01, the capacitor voltage ripple based on the traditional PI controller is 9.53V, where the control parameter is set to K. p =0.4, K i=20 to simulate parameter mismatch. In this embodiment, the capacitor voltage ripple of the capacitor voltage prediction control method based on virtual parameters is 1.90V, where the forgetting factor ε is set to 0.95, and the initial virtual parameter is set to 0.7Zv to simulate parameter mismatch. It can be seen that under parameter mismatch conditions, compared to Figure 6 In the prior art, the capacitor voltage ripple based on the traditional PI controller increases significantly, while the capacitor voltage ripple based on the virtual parameter-based capacitor voltage prediction control method in this embodiment does not change significantly.

[0059] In summary, the technical advantages of this application are as follows: Compared with traditional methods that rely on PI parameter tuning, this application avoids the steady-state error and parameter sensitivity issues of PI controllers. During steady-state operation, it can suppress capacitor voltage ripple to a lower level, thereby reducing capacitor charging and discharging losses and extending capacitor lifespan. Simultaneously, this application uses recursive least squares to update virtual parameters in real time, exhibiting adaptive capability to changes in circuit parameters such as inductance and capacitance values. It maintains excellent control robustness under conditions such as parameter mismatch, load disturbances, or temperature drift, without requiring parameter retuning. Finally, this method achieves precise compensation of the drive signal through a prediction mechanism of the direction function and the corrected phase angle amplitude, avoiding the introduction of additional current overshoot, ensuring a smooth inductor current waveform, and further improving the dynamic response performance and reliability of the converter.

[0060] Example 2 Please refer to Figure 8 A three-level dual active bridge converter model prediction control device 1 includes a memory 3, a processor 2, and a computer program stored in the memory 3 and executable on the processor 2. When the processor 2 executes the computer program, it implements the steps in the above embodiment 1.

[0061] Since the systems / devices described in the above embodiments of the present invention are systems / devices used to implement the methods of the above embodiments of the present invention, those skilled in the art can understand the specific structure and modifications of the systems / devices based on the methods described in the above embodiments of the present invention, and therefore will not be repeated here. All systems / devices used in the methods of the above embodiments of the present invention fall within the scope of protection of the present invention.

[0062] Those skilled in the art will understand that embodiments of the present invention can be provided as methods, apparatus, or computer program products. Therefore, the present invention can take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, the present invention can take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code.

[0063] This invention is described with reference to flowchart illustrations and / or block diagrams of methods, apparatus (devices), and computer program products according to embodiments of the invention. It will be understood that each block of the flowchart illustrations and / or block diagrams, as well as combinations of blocks in the flowchart illustrations and / or block diagrams, can be implemented by computer program instructions.

[0064] It should be noted that any reference numerals placed between parentheses in the claims should not be construed as limiting the claims. The word "comprising" does not exclude the presence of components or steps not listed in the claims. The word "a" or "an" preceding a component does not exclude the presence of a plurality of such components. The invention can be implemented by means of hardware comprising several different components and by means of a suitably programmed computer. In claims that enumerate several means, several of these means may be embodied by the same hardware. The use of the terms first, second, third, etc., is merely for convenience of expression and does not indicate any order. These terms can be understood as part of the component names.

[0065] Furthermore, it should be noted that in the description of this specification, the terms "one embodiment," "some embodiments," "embodiment," "example," "specific example," or "some examples," etc., refer to specific features, structures, materials, or characteristics described in connection with that embodiment or example, which are included in at least one embodiment or example of the present invention. In this specification, the illustrative expressions of the above terms do not necessarily refer to the same embodiment or example. Moreover, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or more embodiments or examples. Furthermore, without contradiction, those skilled in the art can combine and integrate the different embodiments or examples described in this specification, as well as the features of different embodiments or examples.

[0066] Although preferred embodiments of the invention have been described, those skilled in the art, upon learning the basic inventive concept, can make other changes and modifications to these embodiments. Therefore, the claims should be interpreted to include both the preferred embodiments and all changes and modifications falling within the scope of the invention.

[0067] Obviously, those skilled in the art can make various modifications and variations to this invention without departing from its spirit and scope. Therefore, if these modifications and variations fall within the scope of the claims of this invention and their equivalents, then this invention should also include these modifications and variations.

Claims

1. A model predictive control method for a three-level dual active bridge converter, characterized in that, include: Step S1: Collect the upper capacitor voltage and lower capacitor voltage of the three-level dual active bridge converter at the current moment, and calculate the capacitor voltage difference between the upper capacitor voltage and the lower capacitor voltage. Step S2: Substitute the control degree of the current operating point into the mathematical model of the inductor current of the corresponding power mode to calculate the inductor current parameters at the current operating point. Step S3: Calculate the virtual parameters at the current time step using the recursive least squares algorithm; Step S4: Calculate the predicted corrected phase angle magnitude based on the capacitor voltage difference, the inductor current parameter, and the virtual parameter; Step S5: Multiply the magnitude of the corrected phase angle by the direction function to obtain the actual corrected phase angle. Step S6: Convert the actual corrected phase angle into a drive signal and output it to control the converter.

2. The model predictive control method for a three-level dual active bridge converter according to claim 1, characterized in that, The power mode in step S2 includes: The operating range for power mode 1 is defined as {D0≤D1≤D2≤D1+D}; The operating range for power mode 2 is defined as {D1≤D0≤D2≤D1+D}; The operating range for power mode 3 is defined as {D1≤D2≤D0≤D1+D}; Where D0 is the inner phase shift angle of the primary-side full bridge, D1 is the inner phase shift angle between the drive signals of switching transistors Q1 and Q2 in the secondary-side three-level full bridge, D2 is the inner phase shift angle between the drive signals of switching transistors Q3 and Q4 in the secondary-side three-level full bridge, and D is the inner phase shift angle between the drive signals of Q2 and Q3 in the secondary-side three-level full bridge.

3. The model predictive control method for a three-level dual active bridge converter according to claim 1, characterized in that, The mathematical model of the inductor current in step S2 includes: Inductor current reference value i b The voltage transformation ratio k is defined as: ； ； In the formula, n is the transformer turns ratio, U2 is the converter output voltage, and f s L is the switching frequency. eq U1 is the phase-shifting inductance value, and U1 is the converter input voltage; In power mode 1, the expression for the inductor current at each moment is: ； In the formula, O represents the initial moment of a switching cycle, and A represents the secondary three-level full-bridge switch Q. a1 The conduction timing of the drive signal, B represents the secondary three-level full-bridge switch Q. a2 The conduction timing of the drive signal, C represents the secondary three-level full-bridge switch Q. b3 The turn-on timing of the drive signal, D represents the secondary three-level full-bridge switch Q. b4 The on-time of the drive signal; In power mode 2, the expression for the inductor current at each moment is as follows: ； In power mode 3, the expression for the inductor current at each moment is as follows: 。 4. The model predictive control method for a three-level dual active bridge converter according to claim 1, characterized in that, The virtual parameter Z in step S3 V (k) is defined as: ； In the formula, n is the transformer turns ratio, C is the capacitance of the upper and lower capacitors of the secondary three-level full-bridge, and L... eq This is the inductance value of the phase-shifting inductor.

5. The model predictive control method for a three-level dual active bridge converter according to claim 4, characterized in that, The expression for the recursive least squares algorithm in step S3 is: ； ； In the formula, Z V (k 1) represents the virtual parameters calculated in the previous switching cycle, ε is the forgetting factor, and P(k) is the covariance matrix at the current time. 1) is the covariance matrix calculated for the previous switching cycle, and A(k) and B(k) are the variable matrix and constant matrix at the current time, respectively.

6. The model predictive control method for a three-level dual active bridge converter according to claim 1, characterized in that, In step S4, the magnitude m of the phase angle correction is... a The formula for calculating (k) is: ； In the formula, f s For the switching frequency, ΔU c (k) represents the capacitor voltage difference, Z V (k) is a virtual parameter, U2 is the output voltage, and i LD ∧ (k) is the per-unit value of the inductor current at point D during the conduction time, i LB ∧ (k) is the per-unit value of the inductor current at point B during the conduction time, K m This is the upper limit of the amplitude.

7. The model predictive control method for a three-level dual active bridge converter according to claim 1, characterized in that, The direction function in step S5 is defined as follows: ； ； ΔU c (k)= U C2 (k) AT C1 (k); In the formula, s1(k), s2(k), s3(k), and s4(k) are Q, respectively. a1 Q a2 Q b3 Q b4 The direction function of the corrected phase shift angle applied by the drive signal, sgn() is the sign function, ΔU c (k) represents the capacitor voltage difference, i LD ∧ (k) and i LB ∧ (k) is the per-unit value of the inductor current, U C2 (k) represents the lower capacitor voltage, U C1 (k) represents the voltage across the upper capacitor.

8. The model predictive control method for a three-level dual active bridge converter according to claim 2, characterized in that, The per-unit values ​​of the inductor current at points B and D at the current moment are defined as follows: ； ； In the formula, i L (B) and i L (D) represent the instantaneous inductor current values ​​at points B and D respectively under the corresponding power modes, i b This is the reference value for the inductor current.

9. The model predictive control method for a three-level dual active bridge converter according to claim 5, characterized in that, The variable matrix A(k) and the constant matrix B(k) are defined as follows: ； ； In the formula, m a (k 1) The corrected phase angle magnitude calculated for the previous switching cycle, i LD ∧ (k-1) represents the inductor current related to point D at the previous conduction moment and i LB ∧ (k-1) represents the inductor current related quantity at point B during the previous conduction time.

10. A model predictive control device for a three-level dual active bridge converter, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, characterized in that, When the processor executes the computer program, it implements the model predictive control method for a three-level dual active bridge converter as described in any one of claims 1 to 9.

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