Online control method for synchronous rectification of bidirectional CLLC converter under hybrid modulation

Abstract

The application discloses a kind of mixed modulation under bidirectional CLLC converter synchronous rectification online control method, this method is by collecting input voltage, output voltage, output current, switching frequency, inner phase shift angle and resonant cavity parameter, establish the unified state plane model suitable for PFM and IPS two kinds of modulation mode and CCM / DCM mode switching;Equivalent parameter, working track and mode boundary are calculated on-line, the synchronous rectification time offset corresponding to the zero-crossing of the reduced secondary side resonant current is obtained, and the mode-related initial value and single Newton method iteration are used to complete fast correction, and the synchronous rectification driving signal is output.The application does not need to increase high-frequency sensor, can realize high-precision synchronous rectification control in wide working range, reduce conduction loss, improve the efficiency and dynamic performance of bidirectional CLLC converter.

Description

Technical Field

[0001] This invention belongs to the field of power electronic control technology, specifically relating to an online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation. Background Technology

[0002] With the rapid development of energy storage systems, DC microgrids, and electric vehicle on-board chargers, isolated DC-DC converters with bidirectional power flow capabilities have become core power interfaces. CLLC (capacitor-inductor-inductor-capacitor) resonant converters, due to their symmetrical resonant cavity structure, wide soft-switching range, and high power density, have been widely used in bidirectional energy conversion scenarios.

[0003] Especially for wide voltage gain applications, bidirectional CLLC converters typically require a hybrid control strategy combining pulse frequency modulation (PFM) and internal phase-shift modulation (IPS). Under this hybrid modulation, the high-frequency operating characteristics of the converter are more complex, including not only continuous conduction mode (CCM) and discontinuous conduction mode (DCM), but also dynamic switching between modes. Under these conditions, if the traditional time-domain analytical method is used (A. Sankar, A. Mallik and A. Khaligh. Synchronous rectification phase tracking method based on extended harmonics in CLLC converters, IEEE Transactions on Industrial Electronics, Vol. 66, No. 8, pp. 6592-6603, 2019), it is necessary to establish and solve cumbersome nonlinear transcendental equations for each subdivision mode, which places a great burden on the online real-time operation of the digital controller. If the fitting or lookup table method is used (H. Li, Y. Sun and S. Wang. Bidirectional control of SiC bidirectional CLLC electric vehicle charger based on synchronous rectification and input ripple current feedforward based on fitting model, IEEE Transactions on Industrial Electronics, Vol. 70, No. 9, September 2023), facing the high-dimensional control variable space composed of switching frequency and phase shift angle, not only is a huge amount of data storage required, but the calibration process is also extremely cumbersome and has poor robustness to changes in resonance parameters.

[0004] In recent years, state-plane analysis (Z. Sun, T. Zhang, C. Ruan, Q. Yu, and C. Feng. Synchronous Rectification Method for CLLC Resonant Converters Based on State Trajectory Model, Applied Science, Vol. 15, No. 8, p. 4372, 2025) has shown great potential in synchronous rectification calculations due to its ability to transform complex time-domain nonlinear waveforms into intuitive geometric trajectories, making it possible to directly extract accurate switching timing from state trajectories. However, most existing state-plane-based synchronous rectification strategies are limited to single frequency modulation operation conditions and lack a unified mathematical description and processing mechanism for PFM and IPS hybrid modulation and CCM / DCM boundary crossing.

[0005] Therefore, breaking down the theoretical barriers between different modulation methods and conduction modes, and exploring a unified online synchronous rectification control method that is compatible with hybrid modulation strategies, free from high-frequency sensing dependence, and has both high computational accuracy and low computational overhead, has become a key technical challenge for the efficient operation of bidirectional CLLC resonant converters. Summary of the Invention

[0006] In view of the above, the present invention provides an online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation, which uses a unified state plane model as the core to realize online high-precision calculation of synchronous rectification timing under hybrid modulation.

[0007] An online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation includes the following steps: (1) Collect the input voltage, output voltage, output current and switching frequency of the bidirectional CLLC converter, and call the pre-stored resonant cavity parameters; (2) Establish a unified state plane model based on the equivalent circuits of different conduction stages in the first half of the switching cycle, and map the synchronous rectification process under PFM and IPS to the unified state plane model. (3) Based on the resonant cavity parameters and the unified state plane model, the decoupling parameters, equivalent circuit parameters and trajectory center coordinates are obtained online to obtain the state plane trajectory under the current working mode; (4) Identify the current operating mode based on the relationship between the switching frequency and the resonant frequency and the conduction mode boundary criterion under IPS; under the current operating mode, construct an implicit equation with the target conduction angle as the unknown variable based on the geometric relationship of the state plane trajectory; (5) In the current working mode, after performing a first-order Taylor approximation on the trigonometric function, the initial value of the target conduction angle is obtained, and then substituted into the implicit equation to obtain the target conduction angle through a single iteration of the Newton method. (6) Convert the obtained target conduction angle into the time offset of synchronous rectification, and generate and output the final synchronous rectification drive signal based on the time offset.

[0008] The bidirectional CLLC converter includes a primary-side active full-bridge, a secondary-side synchronous rectifier full-bridge, and a resonant cavity connecting the two. The resonant cavity includes a primary-side resonant inductor, a secondary-side resonant inductor, a magnetizing inductor, and a primary-side resonant capacitor. 、 The secondary resonant capacitor and the high-frequency isolation transformer are configured such that one end of the primary resonant inductor is connected to the positive AC terminal of the primary active full-bridge circuit, the other end of the primary resonant inductor is connected to one end of the magnetizing inductor and the same-name terminal of the primary winding of the high-frequency isolation transformer, the other end of the magnetizing inductor is connected to one end of the primary resonant capacitor and the opposite-name terminal of the primary winding of the high-frequency isolation transformer, the other end of the primary resonant capacitor is connected to the negative AC terminal of the primary active full-bridge circuit, one end of the secondary resonant inductor is connected to the same-name terminal of the secondary winding of the high-frequency isolation transformer, the other end of the secondary resonant inductor is connected to the positive AC terminal of the secondary synchronous rectifier full-bridge circuit, one end of the secondary resonant capacitor is connected to the opposite-name terminal of the secondary winding of the high-frequency isolation transformer, and the other end of the secondary resonant capacitor is connected to the negative AC terminal of the secondary synchronous rectifier full-bridge circuit.

[0009] Further, in step (3), the decoupling parameters are calculated using the following expression: in: k and g For decoupling parameters, C 1 represents the capacitance value of the primary resonant capacitor. C 2 represents the capacitance value of the secondary resonant capacitor. L 1 represents the inductance value of the primary resonant inductor. L 2 represents the inductance value of the secondary resonant inductor. L m The inductance value of the magnetizing inductor. The capacitance value of the secondary resonant capacitor referred to the primary side and , The inductance value of the secondary resonant inductor referred to the primary side and , n Δ represents the turns ratio of the primary winding to the secondary winding of the high-frequency isolation transformer, and Δ is an intermediate variable.

[0010] Furthermore, the equivalent circuit parameters in step (3) include the equivalent capacitance value. C eq and equivalent inductance value L eq ,in Ceq = C 1, .

[0011] Furthermore, the calculation of the trajectory center coordinates in step (3) is expressed as follows: In PFM-DCM mode , , ; In PFM-CCM mode , , ; In IPS-CCM mode , , ; In IPS-DCM mode , , ; in: O 1 represents the center coordinate of the state plane trajectory during conduction phase I. O 2 represents the center coordinates of the state plane trajectory during conduction phase II. O 3 represents the center coordinates of the state plane trajectory during conduction phases III, IV, or V. V C2A' The initial voltage value of the secondary resonant capacitor at the beginning of the half-switching cycle and , The output voltage of the bidirectional CLLC converter is referred to the primary side voltage value. , V 1 and V 2 represents the input voltage and output voltage of the bidirectional CLLC converter, respectively. I oN This refers to the normalized output current of the bidirectional CLLC converter. f N intermediate variable and f N = f s / f eq , f s The switching frequency of the bidirectional CLLC converter. f eq The resonant frequency of the equivalent circuit and .

[0012] Furthermore, the specific criteria for identifying the current working mode in step (4) are as follows: when and fs < f r In this case, the current working mode is determined to be PFM-DCM mode; when and f s ≥ f r In this case, the current working mode is determined to be PFM-CCM mode; when and f s ≥ f r And Δ boundary If the value is less than 0, the current working mode is determined to be IPS-CCM mode; when and f s ≥ f r And Δ boundary If the value is ≥0, then the current working mode is determined to be IPS-DCM mode; in: f r The resonant frequency of the resonant cavity is and Δ boundary To determine the indicators and , X load , , As an intermediate variable, , This refers to the inner phase shift angle of the switching transistors in the primary-side active full-bridge circuit. Z eq The equivalent impedance of the equivalent circuit and .

[0013] Furthermore, the implicit equation expression constructed in step (4) is as follows: If the current working mode is PFM-DCM mode ; If the current working mode is PFM-CCM mode ; If the current operating mode is IPS-CCM mode ; If the current operating mode is IPS-DCM mode ; in: F ( i 3) Indicates the target conduction angle i 3 is an implicit equation with unknown variables. Z ( i 3) is an auxiliary vector. X θ intermediate variable and I is a 2×2 identity matrix.

[0014] Furthermore, the auxiliary vector Z ( i The expression for 3) is as follows: in: Red ( i ) is about the variable i The standard rotation matrix, Red e ( i ) is about the variable i The generalized rotation matrix, i 1 represents the conduction angle of stage I. i 2 is the conduction angle for stage II. d The trajectory correction coefficient and If the current operating mode is PFM-CCM mode or IPS-CCM mode, If the current operating mode is PFM-DCM mode or IPS-DCM mode, If the current working mode is PFM-DCM mode or PFM-CCM mode, i 2=0; if the current working mode is IPS-CCM mode, If the current operating mode is IPS-DCM mode, .

[0015] Furthermore, in step (5), the target conduction angle is calculated using the following expression. i 3: in: i init The initial value of the target conduction angle is obtained by using sin i 3 approximately equals i 3. cosplay i 3 is approximately equal to 1 for the first-order approximation of implicit equations F ( i 3) Solve for the result; F ( i init ) indicates that the initial value i init replace i 3 Substitute into implicit equations F ( i 3) yields the function value on the left side of the equals sign.F' ( i init ) is the function value F ( i init The derivative of ).

[0016] Furthermore, in step (6), the expression Δ t offset = i 3 / oh The time offset Δ of synchronous rectification is calculated. t offset ,in oh The equivalent angular frequency of the equivalent circuit. If the current operating mode is PFM-CCM mode or IPS-CCM mode, If the current operating mode is PFM-DCM mode or IPS-DCM mode, .

[0017] A computer device includes a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the above-described online synchronous rectification control method for a bidirectional CLLC converter under hybrid modulation.

[0018] A computer-readable storage medium storing a computer program, which, when executed by a processor, implements the above-described online synchronous rectification control method for a bidirectional CLLC converter under hybrid modulation.

[0019] Compared with the prior art, the present invention has the following beneficial technical effects: 1. Based on a unified state plane model, this invention unifies different working regions under hybrid modulation and CCM / DCM mode switching into the same synchronous rectification solution framework, avoiding the need to establish independent algorithms for different regions.

[0020] 2. This invention can complete the online calculation of synchronous rectification using only the existing low-frequency sampling of the controller, without the need to add high-frequency sensors, nor rely on lookup tables and complex offline calibration, making it easy for DSP (Digital Signal Processor) to implement in real time.

[0021] 3. This invention has high synchronous rectification calculation accuracy and good robustness over a wide operating range. Experimental results show that the synchronous rectification timing error in the IPS light load region can be controlled within 8 ns, the error in the PFM region can be controlled within 15 ns, and the peak efficiency of the prototype can reach 98.06%. Attached Figure Description

[0022] Figure 1 This is a schematic diagram of the bidirectional CLLC resonant converter topology to which this invention applies.

[0023] Figure 2 This is a flowchart illustrating the online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to the present invention.

[0024] Figure 3 This is a typical operating waveform diagram of a bidirectional CLLC converter during forward transmission under PFM modulation. Figure 3 (a) corresponds to the PFM-CCM working mode, and (b) corresponds to the PFM-DCM working mode.

[0025] Figure 4 This is a typical operating waveform diagram of a bidirectional CLLC converter under IPS modulation during forward transmission. Figure 4 (a) corresponds to the IPS-CCM working mode, and (b) corresponds to the IPS-DCM working mode.

[0026] Figure 5 This is a schematic diagram of the equivalent circuit and decoupling circuit for each stage of the half-switching cycle during forward transmission. Figure 5 (a) corresponds to the equivalent circuit of the five stages of the half-switching cycle during forward transmission, and (b) corresponds to the decoupling circuit of the five stages of the half-switching cycle during forward transmission.

[0027] Figure 6 This is a schematic diagram of the unified state plane trajectory of the bidirectional CLLC converter of the present invention under different operating modes during forward transmission. Figure 6 In the diagram, (a) corresponds to the unified state plane trajectory under PFM-CCM mode, (b) corresponds to the unified state plane trajectory under PFM-DCM mode, (c) corresponds to the unified state plane trajectory under IPS-CCM mode, and (d) corresponds to the unified state plane trajectory under IPS-DCM mode.

[0028] Figure 7 This is a typical waveform diagram of the bidirectional CLLC converter in the PFM-CCM operating region during forward transmission in an embodiment of the present invention.

[0029] Figure 8 This is a typical waveform diagram of the bidirectional CLLC converter in the PFM-DCM operating region during forward transmission in an embodiment of the present invention.

[0030] Figure 9 This is a typical waveform diagram of the bidirectional CLLC converter in the IPS-CCM operating region during forward transmission in an embodiment of the present invention.

[0031] Figure 10 This is a typical waveform diagram of the bidirectional CLLC converter in the IPS-DCM operating region during forward transmission in an embodiment of the present invention. Detailed Implementation

[0032] To describe the present invention in more detail, the technical solution of the present invention will be described in detail below with reference to the accompanying drawings and specific embodiments.

[0033] like Figure 1 As shown, the bidirectional CLLC resonant converter applicable to this embodiment includes a primary-side active full-bridge, a secondary-side synchronous rectified full-bridge, and a primary-side resonant inductor. L 1. Secondary resonant inductor L 2. Magnetizing inductor L m Primary resonant capacitor C 1. Secondary resonant capacitor C 2. And the high-frequency isolation transformer, the primary side active full bridge consists of four switching transistors with anti-parallel diodes. S 1~ S The secondary-side synchronous rectifier full-bridge consists of four switching transistors with anti-parallel diodes. S 5~ S The system consists of 8 components. Because the resonant cavity structure exhibits symmetry in both forward and reverse power transmission, the synchronous rectification online control method of this invention can be applied to both bidirectional operating conditions. Figure 1 middle V 1 represents the input voltage. V 2 represents the output voltage. u ab The voltage at the midpoint of the arm of the primary-side active full-bridge is [voltage value]. u cd This is the midpoint voltage of the bridge arm of the secondary-side synchronous rectification full bridge; v C1 Primary resonant capacitor C The voltage on 1, v C2 Secondary resonant capacitor C The voltage on 2, i L1 For primary resonant inductors L The current on 1, i L2 Secondary resonant inductor L The current on 2, i Lm Magnetizing inductor L m The current on it.

[0034] In this embodiment, the converter employs a hybrid modulation strategy combining PFM and IPS. PFM is primarily used for frequency regulation in the high-gain region, while IPS is used to reduce frequency offset in the light-load region and extend the high-efficiency operating range. Correspondingly, the converter may operate in four typical regions during operation: PFM-CCM, PFM-DCM, IPS-CCM, and IPS-DCM, with typical waveforms as shown below. Figure 3 and Figure 4 As shown, Figure 3 and Figure 4 middle V gs This indicates the drive signal, and the subscript number corresponds to the specific switch transistor number.

[0035] Taking the first half-switching cycle as the analysis object, the bidirectional CLLC converter under hybrid modulation experiences a maximum of five conduction stages. PFM mainly involves stages I, III, and V, while IPS mainly involves stages I, II, III, and IV; stage I corresponds to... S 1 and S 4 conduction, S 5 and S 8 conduction; corresponding in stage II S 2 and S 4 conduction, S 5 and S 8 conduction; corresponding in stage III S 2 and S 3 conduction, S 5 and S 8 conduction; corresponding in stage IV S 2 and S 4 conduction, S 5~ S 8 shutdowns; corresponding to stage V S 1 and S 4 conduction, S 5~ S 8. Turn-off. The second half of the switching cycle also has five stages, namely Stage I. '、 Phase II '、 Phase III '、 Phase IV '、 Phase V '， The corresponding switching state is symmetrical to the first half of the switching cycle. Figure 5 Equivalent circuits and decoupling circuits for each stage are presented. By modeling the circuits of each stage in a unified manner, the synchronous rectification timing problem can be transformed into a state-plane geometric problem.

[0036] When establishing the unified state plane model, two independent, decoupled state variables are selected to describe the resonant cavity dynamics, and the primary side state plane is used as the main analysis plane for synchronous rectification. During continuous conduction, the state trajectory is circular; during intermittent conduction, due to the disappearance of the secondary clamp, the state trajectory is elliptical. The unified state plane trajectories under different operating modes are as follows: Figure 6 As shown.

[0037] To achieve pattern recognition, in the PFM region, the switching frequency can be used as a reference. f s With resonant frequencyf r The relationship between them distinguishes between continuous and intermittent conduction; in the IPS region, the operating mode is identified by constructing a CCM / DCM boundary criterion. The boundary criterion takes the coincidence of the zero-crossing time of the reduced secondary resonant current and the turn-off time of the phase-shifting bridge arm as the boundary condition, and establishes the mode discrimination quantity Δ by combining the half-cycle steady-state symmetry relationship. boundary This allows for the differentiation between CCM and DCM under IPS.

[0038] After pattern recognition is completed, the synchronous rectification time offset is calculated based on the unified state plane model. This offset is defined as the signed time difference from the primary side reference commutation time to the zero-crossing time of the converted secondary side resonant current. In the PFM-CCM and IPS-CCM regions, the offset typically manifests as delayed commutation; in the PFM-DCM and IPS-DCM regions, the offset typically manifests as early turn-off.

[0039] To improve online computation efficiency, this embodiment reduces the synchronous rectification timing solution to an implicit equation containing only a single conduction angle variable. Approximate initial values ​​are constructed for different operating modes, and then rapid correction is achieved through a single iteration of the Newton-Raphson algorithm. In the IPS-DCM region, the initial seed is adaptively selected based on the residual size under different expansion variables, thus balancing computational speed and solution accuracy.

[0040] Figure 2 The specific implementation flow of the synchronous rectification online control method of the present invention is given. The controller first determines whether it has entered the hybrid modulation region, and then updates the equivalent parameters according to the sampled input quantity, completes the working mode identification, initial value generation, single iteration correction and time offset calculation based on geometric relationship, and finally outputs the synchronous rectification drive signal. The whole process only relies on the low-frequency variables that the controller can directly collect or know, and does not require additional high-frequency detection circuit.

[0041] The following is the derivation of the formula for the synchronous rectification time offset. First, the secondary side resonance quantity is converted to the primary side, and defined as follows: in: n This represents the turns ratio between the primary and secondary sides of the transformer.

[0042] Depend on Figure 3 Figure 4 It can be seen that the hybrid control has different operating modes, resulting in different operating stages under different modes. Due to the symmetry of the circuit in the positive and negative half-cycles, the following calculations only consider the positive half-cycle, and the stages are defined as follows: The continuous conduction mode (PFM-CCM) of frequency modulation control occurs in the following stages: Stage I 、 III; The stages in which the intermittent conduction mode (PFM-DCM) of frequency modulation control occurs are: Stage I 、 V; The stages in which the continuous conduction mode (IPS-CCM) of phase-shift control occurs are: Stage I 、 II, III; The phase-shift control method's intermittent conduction mode (IPS-DCM) occurs in the following stages: Stage I 、 II, IV.

[0043] Define two decoupled state variables v Ceq and i Leq This is used to represent a fourth-order CLLC resonator as a second-order state-plane model. , Decoupling coefficients k and g It is obtained by matching the coefficients of the original state equations and the equivalent second-order model for each conduction stage, and can be pre-calculated under the same set of resonance parameters.

[0044] In continuous conduction mode, the time-domain state equation of the CLLC topology is: in: .

[0045] The equivalent model after decoupling can be written as: By comparing the two equations above, the decoupling coefficients can be calculated. k , g and equivalent parameters C eq , L eq , V eq1 , V eq2 : in: .

[0046] In discontinuous conduction mode, since the current is discontinuous in stages IV and V, it has a special equivalent model, which can be written as: The equivalent parameters of stage IV and V can be obtained by combining the above formula. C eq , L eq , V eq1 , Veq2 : in: V C2A' Secondary resonant capacitor C 2. Initial voltage value at the start of the half-switching cycle.

[0047] Define variables and take the normalized variable , During the continuous conduction phase, i.e., phases I, II, and III, the state trajectories satisfy the circular trajectory equation; during the intermittent conduction phase, the secondary current is zero, the secondary clamping effect disappears, and the equivalent resonant branch... L 1. L m and C 1. Dominant. At this point, the same state variables are still used, but the ratios of the normalized voltage axis and current axis are different. The trajectory satisfies the elliptical equation. Trajectory correction coefficients are used to unify the trajectory equations of the two: in: v CeqN This is the normalized voltage of the equivalent capacitance in the equivalent circuit. V CeqN0 The initial value of the normalized voltage of the equivalent capacitance is given. V eq1N and V eq2N For the normalized excitation voltage source in the equivalent circuit, the normalized reference voltage value is V 1, i LeqN This is the normalized current of the equivalent inductance in the equivalent circuit. I LeqN0 The initial normalized current value is the equivalent inductance, and the normalized reference current value is... V 1 / Z eq , , e This is the trajectory correction coefficient. Therefore, the center coordinates of the state plane trajectory are: In PFM-DCM mode , , ; In PFM-CCM mode , , ; In IPS-CCM mode , , ; In IPS-DCM mode , , ; in: O 1 represents the center coordinate of the state plane trajectory during conduction phase I. O 2 represents the center coordinates of the state plane trajectory during conduction phase II. O 3 represents the center coordinates of the state plane trajectory during conduction phases III, IV, or V.

[0048] Let the adjacent boundary points within half a switching cycle be... A =[ X A , Y A ] T , B =[ X B , Y B ] T , D =[ X D , Y D ] T The steady-state half-period symmetry point is A '=- A The state plane trajectories of the four working modes are as follows: Figure 6 As shown.

[0049] Since each operating mode contains at most three main conduction phases within half a switching cycle, the corresponding conduction angle is defined as follows: i 1. i 2 and i 3. The angle constraint can then be uniformly written as: The initial boundary points are determined below. A x-coordinate X A By the definition of equivalent capacitance voltage, point A The x-coordinate can be represented as: in: Point A The voltage of the secondary capacitor referred to the primary side.

[0050] Normalized output current I oN It can be obtained from the average value of the secondary current over half a switching cycle, from which the following can be calculated.V C2NA . On the other hand, normalized output current I oN It can also be calculated from the difference between the primary resonant current and the excitation current: in: i LmN For the current flowing through the magnetizing inductor L m The normalized excitation current.

[0051] Since the excitation current trajectory and secondary conduction interval differ under different operating modes, the above equation should be integrated piecewise according to the clamping state and conduction interval of each stage. For PFM-DCM, the duration of stage I is determined by the resonant frequency. The corresponding time is approximated; for IPS-DCM, the secondary current is zero during the IV phase duration, and v C2 Since it remains unchanged, the output current integral is not considered in this interval. At this point, the normalized voltage of the primary-side capacitor at the starting point D of stage IV can be written as: .

[0052] Under the above approximation conditions, the normalized output current obtained by integrating the primary current can be uniformly expressed as: in: .

[0053] By combining the secondary-side output current expression with the mode expressions in the normalized output current expression, we obtain... X A for: according to Figure 6 The geometric relationships shown are given, and the rotation matrix equations between A, B, and D are written respectively: in: .

[0054] Combination A' =- A With angular constraints, intermediate variables can be eliminated. B , D and i 1. i 2. Define auxiliary vectors Z ( i 3): Where: I represents the second-order identity matrix.

[0055] After elimination, the synchronous rectification timing solution is reduced to only containing i 3. Implicit equations for a single variable: because F ( i 3) = 0 usually does not have an explicit closed-form solution; the controller first determines the solution based on... i 3. Smaller characteristics using sin i 3 approximately equals i 3. cosplay i The first-order approximation of 3 being approximately equal to 1 yields the initial value. i init Perform the Newton correction again: In the IPS-DCM region, i 2 and i The relative magnitude of 3 will change with the inward phase angle. Changes. To avoid errors caused by a single unfolding point, this embodiment constructs different... i 2 and i 3 represents the candidate initial values ​​for the expanded variables. The corresponding residuals are calculated, and the one with the smaller residual is selected as the initial seed for the Newton correction.

[0056] The corrected i 2 and i After 3, the angle constraints are unified as follows: i 3. Convert it into synchronous rectification time offset Δ according to the actual angular frequency of the stage. t offset = i 3 / oh ,in oh The equivalent angular frequency is defined as follows at different stages: For delayed commutation scenarios, the switching on and off of the synchronous rectifier bridge arm are both shifted later than the primary side reference commutation time: , For cases of early shutdown, the activation time remains synchronized with the primary side reference time, while the shutdown time is advanced by a negative offset. , ;in t ref,on This refers to the turn-on time of the primary-side switch transistor. t ref,off This is the turn-off time of the primary-side switch.

[0057] When phase shift angle At this time, the converter operates at PFM, and its CCM / DCM boundary is determined by the switching frequency. f s With resonant frequency f r The relationship is established; when When the converter operates in IPS mode, it is necessary to further determine whether it operates in CCM or DCM based on the critical boundary conditions.

[0058] Therefore, the key to mode determination lies in identifying the CCM / DCM critical boundary under IPS modulation. Under the CCM / DCM critical state of IPS, the secondary resonant current... i L2 The zero-crossing time coincides exactly with the turn-off time of the phase-shifting bridge arm, therefore we have i 3 = 0, at this time we have: There is a geometric relationship at the critical state. D = A '=- A Therefore, there is X A =- X D ,Right now: Bring this relationship into X A The discriminant is obtained from the relation: Therefore, the operating mode under hybrid control is determined as follows: Therefore, the controller only needs to sample V 1. V 2. I o , f s , By calling pre-stored resonant cavity parameters, equivalent parameter updates and operating mode identification can be completed online. i 3. Solve for Δ t offset It can convert without the need for additional high-frequency current or high-frequency voltage detection circuits.

[0059] This embodiment was verified on an 8.6kW bidirectional CLLC converter prototype, which was implemented based on the TMS320F280049 digital signal processor.

[0060] like Figure 7~Figure 10The experimental results show that under four typical operating conditions—PFM-DCM, PFM-CCM, IPS-DCM, and IPS-CCM—the synchronous rectification drive pulse always falls within the effective conduction range of the secondary resonant current. The maximum phase deviation between the calculated CCM / DCM boundary and the extracted accurate time-domain model under IPS is approximately 1.79 degrees. Regarding synchronous rectification time error, the average absolute error in the IPS light-load region can be controlled within 8 ns, and in the PFM region within 15 ns. The prototype achieves a peak efficiency of 98.06%, representing a maximum improvement of approximately 1.87% compared to the asynchronous rectification scheme, while maintaining a bounded offset under ±5% disturbance of the resonant parameters. Figure 7~Figure 10 middle V gs This indicates the drive signal, and the subscript corresponds to the specific switching transistor.

[0061] The above description of the embodiments is provided to enable those skilled in the art to understand and apply the present invention. Those skilled in the art can readily make various modifications to the above embodiments and apply the general principles described herein to other embodiments without creative effort. Therefore, the present invention is not limited to the above embodiments, and any improvements and modifications made to the present invention by those skilled in the art based on the disclosure thereof should be within the scope of protection of the present invention.

Claims

1. A method for online synchronous rectification control of a bidirectional CLLC converter under hybrid modulation, characterized in that, Includes the following steps: (1) Collect the input voltage, output voltage, output current and switching frequency of the bidirectional CLLC converter, and call the pre-stored resonant cavity parameters; (2) Establish a unified state plane model based on the equivalent circuits of different conduction stages in the first half of the switching cycle, and map the synchronous rectification process under PFM and IPS to the unified state plane model. (3) Based on the resonant cavity parameters and the unified state plane model, the decoupling parameters, equivalent circuit parameters and trajectory center coordinates are obtained online to obtain the state plane trajectory under the current working mode; (4) Identify the current operating mode based on the relationship between the switching frequency and the resonant frequency and the conduction mode boundary criterion under IPS; under the current operating mode, construct an implicit equation with the target conduction angle as the unknown variable based on the geometric relationship of the state plane trajectory; (5) In the current working mode, after performing a first-order Taylor approximation on the trigonometric function, the initial value of the target conduction angle is obtained, and then substituted into the implicit equation to obtain the target conduction angle through a single iteration of the Newton method. (6) Convert the obtained target conduction angle into the time offset of synchronous rectification, and generate and output the final synchronous rectification drive signal based on the time offset.

2. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 1, characterized in that, In step (3), the decoupling parameters are calculated using the following expression: in: k and g For decoupling parameters, C 1 represents the capacitance value of the primary resonant capacitor. C 2 represents the capacitance value of the secondary resonant capacitor. L 1 represents the inductance value of the primary resonant inductor. L 2 represents the inductance value of the secondary resonant inductor. L m The inductance value of the magnetizing inductor. The capacitance value of the secondary resonant capacitor referred to the primary side and , The inductance value of the secondary resonant inductor referred to the primary side and , n Δ represents the turns ratio of the primary winding to the secondary winding of the high-frequency isolation transformer, and Δ is an intermediate variable.

3. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 2, characterized in that, The equivalent circuit parameters in step (3) include the equivalent capacitance value. C eq and equivalent inductance value L eq ,in C eq = C 1, .

4. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 3, characterized in that, The calculation of the trajectory center coordinates in step (3) is expressed as follows: In PFM-DCM mode , , ; In PFM-CCM mode , , ; In IPS-CCM mode , , ; In IPS-DCM mode , , ; in: O 1 represents the center coordinate of the state plane trajectory during conduction phase I. O 2 represents the center coordinates of the state plane trajectory during conduction phase II. O 3 represents the center coordinates of the state plane trajectory during conduction phases III, IV, or V. V C2A' The initial voltage value of the secondary resonant capacitor at the beginning of the half-switching cycle and , The output voltage of the bidirectional CLLC converter is referred to the primary side voltage value. , V 1 and V 2 represents the input voltage and output voltage of the bidirectional CLLC converter, respectively. I oN This refers to the normalized output current of the bidirectional CLLC converter. f N intermediate variable and f N = f s / f eq , f s The switching frequency of the bidirectional CLLC converter. f eq The resonant frequency of the equivalent circuit and .

5. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 4, characterized in that, The specific criteria for identifying the current working mode in step (4) are as follows: when and f s < f r In this case, the current working mode is determined to be PFM-DCM mode; when and f s ≥ f r In this case, the current working mode is determined to be PFM-CCM mode; when and f s ≥ f r And Δ boundary If the value is less than 0, the current operating mode is determined to be IPS-CCM mode; when and f s ≥ f r And Δ boundary If the value is ≥0, then the current working mode is determined to be IPS-DCM mode; in: f r The resonant frequency of the resonant cavity is and Δ boundary To determine the indicators and , X load , , As an intermediate variable, , This refers to the inner phase shift angle of the switching transistors in the primary-side active full-bridge circuit. Z eq The equivalent impedance of the equivalent circuit and .

6. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 5, characterized in that, The implicit equation expression constructed in step (4) is as follows: If the current working mode is PFM-DCM mode ; If the current working mode is PFM-CCM mode ; If the current operating mode is IPS-CCM mode ; If the current operating mode is IPS-DCM mode ; in: F ( θ 3) Indicates the target conduction angle θ 3 is an implicit equation with unknown variables. Z ( θ 3) is an auxiliary vector. X θ intermediate variable and I is a 2×2 identity matrix; The auxiliary vector Z ( θ The expression for 3) is as follows: in: Rot ( θ ) is about the variable θ The standard rotation matrix, Rot e ( θ ) is about the variable θ The generalized rotation matrix, θ 1 represents the conduction angle of stage I. θ 2 is the conduction angle for stage II. δ The trajectory correction coefficient and If the current operating mode is PFM-CCM mode or IPS-CCM mode, If the current operating mode is PFM-DCM mode or IPS-DCM mode, If the current working mode is PFM-DCM mode or PFM-CCM mode, θ 2=0; if the current working mode is IPS-CCM mode, If the current operating mode is IPS-DCM mode, .

7. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 6, characterized in that, In step (5), the target conduction angle is calculated using the following expression. θ 3: in: θ init The initial value of the target conduction angle is obtained by using sin θ 3 approximately equals θ 3. cosplay θ 3 is approximately equal to 1 for the first-order approximation of implicit equations F ( θ 3) Solve for the result; F ( θ init ) indicates that the initial value θ init replace θ 3 Substitute into implicit equations F ( θ 3) yields the function value on the left side of the equals sign. F' ( θ init ) is the function value F ( θ init The derivative of ).

8. The online control method for synchronous rectification of a bidirectional CLLC converter under hybrid modulation according to claim 6, characterized in that, In step (6), the expression Δ t offset = θ 3 / ω The time offset Δ of synchronous rectification is calculated. t offset ,in ω The equivalent angular frequency of the equivalent circuit. If the current operating mode is PFM-CCM mode or IPS-CCM mode, If the current operating mode is PFM-DCM mode or IPS-DCM mode, .

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